FAIRCHILD AN-8035

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AN-8035
Design Consideration for Boundary Conduction Mode
Power Factor Correction (PFC) Using FAN7930
1. Introduction
around the PFC controller and saves total BOM cost. The
internal proprietary logic for detecting input voltage
greatly improves the stability of PFC operation. Together
with the maximum switching frequency clamping at
300kHz, FAN7930 can limit inductor current within predesigned value at one or two cycles of the AC-inputabsent test to simulate a sudden blackout. Due to the
startup-without-overshoot design, audible noise from
repetitive OVP triggering is eliminated. Protection
functions include output over-voltage, over-current, openfeedback, and under-voltage lockout.
This application note presents practical step-by-step
design considerations for a Boundary-Conduction-Mode
(BCM) Power-Factor-Correction (PFC) converter
employing Fairchild PFC controller, FAN7930. It
includes designing the inductor and Zero-CurrentDetection (ZCD) circuit, selecting the components, and
closing the control loop. The design procedure is verified
through an experimental 200W prototype converter.
Unlike the Continuous Conduction Mode (CCM)
technique often used at this power level, BCM offers
inherent zero-current switching of the boost diodes (no
reverse-recovery losses), which permits the use of less
expensive diodes without sacrificing efficiency.
An Excel®-based design tool is available with this
application note and the design result is shown with the
calculation results as an example.
FAN7930 has a PFC-ready pin to acknowledge when
PFC output voltage reaches stable operation range. This
signal can be used as the VCC trigger signal for another
power stage controller after PFC stage or be transferred to
the secondary side to synchronize the operation with PFC
voltage condition. This simplifies the external circuit
Visit http://www.fairchildsemi.com/design_tools/ to
download the design tool.
LBOOST
VCC
VOUT
Q1
FAN7930
8
5
3
2
VCC
Out
ZCD
7
CS
4
INV
1
Comp
RDY
GND
6
VOUT acknowledge
Figure 1. Typical Application Circuit
Excel® is a registered trademark of Microsoft Corporation.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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AN-8035
2. Operation Principle of BCM Boost PFC Converter
The most widely used operation modes for the boost
converter are continuous conduction mode (CCM) and
boundary conduction mode (BCM). These two descriptive
names refer to the current flowing through the energy
storage inductor of the boost converter, as depicted in
Figure 2. As the names indicate, the inductor current in
CCM is continuous; while in BCM, the new switching
period is initiated when the inductor current returns to
zero, which is at the boundary of continuous conduction
and discontinuous conduction operations. Even though the
BCM operation has higher RMS current in the inductor
and switching devices, it allows better switching condition
for the MOSFET and the diode. As shown in Figure 2, the
diode reverse recovery is eliminated and a fast-recovery
diode is not needed. The MOSFET is also turned on with
zero current, which reduces the switching loss.
A by-product of BCM is that the boost converter runs
with variable switching frequency that depends primarily
on the selected output voltage, the instantaneous value of
the input voltage, the boost inductor value, and the output
power delivered to the load. The operating frequency
changes as the input current follows the sinusoidal input
voltage waveform, as shown in Figure 3. The lowest
frequency occurs at the peak of sinusoidal line voltage.
Figure 3. Operation Waveforms of BCM PFC
The voltage-second balance equation for the inductor is:
VIN ( t ) ⋅ t ON = (VOUT − VIN ( t )) ⋅ t OFF
(1)
where VIN(t) is the rectified line voltage and VOUT is the
output voltage.
The switching frequency of BCM boost PFC converter is:
fSW =
=
Figure 2. CCM vs. BCM Control
The fundamental idea of BCM PFC is that the inductor
current starts from zero in each switching period, as
shown in Figure 3. When the power transistor of the boost
converter is turned on for a fixed time, the peak inductor
current is proportional to the input voltage. Since the
current waveform is triangular; the average value in each
switching period is proportional to the input voltage. In a
sinusoidal input voltage, the input current of the converter
follows the input voltage waveform with very high
accuracy and draws a sinusoidal input current from the
source. This behavior makes the boost converter in BCM
operation an ideal candidate for power factor correction.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
1
1 VOUT − VIN ( t )
=
⋅
+ tOFF tON
VOUT
tON
1
tON
⋅
VOUT − VIN ,PK ⋅ sin (2π ⋅ fLINE ⋅ t )
(2)
VOUT
where VIN,PK is the amplitude of the line voltage and fLINE
is the line frequency.
Figure 4 shows how the MOSFET on time and switching
frequency changes as output power decreases. When the
load decreases, as shown in the right side of Figure 4, the
peak inductor current diminishes with reduced MOSFET
on time and, therefore, the switching frequency increases.
Since this can cause severe switching losses at light-load
condition and too-high switching frequency operation
may occur at startup, the maximum switching frequency
of FAN7930 is limited to 300kHz.
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AN-8035
3. Startup without Overshoot and
AC-Absent Detection
Figure 4. Frequency Variation of BCM PFC
Since the design of the filter and inductor for a BCM PFC
converter with variable switching frequency should be at
minimum frequency condition, it is worthwhile to
examine how the minimum frequency of BCM PFC
converter changes with operating conditions.
Feedback control speed of the PFC is typically quite slow.
Due to the slow response, there is a gap between output
voltage and feedback control. That is why Over-Voltage
Protection (OVP) is critical at the PFC controller. Voltage
dip caused by fast load change from light to heavy is
diminished by a large bulk capacitor. OVP is easily
triggered at startup. Switching starting and stopping by
OVP at startup may cause audible noise and can increase
voltage stress at startup, which may be higher than normal
operation. This operation is improved when soft-start time
is very long. However, too-long startup time raises the
time needed for the output voltage to reach the rated
value, especially at light load. FAN7930 includes a startup
without overshoot feature. During startup, the feedback
loop is controlled by an internal proportional gain
controller and, when the output voltage reaches the
vicinity of the rated value, changed to the external
compensator after an internally fixed transition time
described in the Figure 6. In short, an internal
proportional gain controller prevents overshoot at startup;
external conventional compensator takes over after startup.
VOUT
Conventional controller
Startup overshoot
Overshoot protection startup
control
Figure 5 shows the minimum switching frequency, which
occurs at the peak of line voltage as a function of the
RMS line voltage for three output voltage settings. It is
interesting that, depending on where the output voltage is
set, the minimum switching frequency may occur at the
minimum or at the maximum line voltage. When the
output voltage is approximately 405V, the minimum
switching frequency is the same for both low line (85VAC)
and high line (265VAC).
Control transition
VCOMP
Depend on load
Internal controller
t
Figure 6. Startup Without Overshoot
Figure 5. Minimum Switching Frequency vs. RMS
Line Voltage (L = 390µH, POUT = 200W)
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
FAN7930 eliminates AC input voltage detection to save
the power loss caused by an input-voltage-sensing resistor
array and to optimize THD. Therefore, no information
about input voltage is available at the internal controller.
In many cases, the VCC of PFC controller is supplied by
an independent power source, like standby power, so
when the electric power is suddenly interrupted during
one or two AC line periods, VCC is still alive during that
time and PFC output voltage drops. Accordingly, the
control loop tries to compensate output voltage drop and
control voltage reaches its maximum. When AC line input
voltage is live, control voltage allows high switching
current and creates stress on the MOSFET and diode. To
protect against this, FAN7930 checks if the input AC
voltage exists. Once controller verifies that the input
voltage does not exist, soft-start is reset and waits until
AC input voltage is applied again. Soft-start manages the
turn-on time for smooth operation after detecting that the
AC voltage is live and results in less voltage and current
stress during startup.
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AN-8035
Figure 7. AC-Off Operation without AC-Absent
Detection Circuit
Figure 8. AC-Off Operation with AC-Absent
Detection Circuit
4. Design Considerations
In this section, a design procedure is presented using the
schematic in Figure 9 as a reference. A 200W PFC
application with universal input range is selected as a
design example. The design specifications are:
ƒ Line Voltage Range: 90~265VAC (Universal Input),
50Hz
ƒ Nominal Output Voltage and Current: 400V/0.5A
ƒ Hold-up Time Requirement: Output Voltage Should
ƒ
ƒ
ƒ
ƒ
Not Drop Below 330V During One Line Cycle
Output Voltage Ripple: Less than 8VPP
Minimum Switching Frequency: Higher than 40kHz
Control Bandwidth: 5~15Hz
VCC supplied from auxiliary power supply.
(200W)
CCOMP.HF
Figure 9. Reference Circuit for Design Example of BCM Boost PFC
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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AN-8035
[STEP-1] Define System Specifications
ƒ Line Frequency Range (VLINE,MIN and VLINE,MAX)
ƒ Line Frequency (fLINE)
ƒ Output-Voltage (VOUT)
ƒ Output Load Current (IOUT)
ƒ Output Power (POUT =VOUT × IOUT)
ƒ Estimated Efficiency (η)
(Design Example) Input voltage range is universal input,
output load is 500mA, and estimated efficiency is selected
as 0.9.
VLINE ,MIN = 90VAC , VLINE ,MAX = 265VAC
fLINE = 50 Hz
VOUT = 400V , IOUT = 500 mA
To calculate the maximum input power, it is necessary to
estimate the power conversion efficiency. At universal
input range, efficiency is recommended at 0.9; 0.93~0.95
is recommended when input voltage is high.
η = 0. 9
When input voltage is set at the minimum, input current
becomes the maximum to deliver the same power
compared at high line. Maximum boost inductor current
can be detected at the minimum line voltage and at its
peak. Inductor current can be divided into two categories;
one is rising current when MOSFET is on and the other is
output diode current when MOSFET is off, as shown in
Figure 10.
I IN ,MAX =
I L,PK =
4 ⋅ POUT
η ⋅ 2 ⋅ VLINE ,MIN
I L,PK
I IN ,MAXRMS
=
4 ⋅ 400V ⋅ 0.5 A
0.9 ⋅ 2 ⋅ 90
= 6.984 A
6.984 A
= 3.492 A
2
I
3.492 A
= IN ,MAX =
= 2.469 A
2
2
2
=
Figure 10. Inductor and Input Current
Because switching frequency is much higher than line
frequency, input current can be assumed to be constant
during a switching period, as shown in Figure 11.
[STEP-2] Boost Inductor Design
The boost inductor value is determined by the output
power and the minimum switching frequency. The
minimum switching frequency must be higher than the
maximum audible frequency band of 20kHz. Minimum
frequency near 20kHz can decrease switching loss with
the cost of increased inductor size and line filter size. Toohigh minimum frequency may increase the switching loss
and make the system respond to noise. Selecting in the
range of about 30~60kHz is a common choice; 40~50kHz
is recommended with FAN7930.
Figure 11. Inductor and Input Current
With the estimated efficiency, Figure 10 and Figure 11
inductor current peak (IL,PK), maximum input current
(IIN,MAX), and input RMS (Root Mean Square) current
(IIN,MAXRMS) are given as:
I L , PK =
4 ⋅ POUT
η ⋅ 2 ⋅ V LINE , MIN
[ A]
(3)
L=
IIN ,MAX = I L,PK / 2 [ A ]
(4)
IIN ,MAXRMS = IIN ,MAX / 2 [ A ]
(5)
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
The minimum switching frequency may appear at
minimum input voltage or maximum input voltage,
depending on the output voltage level. When PFC output
voltage is less than 405V, minimum switching appears at
the maximum input voltage, according to Fairchild
application note AN-6086. The inductance is obtained
using the minimum switching frequency:
η⋅
( 2V )
4 ⋅ fSW ,MIN ⋅ POUT
2
LINE
⎛
2VLINE
⋅ ⎜1 +
⎜
V
OUT − 2VLINE
⎝
⎞
⎟
⎟
⎠
[H]
(6)
where L is boost inductance and fSW,MIN is the minimum
switching frequency.
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AN-8035
The maximum on time needed to carry peak inductor
current is calculated as:
tON,MAX = L ⋅
IL,PK
2 ⋅ VLINE,MIN
[s]
(7)
Once inductance and the maximum inductor current are
calculated, the number of turns of the boost inductor
should be determined considering the core saturation. The
minimum number of turns is given as:
N BOOST ≥
I L ,PK ⋅ L [ μH ]
Ae [ mm 2 ] ⋅ ΔB
Figure 13. Ae and AW
[ Turns ]
(8)
where Ae is the cross-sectional area of core and ΔB is the
maximum flux swing of the core in Tesla. ΔB should be
set below the saturation flux density.
(Design Example) Since the output voltage is 400V, the
minimum frequency occurs at high-line (265VAC) and
full-load condition. Assuming the efficiency is 90% and
selecting the minimum frequency as 50kHz, the inductor
value is obtained as:
Figure 12 shows the typical B-H characteristics of ferrite
core from TDK (PC45). Since the saturation flux density
(ΔB) decreases as the temperature increases, the high
temperature characteristics should be considered.
L=
RMS inductor current (IL,RMS) and current density of the
coil (IL,DENSITY) can be given as:
=
(
η ⋅ 2VLINE
)
2
⎛
2VLINE
4 ⋅ fSW ,MIN ⋅ POUT ⋅ ⎜1 +
⎜ V
OUT − 2VLINE
⎝
0.9 ⋅
( 2 × 265)
⎞
⎟
⎟
⎠
2
⎛
2 ⋅ 265 ⎞⎟
4 ⋅ 50 ⋅10 ⋅ 200 ⋅ ⎜1 +
⎜ 400 − 2 ⋅ 265 ⎟
⎝
⎠
= 199.4 [ μH ]
3
I L,RMS =
I L ,DENSITY =
I L,PK
6
(9)
[ A]
I L ,RMS
2
π ⋅ ⎛⎜ d wire 2 ⎞⎟ ⋅ Nwire
⎝
⎠
[ A / mm 2 ]
Assuming EER3019N core (PL-7, Ae=137mm2) is used
and setting ΔB as 0.3T, the primary winding should be:
(10)
N BOOST ≥
I L ,PK ⋅ L [ μH ]
Ae [ mm ] ⋅ ΔB
2
=
6.984 ⋅ 209
= 34 [ T ]
137 ⋅ 0.3
where dWIRE is the diameter of winding wire and NWIRE is
the number of strands of winding wire.
The number of turns (NBOOST) of the boost inductor is
determined as 34 turns.
When selecting wire diameter and strands; current
density, window area (AW, refer to Figure 13) of selected
core, and fill factor need to be considered. Winding
sequence of the boost inductor is relatively simple
compared to a DC-DC converter, so fill factor can be
assumed about 0.2~0.3.
When 0.10mm diameter and 50-strand wire is used, RMS
current of inductor coil and current density are:
I
6.984
I L,RMS = L,PK =
= 2.85 [ A ]
6
6
Layers cause the skin effect and proximity effect in the
coil, so real current density may be higher than expected.
IL,DENSITY=
IL,RMS
2.85
=
= 7.3 [ A / mm2 ]
2
π ⋅ (0.1 / 2)2 ⋅ 50
dwire ⎞
⎛
π ⋅⎜
⋅ Nwire
2 ⎟⎠
⎝
Figure 12. Typical B-H Curves of Ferrite Core
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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AN-8035
[STEP-3] Inductor Auxiliary Winding Design
Figure 14 shows the application circuit of nearby ZCD pin
from auxiliary winding.
Auxiliary winding must give enough energy to trigger
ZCD threshold to detect zero current. Minimum auxiliary
winding turns are given as:
NAUX ≥
1.5V ⋅ NBOOST
VOUT − 2VLINE,MAX
[ Turns]
(11)
where 1.5V is the positive threshold of the ZCD pin.
To guarantee stable operation, auxiliary winding turns are
recommended to add 2~3 turns to the calculation result of
Equation (11). However, too many auxiliary winding
turns raise the negative clamping loss at high line and
positive clamping loss at low line.
(Design Example) 34 turns are selected as boost inductor
turns and auxiliary winding turns are calculated as:
NAUX ≥
Figure 14. Application Circuit of ZCD Pin
The first role of ZCD winding is detecting the zerocurrent point of the boost inductor. Once the boost
inductor current becomes zero, the effective capacitor
shown at the MOSFET drain pin (Ceff) and the boost
inductor resonate together. To minimize the constant turnon time deterioration and turn-on loss, the gate is turned
on again when the drain source voltage of the MOSFET
(VDS) reaches the valley point shown in Figure 15. When
input voltage is lower than half of the boosted output
voltage, zero voltage switching (ZVS) is possible if
MOSFET turn-on is triggered at valley point.
1.5V ⋅ NBOOST
VOUT − 2VLINE,MAX
=
1.5 ⋅ 34
400 − 2 ⋅ 265
= 2.02[ Turns]
Choice should be around 4~5 turns after adding 2~3 turns.
[STEP-4] ZCD Circuit Design
If a transition time when VAUXILIARY drops from 1.4V to
0V is ignored from Figure 15, the needed additional delay
by the external resistor and capacitor is one quarter of the
resonant period. The time constant made by ZCD resistor
and capacitor should be the same as one quarter of the
resonant period:
2π Ceff ⋅ L
(12)
RZCD ⋅ CZCD =
4
where Ceff is the effective capacitor shown at the
MOSFET drain pin; CZCD is the external capacitance at
the ZCD pin; and RZCD is the external resistance at the
ZCD pin.
The second role of RZCD is the current limit of the internal
negative clamp circuit when auxiliary voltage drops to
negative due to MOSFET turn on. ZCD voltage is
clamped 0.65V and minimum RZCD can be given as:
RZCD
⎛ N AUX
⎞
⎜
2VLINE,MAX − 0.65V ⎟⎟
⎜N
BOOST
⎠ [Ω]
≥⎝
3mA
(13)
where 3mA is the clamping capability of the ZCD pin.
Figure 15. ZCD Detection Waveforms
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
The calculation result of Equation (13) is normally higher
than 15kΩ. If 20kΩ is assumed as RZCD, calculated CZCD
from Equation (12) is around 10pF when the other
components are assumed as conventional values used in
the field. Because most IC pins have several pF parasitic
capacitance, CZCD can be eliminated when RZCD is higher
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AN-8035
than 30kΩ. However, a small capacitor would be helpful
when auxiliary winding suffers from operating noise.
The PFC control loop has two conflicting goals: output
voltage regulation and making the input current shape the
same as input voltage. If the control loop reacts to regulate
output voltage smoothly, as shown in Figure 16, control
voltage varies widely with the input voltage variation.
Input current acts to the control loop and sinusoidal input
current shape can not be attained. This is the reason
control response of most PFC topologies is very slow and
turn-on time over AC period is kept constant. This is also
the reason output voltage ripple is made by input and
output power relationship, not by control-loop
performance.
Figure 17. Inductor Current at AC Voltage Peak
Figure 16. Input Current Deterioration by Fast Control
If on-time is controlled constantly over one AC period,
inductor current peak follows AC input voltage shape and
achieves good power factor. Off-time is basically inductor
current reset time due to the boundary mode and is
determined by the input and output voltage difference.
When input voltage is at its peak, the voltage difference
between input and output voltage is small, and long turnoff time is necessary. When input voltage is near zero,
turn-off time is short, as shown in Figure 17 and Figure
18. Though inductor current drops to zero, there is a
minor delay, explained above. The delay can be assumed
as fixed when AC is at line peak and zero. Near AC line
peak, the inductor current decreasing slope is slow and
inductor current slope is also slow during the ZCD delay.
The amount of negative current is not much higher than
the inductor current peak. Near the AC line zero, inductor
current decreasing slope is very high and the amount of
negative current is higher than positive inductor current
peak because input voltage is almost zero.
Figure 18. Inductor Current at AC Voltage Zero
Negative inductor current creates zero current distortion
and degrades the power factor. Improve this by extending
turn-on time at the AC line input near the zero cross.
Negative auxiliary winding voltage, when MOSFET is
turned on, is linearly proportional to the input voltage.
Sourcing current generated by the internal negative
clamping circuit is also proportional to sinusoidal input
voltage. That current is detected internally and added to
the internal sawtooth generator, as shown in Figure 19.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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AN-8035
Figure 19. ZCD Current and Sawtooth Generator
When AC input voltage is almost zero, no negative
current is generated from inside, but sourcing current
when input voltage is high is used to raise the sawtooth
generator slope and turn-on time is shorter. As a result,
turn-on time when AC voltage is zero is longer compared
to AC voltage, in peaks shown in Figure 20.
Figure 20. THD Improvement
Figure 22. Internal Sawtooth Wave Slope Variation
RZCD also influences control range. Because FAN7930
doesn’t detect input voltage, voltage-mode control value
is determined by the turn-on time to deliver needed
current to boost output voltage. When input voltage
increases, control voltage decreases rapidly. For example,
if input voltage doubles, control voltage drops to one
quarter. Making control voltage maximum when input
voltage is low and at full load is necessary to use the
whole control range for the rest of the input voltage
conditions. Matching maximum turn-on time needed at
low line is calculated in Equation (7) and turn-on time
adjustment by RZCD guarantees use of the full control
range. RZCD for control range optimization is obtained as:
The current that comes from the ZCD pin, when auxiliary
voltage is negative, depends on RZCD. The second role of
RZCD is also related with the improving the Total
Harmonic Distortion (THD).
The third role of RZCD is making the maximum turn-on
time adjustment. Depending on sourcing current from the
ZCD pin, the maximum on-time varies as in Figure 21.
R ZCD ≥
2 ⋅ VLINE ,MIN ⋅ N AUX
28 μs
⋅
tON ,MAX 1 − tON ,MAX
0.469 mA ⋅ N BOOST (14)
where:
tON,MAX is calculated by Equation (7);
tON,MAX1 is maximum on-time programming 1;
NBOOST is the winding turns of boost inductor; and
NAUX is the auxiliary winding turns.
RZCD calculated by Equation (13) is normally lower than
the value calculated in Equation (14). To guarantee the
needed turn on-time for the boost inductor to deliver rated
power, the resulting RZCD from Equation (13) is normally
not suitable. RZCD should be higher than the result of
Equation (14) when output voltage drops as a result of
low line voltage.
Figure 21. Maximum On-Time Variation vs. IZCD
With the aid of IZCD, an internal sawtooth generator slope
is changed and turn-on time varies as shown in Figure 22.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
When input voltage is high and load is light, not much
input current is needed and control voltage of VCOMP
touches switching stop level, such as if FAN7930 is 1V.
However, in some applications, a PFC block is needed to
operate normally at light load. To compensate control
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AN-8035
range correctly, input voltage sensing is necessary, such
as with Fairchild’s interleaved PFC controller FAN9612,
or special care on sawtooth generator is necessary.
Without it, optimizing RZCD is only slightly helpful for
control range. This is explained and depicted in the
associated Excel® design tool “COMP Range” worksheet.
To guarantee enough control range at high line, clamping
output voltage lower than rated output on the minimum
input condition can help.
COUT ≥
2 ⋅ POUT ⋅ tHOLD
(VOUT − 0.5 ⋅ ΔVOUT,RIPPLE)2 −VOUT,MIN2
[f ]
(16)
where tHOLD is the required hold-up time and VOUT,MIN is
the minimum output voltage during hold-up time.
(Design Example) Minimum RZCD for clamping
capability is calculated as:
⎛ NAUX
⎞
⎜
2VLINE,MAX − 0.65V ⎟⎟
⎜N
BOOST
⎝
⎠
RZCD ≥
3mA
⎛5
⎞
2 ⋅ 265 − 0.65V ⎟
⎜
⎝ 34
⎠
= 18.2kΩ
=
3mA
Minimum RZCD for control range is calculated as:
RZCD ≥
=
2 ⋅ VLINE ,MIN ⋅ N AUX
28 μs
⋅
tON ,MAX1 − tON ,MAX 0.469 mA ⋅ NBOOST
28 μs
2 ⋅ 90 ⋅ 5
⋅
= 37.2 kΩ
42 μs − 10.9μs 0.469 mA ⋅ 34
A choice close to the value calculated by the control
range is recommended. 39kΩ is chosen in this case.
Figure 23. Output Voltage Ripple
The voltage rating of capacitor can be obtained as:
VST ,COUT =
VOVP ,MAX
⋅VOUT [ V ]
VREF
(17)
where VOVP,MAX and VREF are the maximum tolerance
specifications of over-voltage protection triggering
voltage and reference voltage at error amplifier.
[STEP-5] Output Capacitor Selection
(Design Example) With the ripple specification of 8Vp-p,
the capacitor should be:
The output voltage ripple should be considered when
selecting the output capacitor. Figure 23 shows the line
frequency ripple on the output voltage. With a given
specification of output ripple, the condition for the output
capacitor is obtained as:
COUT
IOUT
≥
[F]
2π ⋅ fLINE ⋅ ΔVOUT,RIPPLE
CO ≥
IOUT
0.5
=
= 198.9[ μF ]
2π ⋅ fLINE ⋅ ΔVOUT,ripple 2π ⋅ 50 ⋅ 8
Since minimum allowable output voltage during one cycle
line (20ms) drop-outs is 330V, the capacitor should be:
POUT ⋅ t HOLD
CO ≥
(VOUT − 0.5 ⋅ ΔVOUT,ripple)2 −VOUT,MIN2
(15)
where VOUT,RIPPLE is the peak-to-peak output voltage
ripple specification.
=
2 ⋅ 200 ⋅ 20 ×10−3
(400 − 0.5 ⋅ 8)2 − 3302
= 167[ μF ]
The output voltage ripple caused by ESR of electrolytic
capacitor is not as serious as other power converters
because output voltage is high and load current is small.
Since too much ripple on the output voltage may cause
premature OVP during normal operation, the peak-to-peak
ripple specification should be smaller than 15% of the
nominal output voltage.
To meet both conditions, the output capacitor must be larger
than 199μF. A 220μF capacitor is selected for the output
capacitor.
The hold-up time should also be considered when
determining the output capacitor as:
VST ,COUT =
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
The voltage stress of selected capacitor is calculated as:
VOVP,MAX
2.730
⋅ 400 = 436.8[ V ]
⋅ VOUT =
2.500
VREF
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AN-8035
The precise turn-off loss calculation is difficult because of
the nonlinear characteristics of MOSFET turn off. When
piecewise linear current and voltage of MOSFET during
turn-off and inductive load are assumed, MOSFET turnoff loss is obtained as:
PQ,SWOFF =
1
⋅ VOUT ⋅ I L ⋅ tOFF ⋅ fSW [ W ]
2
(21)
where tOFF is the turn-off time and fSW is the switching
frequency.
Boundary mode PFC inductor current and switching
frequency vary at every switching moment. RMS inductor
current and average switching frequency over one AC
period can be used instead of instantaneous values.
[STEP-6] MOSFET and DIODE Selection
Selecting the MOSFET and diode needs extensive
knowledge and calculation regarding loss mechanisms
and gets more complicated if proper selection of a
heatsink is added. Sometimes the loss calculation itself is
based on assumptions that may be far from reality. Refer
to industry resources regarding these topics. This note
shows the voltage rating and switching loss calculations
based on the linear approximation.
The voltage stress of the MOSFET is obtained as:
VST ,Q =
VOVP ,MAX
⋅ VOUT + VDROP,DOUT [ V ]
VREF
(18)
where VDROP,DOUT is the maximum forward-voltage drop
of output diode.
After the MOSFET is turned off, the output diode turns on
and a large output electrolytic capacitor is shown at the
drain pin, thus a drain voltage clamping circuit that is
necessary on other topologies is not necessary in PFC.
During the turn-off transient, boost inductor current
changes the path from MOSFET to output diode and
before the output diode turns on; a minor voltage peak can
be shown at drain pin, which is proportional to MOSFET
turn-off speed.
MOSFET loss can be divided into three parts: conduction
loss, turn-off loss, and discharge loss. Boundary mode
guarantees ZCS (Zero Current Switching) of MOSFET
when turned on, so turn-on loss is negligible.
The MOSFET RMS current and conduction loss are
obtained as:
IQ,RMS = IL,PK ⋅
1 4 2 ⋅VLINE
[ A]
−
6 9π ⋅VOUT
PQ,CON = (IQ ,RMS )2 ⋅ RDS,ON [ W ]
Individual loss portions are changed according to the
input voltage; maximum conduction loss appears at low
line because of high input current; and maximum
switching off loss appears at high line because of the high
switching frequency. Thus, resulting loss is always lower
than the summation of the two losses calculated above.
Capacitive discharge loss made by effective capacitance
shown at drain and source, which includes MOSFET
COSS, an externally added capacitor to reduce dv/dt and
parasitic capacitors shown at drain pin, is also dissipated
at MOSFET. That loss is calculated as:
PQ,DISCHG=
1
2
(COSS + CEXT + CPAR) ⋅VOUT
⋅ fSW [ W ]
2
where:
COSS is the output capacitance of MOSFET;
CEXT is an externally added capacitor at drain and source
of MOSFET; and
CPAR is the parasitic capacitance shown at drain pin.
Because the COSS is a function of the drain and source
voltage, it is necessary to refer to graph data showing the
relationship between COSS and voltage.
Estimate the total power dissipation of MOSFET as the
sum of three losses:
PQ = PQ,CON + PQ,SWOFF + PQ,DISCHG [ W ]
(23)
Diode voltage stress is the same as the output capacitor
stress calculated in Equation (17).
The average diode current and power loss are obtained as:
I DOUT,AVE =
(19)
IOUT
η
[ A]
(24)
PDOUT = VDROP,DOUT ⋅ I DOUT,AVE [ W ]
(20)
where IQ,RMS is the RMS value of MOSFET current,
PQ,CON is the conduction loss caused by MOSFET current,
and RDS,ON is the ON resistance of the MOSFET.
(22)
(25)
where VDROP,DOUT is the forward voltage drop of diode.
ON resistance is described as “static ON resistance” and
varies depending on junction temperature. That variation
information is normally supplied as a graph in the
datasheet and may vary by manufacturer. When
calculating conduction loss, generally multiply three with
the RDS,ON for more accurate estimation.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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(Design Example) Internal reference at the feedback pin
is 2.5V and maximum tolerance of OVP trigger voltage
is 2.730V. If Fairchild’s FCPF20N60 MOSFET and
FFPF08H60S diode are selected, VD,FOR is 2.1V at 8A,
25oC, maximum RDS,ON is 0.19Ω at drain current is 10A,
and maximum COSS is 85pF at drain-source voltage is
480V.
VST,Q =
=
(Design Example) Maximum inductor current is 6.984A
and sensing resistor is calculated as:
RCS =
⎛
1 4 2 ⋅VLINE
= ⎜I L,PK ⋅
−
⎜⎜
6 9π ⋅VOUT
⎝
PQ,SWOFF =
(27)
Power rating of the sensing resistor is recommended a
twice the power rating calculated in Equation (27).
VOVP,MAX
⋅VOUT + VDROP,DIODE
VREF
⎛
1 4 2 ⋅ 90
= ⎜6.984 ⋅
−
⎜
6 9π ⋅ 400
⎝
2
⎞
⎟ ⋅ (R
DS,ON )
⎟⎟
⎠
VCS,LIM
pk
I ind
⋅1.1
=
0.8
= 0.104[ Ω ]
6.984 ⋅1.1
Choosing 0.1Ω as RCS, power loss is calculated as:
PRCS,LOSS = IQ2 ,RMS ⋅ RCS = 2.4362 ⋅ 0.1 = 0.59[ W ]
2
⎞
⎟ ⋅ (0.19 × 3) = 3.38[ W ]
⎟
⎠
Recommended power rating of sensing resistor is 1.19W.
1
⋅ VOUT ⋅ IL ⋅ tOFF ⋅ fSW
2
1
⋅ 400 ⋅ 2.469⋅ 50ns ⋅ 50k / 2 =1.54 [ W ]
2
PQ,DISCHG =
=
PRCS = IQ2 ,RMS ⋅ RCS [ W ]
2.73
⋅ 400 + 2.1 = 438.9 [V ]
2.50
PQ,CON
=
Once resistance is calculated, its power loss at low line is
calculated as:
1
2
⋅ (COSS + CEXT + CPAR ) ⋅VOUT
⋅ fSW
2
[STEP-9] Design Compensation Network
1
⋅ 85p ⋅ 4002 ⋅ 50k / 2 = 0.43 [ W ]
2
Diode average current and forward-voltage drop loss as:
I DOUT,AVE =
IOUT
η
=
0.5
= 0.56[ A ]
0.9
The boost PFC power stage can be modeled as shown in
Figure 24. MOSFET and diode can be changed to lossfree resistor model and then be modeled as a voltagecontrolled current source supplying RC network.
PDOUT,LOSS = VDOUT,FOR ⋅ I DOUT,AVE = 2.1 ⋅ 0.56 = 1.46[ W ]
[STEP-8] Determine Current-Sense Resistor
It is typical to set pulse-by-pulse current limit level a little
higher than the maximum inductor current calculated by
Equation (3). For 10% margin, the current-sensing resistor
is selected as:
RCS =
VCS,LIM
I L,PK ⋅1.1
[Ω]
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
Figure 24. Small Signal Modeling of the Power Stage
(26)
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12
AN-8035
By averaging the diode current during the half line cycle,
the low-frequency behavior of the voltage controlled
current source of Figure 24 is obtained as:
I DOUT,AVE = KSAW ⋅
2VLINE 2VLINE
⋅
[ A]
4VOUT
L
(28)
Proportional and integration (PI) control with highfrequency pole is typically used for compensation, as
shown in Figure 27. The compensation zero (fCZ)
introduces phase boost, while the high-frequency
compensation pole (fCP) attenuates the switching ripple.
The transfer function of the compensation network is
obtained as:
where:
L is the boost inductance,
VOUT is the output voltage; and
KSAW is the internal gain of sawtooth generator
(that of FAN7930 is 8.496×10-6).
∧
v COMP
∧
v OUT
Then the low-frequency, small-signal, control-to-output
transfer function is obtained as:
∧
v OUT
= KSAW ⋅
∧
v COMP
where f p =
(VLINE )2 RL ⋅
4VOUT ⋅ L
fI =
1
1+
s
2π f p
(29)
2
and RL is the output load
2π ⋅ RLCOUT
resistance in a given load condition.
Figure 25 and Figure 26 show the variation of the controlto-output transfer function for different input voltages and
different loads. Since DC gain and crossover frequency
increase as input voltage increases, and DC gain increases
as load decreases, high input voltage and light load is the
worst condition for feedback loop design.
where fCZ
2π fI
=
⋅
s
s
2π fCZ
s
1+
2π fCP
1+
2.5
115μmho
⋅
VOUT 2π ⋅ CCOMP, LF + CCOMP, HF
(
)
(30)
1
=
2π ⋅ RCOMP ⋅ CCOMP, LF
1
⎛ CCOMP, LF ⋅ CCOMP, HF ⎞
⎟
2π ⋅ RCOMP ⋅ ⎜
⎜ CCOMP, LF + CCOMP, HF ⎟
⎠
⎝
If CCOMP,LF is much larger than CCOMP,HF, fI and fCP can be
simplified as:
fCP =
fI ≅
2.5
115μmho
[ Hz]
⋅
VOUT 2π ⋅ CCOMP, LF
fCP ≅
(31)
1
[ Hz]
2π ⋅ RCOMP ⋅ CCOMP, HF
GM = 115 μ mho
Figure 25. Control-to-Output Transfer Function for
Different Input Voltages
Figure 27. Compensation Network
Figure 26. Control-to-Output Transfer Function for
Different Loads
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
The feedback resistor is chosen to scale down the output
voltage to meet the internal reference voltage:
RFB1
⋅ VOUT = 2.5V
RFB1 + RFB2
(32)
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(Design Example) IfRFB1 is 13MΩ, then RFB2 is:
Typically, high RFB1 is used to reduce power consumption
and, at the same time, CFB can be added to raise the noise
immunity. The maximum CFB currently used is several
nano farads. Adding a capacitor at the feedback loop
introduces a pole as:
fFP =
≅
1
2π ⋅ (RFB1 // RFB2 ) ⋅ CFB
1
2π ⋅ RFB2 ⋅ CFB
RFB2 =
2.5V
2.5
RFB1 =
13 ×106 = 81.7 kΩ
VOUT − 2.5V
400 − 2.5
Choosing the crossover frequency (control bandwidth) at
15Hz, CCOMP,LF is obtained as:
(33)
CCOMP, LF ≅
[ Hz]
KSAW (VLINE )2 2.5 ⋅ 115μ mho
2 ⋅ VOUT2 ⋅ L ⋅ COUT (2π fC )2
8.496 ×10−6 (230)2 2.5 ⋅ 115 ×10−6
where (RFB1 // RFB 2 ) = RFB1 ⋅ RFB 2
=
Though RFB1 is high, pole frequency made by the
synthesized total resistance and several nano farads is
several kilo hertz and rarely affects control-loop response.
Actual CCOMP,LF is determined as 1000nF since it is the
closest value among the off-the-shelf capacitors. RCOMP is
obtained as:
R FB1 + RFB 2
2 ⋅ 4002 ⋅ 199 ×10−6 ⋅ 220 ×10−6 (2π 15)2
The procedure to design the feedback loop is:
a.
RCOMP =
Determine the crossover frequency (fC) around
1/10~1/5 of line frequency. Since the control-tooutput transfer function of the power stage has
-20dB/dec slope and -90o phase at the crossover
frequency, as shown in Figure 28; it is required
to place the zero of the compensation network
(fCZ) around the crossover frequency so 45°
phase margin is obtained. The capacitor CCOMP,LF
is determined as:
CCOMP, LF ≅
KSAW (VLINE )2 2.5 ⋅115μ mho
2 ⋅VOUT2 ⋅ L ⋅ COUT (2π fC )2
[f ]
= 1038nF
1
1
=
= 10.22kΩ
2π ⋅ fC ⋅ CCOMP,LF 2π ⋅ 15 ⋅ 1038 ×10 −9
Selecting the high-frequency pole as 150Hz, CCOMP,HF is
obtained as:
CCOMP,HF =
1
1
=
= 103nF
2π ⋅ fCP ⋅ RCOMP 2π ⋅ 150 ⋅ 10.22 × 103
These components result in a control loop with a bandwidth
of 19.7Hz and phase margin of 46°. The actual bandwidth is
a little larger than the asymptotic design.
(34)
To place the compensation zero at the crossover
frequency, the compensation resistor is obtained as:
RCOMP =
b.
1
[Ω]
2π ⋅ fC ⋅ CCOMP,LF
(35)
Place this compensator high-frequency pole (fCP)
at least a decade higher than fC to ensure that it
does not interfere with the phase margin of the
voltage regulation loop at its crossover
frequency. It should also be sufficiently lower
than the switching frequency of the converter for
noise to be effectively attenuated. The capacitor
CCOMP,HF is determined as:
CCOMP,HF =
1
[Ω]
2π ⋅ fCP ⋅ RCOMP
(36)
Figure 28. Compensation Network Design
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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[STEP-10] Line Filter Capacitor Selection
It is typical to use small bypass capacitors across the
bridge rectifier output stage to filter the switching current
ripple, as shown in Figure 29. Since the impedance of the
line filter inductor at line frequency is negligible
compared to the impedance of the capacitors, the line
frequency behavior of the line filter stage can be modeled,
as shown in Figure 29. Even though the bypass capacitors
absorb switching ripple current, they also generate
circulating capacitor current, which leads the line voltage
by 90o, as shown in Figure 30. The circulating current
through the capacitor is added to the load current and
generates displacement between line voltage and current.
The displacement angle is given by:
⎛ η ⋅ (VLINE )2 ⋅ 2π ⋅ fLINE ⋅ CEQ ⎞
⎟
(37)
⎜
⎟
POUT
⎝
⎠
where CEQ is the equivalent capacitance that appears
across the AC line (CEQ=CF1+CF2+CHF).
θ = tan−1 ⎜
Figure 29. Equivalent Circuit of Line Filter Stage
The resultant displacement factor is:
DF = cos(θ )
(38)
Since the displacement factor is related to power factor,
the capacitors in the line filter stage should be selected
carefully. With a given minimum displacement factor
(DFMIN) at full-load condition, the allowable effective
input capacitance is obtained as:
CEA <
POUT
η ⋅ (VLINE )2 ⋅ 2π ⋅ fLINE
(
)
⋅ tan cos−1 (DFMN ) [ F ]
(39)
One way to determine if the input capacitor is too high or
PFC control routine has problems is to check power factor
(PF) and Total Harmonic Distortion (THD). PF is the
degree to which input energy is effectively transferred to
the load by the multiplication of displacement factor and
THD that is input current shape deterioration ratio. PFC
control loop rarely has no relation to displacement factor
and input capacitor rarely has no impact on the input
current shape. If PF is low (high is preferable), but THD
is quite good (low is preferable), it can be concluded that
input capacitance is too high and PFC controller is fine.
θ
Figure 30. Line Current Displacement
(Design Example) Assuming the minimum
displacement factor at full load is 0.98, the equivalent
input capacitance is obtained as:
POUT
⋅ tan cos−1 (DFMN )
CEA <
2
η ⋅ (VLINE) ⋅ 2π ⋅ fLINE
200
<
⋅ tan cos−1 (0.98) = 2.0453μF
2
0.9 ⋅ (264) ⋅ 2π ⋅ 50
(
(
)
)
Thus, the sum of the capacitors on the input side should
be smaller than 2.0µF.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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AN-8035
Appendix 1: Use of the RDY Pin
Typically, boosted output voltage from the PFC block is
used as input voltage to the DC-DC conversion block. For
some types of DC-DC converter, it is recommended to
trigger operation after the input voltage raised to some
level. For example, LLC resonant converter or forward
converter’s input voltage is limited to some range to
enhance performance or guarantee the stable operation.
For that purpose, the PFC RDY pin is assigned and can be
used as a acknowledge signal for the DC-DC conversion
stages. When PFC output voltage rises higher than the
internal threshold, PFC RDY output is pulled HIGH by
the external pull-up voltage and drops to zero with
hysteresis.
2.240V
VOUT [V ]
2.500V
1.640V
VOUT,RDYL =
VOUT [V ]
2.500V
VOUT,RDYH =
supply, that breakdown current flows high for a long time.
In this case, the internal MOSFET may be damaged since
the external small-signal bipolar junction transistor current
capability is higher than the internal RDY MOSFET.
Once circuit configuration is settled, voltage after
subtracting forward-voltage drop of the diode and voltage
drop (by the multiplication of base current and RPULLUP)
from the VCC of FAN7930 is available for the LLC
controller’s VCC source.
Another example is using RDY when the secondary side
needs PFC voltage information. When a Cold Cathode
Fluorescent Lamp (CCFL) is used for the backlight source
of an LCD TV, the inverter stage to ignite CCFL can
receive PFC output voltage directly. For that application,
Figure 32 can be a suitable circuit configuration.
(40)
where VOUT,RDYH is the VOUT voltage to trigger PFC RDY
output to pull HIGH and VOUT,RDYL is the VOUT voltage to
trigger PFC ready output to drop to zero.
If rated VOUT is 400VDC, then VOUT,RDYH is 358VDC, and
VOUT,RDYL is 262VDC.
When LLC resonant converter is assumed to connect at
the PFC output, the RDY pin can control the VCC for the
LLC controller, as shown in Figure 31.
Voltage source
from standby block
Figure 32. RDY Application Circuit Using Opto-Coupler
VCC
8
RPULLUP
RDY
PN2222A
2
1N4148
VCC for LLC
controller
-
2.24V/1.64V
With this application circuit, the minimum RPULLUP is
given by Equation (42) and the maximum RPULLUP is
limited by sufficient current to guarantee stable operation
of the opto-coupler. Assuming 1mA is the typical quantity
to drive opto-coupler, the maximum RPULLUP is:
RPULLUP ≤
+
INV
VPULLUP − VOPTO,F
1mA
[Ω]
(42)
where VOPTO,R is the input forward-voltage drop of the
opto-coupler.
1
Figure 31. RDY Application Circuit for VCC Driving
RPULLUP is chosen based on the current capability of
internal open-drain MOSFET and can be obtained as:
RPULLUP ≥
VPULLUP − VRDY,SAT
I RDY,SK
[Ω]
(41)
It may possible that a secondary microcontroller has
authority to give a trigger signal to the CCFL inverter
controller; however, after combining the microcontroller
signal and RDY signal from the primary-side, the inverter
stage is triggered only when the two signals meet the
requirements at the same time.
where VPULLUP is the pull-up voltage, VRDY,SAT is the
saturation voltage of the internal MOSFET, and IRDY,SK is
the allowable sink current for the internal MOSFET.
A fast diode, such as 1N4148, is needed to prohibit the
emitter-base breakdown. Without that diode, when RDY
voltage drops to VRDY,SAT after being pulled up, emitter
voltage maintains operating voltage for LLC controller
and almost all the voltage is applied to the emitter and
base. Breakdown current flows from emitter, base, and
drain of the MOSFET to the source of MOSFET. Because
a large electrolytic capacitor is typically used at the VCC
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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AN-8035
Appendix 2: Gate Driver Design
FAN7930 directly drives the gate of the MOSFET and
various combinations of gate driver circuits are possible.
Figure 33Figure 31 shows the three circuits that are
widely used.
When only one resistor is used, the turn-on and turn-off
paths follow the same routine and turn-on and turn-off
speed cannot be changed simultaneously. To cover this,
make different paths by two resistors and diode if
possible. Turn-off current flows through the diode first,
instead of RON, and then RON and ROFF show together.
Accordingly, faster turn off is possible. However, turn-off
path using internal gate driver’s sinking path and current
is limited by sinking current capability. If a PNP transistor
is added between the gate and source of the MOSFET, the
gate is shorted to source locally without sharing the
current path to the gate driver. This makes the gate
discharge to the much smaller path than that made by the
controller. The possibility of ground bounce is reduced
and power dissipation in the gate driver is reduced. Due to
new high-speed MOSFET types such as SupreMOS® or
SuperFET™, gate speed is getting fast. This decreases the
switching loss of the MOSFET. At the same time, power
systems suffer from the EMI deterioration or noise
problems, like gate oscillation. Therefore, sometimes a
gate discharge circuit is inevitable to use high-speed
characteristics fully.
The most difficult and uncertain task in direct gate drive is
optimizing circuit layout. Gate driving path from the OUT
pin, resistor, MOSFET gate, and MOSFET source to the
GND pin should be as short as possible to reduce parasitic
inductance; which may make MOSFET on/off speed slow
or introduce unwanted gate oscillation. Using a wider
PCB pattern for this lane reduces parasitic inductance. To
damp unwanted gate oscillation made by the capacitance
at the gate pin and parasitic inductance formed by
MOSFET internal bonding wire and PCB pattern, proper
resistance can match the impedance at the resonant
frequency. To meet EMI regulations or for the redundant
system, fast gate speed can be sacrificed by increasing
serial resistance between the gate driver and gate.
An optimal gate driver circuit needs intensive knowledge
of MOSFET turn-on/off characteristics and consideration
of the other critical performance requirements of the
system. This is beyond the scope of this paper and many
reference papers can be found in the industry literature.
Figure 33. Equivalent Circuit of Line Filter Stage
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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17
AN-8035
Appendix 3: Design Summary
Figure 34 shows the schematic of the 200W BCM boost PFC design example used in this application note. EER3019N
core is used for the boost inductor.
Figure 34. Schematic of Design Example
Figure 35. Boost Inductor Specification
Table 1.
Winding Specification
NP
Pin
Diameter / Thickness
Turns
3,4 Æ 1,2
0.1φ × 50 (Litz wire)
0.05mm
34
Insulation Tape
NAUX
9,10 Æ 6,7
Insulation Tape
0.3φ
0.05mm
3
5
3
Core: EER3019N (Ae=137 mm2)
Bobbin: EER3019N
Inductance: 210μH
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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18
AN-8035
Appendix 4. Experimental Verification
To show the validity of the design procedure presented in
this application note, the converter of the design example
was built and tested. All the circuit components are
exactly as designed in the example.
Figure 36 and Figure 37 show the inductor current and
input current for 115VAC and 230VAC condition. Figure 38
and Figure 39 show the output response for 115VAC fullload and no-load conditions. Regardless of load
conditions, output voltage shows no overshoot. Figure 39
shows response when AC input voltage was omitted 20ms
periodically. As observed, control voltage was clamped
when AC absent was detected internally and restarted
smoothly when AC was applied again. Figure 41 shows
RDY output response. The PFC performances are shown
in Table 2. The power factor at full load is 0.992 and
0.990 for 110VAC and 230VAC, respectively.
Figure 38. Output Response at 110VAC, Full Load
Figure 39. Output Response at 110VAC, No Load
Figure 36. Inductor Current Waveforms at 110VAC
Figure 40. AC-Absent Detection Operation
Figure 37. Inductor Current Waveforms at 230VAC
Figure 41. RDY Output Response
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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19
AN-8035
Table 2.
VIN
85VAC
115VAC
230VAC
264VAC
Performance Results
POUT [W]
η [%]
PF
THD [%]
100
150
200
100
150
200
100
150
200
100
150
200
95.5
94.4
93.4
96.5
96.3
95.7
96.9
97.6
97.9
97.1
97.9
98.1
0.996
0.995
0.994
0.995
0.993
0.992
0.965
0.985
0.990
0.939
0.973
0.985
8.52
10.21
11.11
8.26
10.87
12.33
13.59
4.83
7.57
19.99
10.39
4.46
Definition of Terms
η is the efficiency.
θ is the displacement angle.
ΔB is the maximum flux swing of the core at nominal output power in Tesla.
Ae is the cross-sectional area of core.
AW is the window area of core.
BMAX is the maximum flux density of boost inductor at maximum output power in Tesla.
CCOMP,HF is the high-frequency compensation capacitance.
CCOMP,LF is the low-frequency compensation capacitance.
Ceff is the effective capacitance shown at the MOSFET drain pin.
CEA is the effective input capacitance to meet a given displacement factor.
CEXT is the external capacitance at drain-source to decrease the turn-off slope.
CEQ is the equivalent input capacitance.
CFB is the feedback capacitance parallel with RFB2.
COUT is the output capacitance.
COSS is the output capacitance of power MOSFET.
CPAR is the parasitic capacitance at drain-source of power MOSFET.
CZCD is the capacitance connected at ZCD pin to improve noise immunity.
dWIRE is the diameter of boost inductor winding wire.
DF is the displacement factor between input voltage and input current.
fC is the crossover frequency.
fCP is the high-frequency compensation pole to attenuate the switching ripple.
fCZ is the compensation zero.
fLINE is the line frequency.
fI is the integral gain of the compensator.
fP is the pole frequency in the PFC power stage transfer function.
fSW is the switching frequency.
fSW,MIN is the minimum switching frequency.
ICS,LIM is the pulse-by-pulse current limit level determined by sensing resistor.
IDOUT,AVE is the average current of output diode.
IIN,MAX is the maximum input current from the AC outlet.
IIN,MAXRMS is the maximum RMS (Root Mean Square) input current from the AC outlet.
IL is the inductor current at the nominal output power.
IL,PK is the maximum peak inductor current at the nominal output power.
IL,RMS is the RMS value of the inductor current at the nominal output power.
IL,DENSITY is the current density of the boost inductor coil.
IOUT is the nominal output current of the boost PFC stage.
IQ,RMS is the RMS current at the power switch.
IRDY,SK is the allowable sink current for the internal MOSFET in RDY pin.
KSAW is the internal gain of sawtooth generator (that of FAN7930 is 8.496×10-6).
L is the boost inductance.
NAUX is the number of turns of auxiliary winding in boost inductor.
NBOOST is the number of turns of primary winding in boost inductor.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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NWIRE is the number of strands of boost inductor winding wire.
PDOUT is the loss of output diode.
POUT is the nominal output power of boost PFC stage.
PQ,CON is conduction loss of the power MOSFET.
PQ,SWOFF is turn-off loss of power MOSFET.
PQ,DISCHRGE is the drain-source capacitance discharge loss and consumed at power MOSFET.
PQ is the total loss of power MOSFET made by PQ,CON, PQ,SWOFF, and PQ,DISCHARGE.
PRCS is the power loss caused by current-sense resistance.
RCOMP is the compensation resistance.
RCS is the power MOSFET current-sense resistance.
RDS,ON is the static drain-source on resistance of the power switch.
RFB1 is the feedback resistance between the INV pin and output voltage.
RFB2 is the feedback resistance between the INV pin and ground.
RL is the output load resistance in a given load condition.
RPULLUP is the pull-up resistance between the RDY pin and pull-up voltage.
RZCD is the resistor connected at the ZCD pin to optimize THD.
tHOLD is the required hold-up time.
tOFF is the inductor current reset time.
tON,MAX is the maximum on time fixed internally.
tON,MAX1 is the programmed maximum on time.
VCOMP is compensation pin voltage.
VCS,LIM is power MOSFET current-sense limit voltage.
VDROP,DOUT is the forward-voltage drop of output diode.
VIN(t) is the rectified line voltage.
VIN,PK is the amplitude of line voltage.
VLINE is RMS line voltage.
VLINE,MAX is the maximum RMS line voltage.
VLINE,MIN is the minimum RMS line voltage.
VLINE,OVP is the line OVP trip point in RMS.
VOPTO,F is the input forward voltage drop of opto-coupler.
VOUT is the PFC output voltage.
VOUT,MIN is the allowable minimum output voltage during the hold-up time.
VOUT,RDYH is the VOUT to trigger PFC RDY out pulls high.
VOUT,RDYL is the VOUT to trigger PFC RDY out drops to zero.
ΔVOUT,RIPPLE is the peak-to-peak output voltage ripple.
VPULLUP is the pull-up voltage for RDY pin.
VRDY,SAT is the internal saturation voltage of RDY pin.
VREF is the internal reference voltage for the feedback input.
VOVP,MAX is the maximum tolerance of Over-Voltage Protection specification
VST,COUT is the voltage stress at the output capacitor.
VST,Q is the voltage stress at the power MOSFET.
© 2010 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 5/3/10
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References
[1] Fairchild Datasheet FAN9612, Interleaved Dual BCM, PFC Controller
[2] Fairchild Datasheet FAN7930 Critical Conduction Mode PFC Controller
[3] Fairchild Application Note AN-6027, Design of Power Factor Correction Circuit Using FAN7530
[4] Fairchild Application Note AN-6086, Design Consideration for Interleaved BCM PFC using FAN9612
[5] Robert W. Erikson, Dragan Maksimovic, Fundamentals of Power Electronics, Second Edition, Kluwer Academic
Publishers, 2001.
Related Datasheets
FAN7930 — Critical Conduction Mode PFC Controller
FAN9611 / FAN9612 — Interleaved Dual BCM PFC Controllers
1N/FDLL 914/A/B / 916/A/B / 4148 / 4448 Small Signal Diode
PN2222A/MMBT2222A/PZT2222A NPN General Purpose Amplifier
FCPF20N60 — 600V N-Channel MOSFET, SuperFETTM
FFPF08H60S — 8A, 600V Hyperfast Rectifier
Author
YoungBae Park, System and Application Engineer
Important Notice
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HEREIN TO IMPROVE RELIABILITY, FUNCTION, OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF
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FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR
SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION.
As used herein:
1.
Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body, or
(b) support or sustain life, or (c) whose failure to perform
when properly used in accordance with instructions for use
provided in the labeling, can be reasonably expected to
result in significant injury to the user.
© 2010 Fairchild Semiconductor Corporation
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2.
A critical component is any component of a life support
device or system whose failure to perform can be reasonably
expected to cause the failure of the life support device or
system, or to affect its safety or effectiveness.
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