TI TPS61006DGS

TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
D Start-Up Into a Full Load With Supply
D
D
D
D
D
D Low-EMI Converter (Integrated Antiringing
Voltages as Low as 0.9 V Over Full
Temperature Range
Minimum 100-mA Output Current From
0.8-V Supply Voltage, 250 mA From 1.8 V
High Power Conversion Efficiency,
up to 90%
Power-Save Mode for Improved Efficiency
at Low Output Currents
Device Quiescent Current Less Than 50 µA
Added System Security With Integrated
Low-Battery Comparator
Switch Across Inductor)
D Micro-Size 10-Pin MSOP Package
D Evaluation Modules Available
(TPS6100xEVM–156)
Applications Include:
– Single- and Dual-Cell Battery Operated
Products
– MP3-Players and Wireless Headsets
– Pagers and Cordless Phones
– Portable Medical Diagnostic Equipment
– Remote Controls
D
·
description
The TPS6100x devices are boost converters intended for systems that are typically operated from a single- or
dual-cell nickel-cadmium (NiCd), nickel-metal hydride (NiMH), or alkaline battery. The converter output voltage
can be adjusted from 1.5 V to a maximum of 3.3 V and provides a minimum output current of 100 mA from a
single battery cell and 250 mA from two battery cells. The converter starts up into a full load with a supply voltage
of 0.9 V and stays in operation with supply voltages as low as 0.8 V.
The converter is based on a fixed-frequency, current-mode pulse-width-modulation (PWM) controller that goes
into power-save mode at low load currents. The current through the switch is limited to a maximum of 1100 mA,
depending on the output voltage. The current sense is integrated to further minimize external component count.
The converter can be disabled to minimize battery drain when the system is put into standby.
A low-EMI mode is implemented to reduce interference and radiated electromagnetic energy that is caused by
the ringing of the inductor when the inductor discharge-current decreases to zero. The device is packaged in
the space-saving 10-pin MSOP package.
L1
6 VBAT
140
7
SW
LBO 10
9 LBI
R2
Low Battery
Warning
TPS61006
FB 3
8 NC
ON
OFF
COMP 2
1 EN
GND
4
C1
100 pF
VOUT
Co
22 µF
VOUT 5
R3
R1
START-UP TIMING INTO 33-Ω LOAD
R4
10 kΩ
120
3
100
IOUT
2
80
60
40
1
C2
33 nF
20
EN
0
0
TYPICAL APPLICATION CIRCUIT FOR FIXED
OUTPUT VOLTAGE OPTION
I O– Output Current – mA
33 µH
VO = 3.3 V
VO – Output Voltage – V
Ci
10 µF
TPS61006
D1
0
2
4
6
8
10 12
Time – ms
14
16
18
20
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright  2000–2003, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
AVAILABLE OPTIONS
TA
PACKAGE
– 40°C to 85°C
OUTPUT VOLTAGE
(V)
PART NUMBER†
MARKING DGS
PACKAGE
Adj. from 1.5 V to 3.3 V
TPS61000DGS
ADA
1.5
TPS61001DGS
ADB
1.8
TPS61002DGS
ADC
2.5
TPS61003DGS
ADD
2.8
TPS61004DGS
ADE
3.0
TPS61005DGS
ADF
3.3
TPS61006DGS
ADG
10 Pin MSOP DGS
10-Pin
Adj. from 1.5 V to 3.3 V
TPS61007DGS
ADH
† The DGS package is available taped and reeled. Add R suffix to device type (e.g. TPS61000DGSR) to order quantities of
2500 devices per reel.
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
Compensation of error amplifier. Connect R-C-C network to set frequency response of control loop. See the
Application section for more details.
COMP
2
EN
1
I
Chip-enable input. The converter is switched on if EN is set high, and is switched off when EN is connected to
ground (shutdown mode).
FB
3
I
Feedback input for adjustable output voltage (TPS61000 only). The output voltage is programmed depending on
the values of resistors R1 and R2. For the fixed output voltage versions (TPS61000, TPS61002, TPS61003,
TPS61004, TPS61005, TPS61006), leave the FB pin unconnected.
NC/FBGND
8
Not connected (TPS61000, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006). A ground pin for the
feedback resistor divider for the TPS61007 only.
GND
4
Ground
LBI
9
I
Low-battery detector input. A low-battery signal is generated at the LBO pin when the voltage on LBI drops below
the threshold of 500 mV. Connect LBI to GND or VBAT if the low-battery detector function is not used. Do not leave
this pin floating.
LBO
10
O
Open-drain low-battery detector output. This pin is pulled low if the voltage on LBI drops below the threshold of
500 mV. A pullup resistor should be connected between LBO and VOUT.
SW
7
I
Switch input pin. The node between inductor and anode of the rectifier diode is connected to this pin.
VBAT
VOUT
6
I
Supply pin
5
O
Output voltage. For the fixed output voltage versions, the integrated resistive divider is connected to this pin.
DGS PACKAGE
(TOP VIEW)
EN
COMP
FB
GND
VOUT
1
10
2
9
3
8
4
7
5
6
LBO
LBI
NC/FBGND‡
SW
VBAT
‡ TPS61007 only
2
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
functional block diagram
fixed output-voltage option
L1
D1
CI
VOUT
SW
CO
Antiringing
Comparator
and Switch
VBAT
UVLO
Control Logic
Oscillator
Gate Drive
EN
LBI/LBO
Comparator
Current Sense
Current Limit
Slope Compensation
LBI
VREF
Comparator
Error
Amplifier
LBO
GND
Bandgap
Reference
COMP
adjustable output-voltage option (TPS61000 only)
L1
D1
CI
CO
SW
Antiringing
Comparator
and Switch
VBAT
UVLO
EN
LBI/LBO
Comparator
VOUT
Control Logic
Oscillator
Gate Drive
Current Sense
Current Limit
Slope Compensation
LBI
FB
VREF
Comparator
Error
Amplifier
LBO
GND
POST OFFICE BOX 655303
Bandgap
Reference
COMP
• DALLAS, TEXAS 75265
3
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
functional block diagram (continued)
adjustable output-voltage option (TPS61007 only)
L1
D1
CI
CO
SW
Antiringing
Comparator
and Switch
VBAT
UVLO
EN
LBI/LBO
Comparator
VOUT
Control Logic
Oscillator
Gate Drive
Current Sense
Current Limit
Slope Compensation
LBI
FB
VREF
Comparator
Error
Amplifier
LBO
Bandgap
Reference
FBGND
GND
COMP
detailed description
controller circuit
The device is based on a current-mode control topology using a constant-frequency pulse-width modulator to
regulate the output voltage. It runs at an oscillator frequency of 500 kHz. The current sense is implemented by
measuring the voltage across the switch. The controller also limits the current through the power switch on a
pulse-by-pulse basis. Care must be taken that the inductor saturation current is higher than the current limit of
the TPS6100x. This prevents the inductor from going into saturation and therefore protects both device and
inductor. The current limit should not become active during normal operating conditions.
The TPS6100x is designed for high efficiency over a wide output current range. Even at light loads the efficiency
stays high because the controller enters a power-save mode, minimizing switching losses of the converter. In
this mode, the controller only switches if the output voltage trips below a set threshold voltage. It ramps up the
output voltage with one or several pulses, and again goes into the power-save mode once the output voltage
exceeds the threshold voltage. The controller enters the power-save mode when the output current drops to
levels that force the discontinuous current mode. It calculates a minimum duty cycle based on input and output
voltage and uses the calculation for the transition out of the power-save mode into continuous current mode.
The control loop must be externally compensated with an R/C/C network connected to the COMP pin. See the
application section for more details on the design of the compensation network.
device enable
The device is put into operation when EN is set high. During start-up of the converter the input current from the
battery is limited until the voltage on COMP reaches its operating point. The device is put into a shutdown mode
when EN is set to GND. In this mode, the regulator stops switching and all internal control circuitry including
the low-battery comparator is switched off. The output voltage drops to one diode drop below the input voltage
in shutdown.
4
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
detailed description (continued)
under-voltage lockout
An under-voltage lockout function prevents the device start-up if the supply voltage on VBAT is lower than
approximately 0.7 V. This undervoltage lockout function is implemented in order to prevent the malfunctioning
of the converter. When in operation and the battery is being discharged, the device automatically enters the
shutdown mode if the voltage on VBAT drops below approximately 0.7 V.
If the EN pin is hardwired to VBAT and if the voltage at VBAT drops temporarily below the UVLO threshold voltage,
the device switches off and does not start up again automatically, even if the supply voltage rises above 0.9 V.
The device starts up again only after a signal change from low to high on EN or if the battery voltage is completely
removed.
low Battery detector circuit (LBI and LBO)
The low-battery detector circuit is typically used to supervise the battery voltage and to generate an error flag
when the battery voltage drops below a user-set threshold voltage. The function is active only when the device
is enabled. When the device is disabled, the LBO pin is high impedance. The LBO pin goes active low when
the voltage on the LBI pin decreases below the set threshold voltage of 500 mV ± 15 mV, which is equal to the
internal reference voltage. The battery voltage, at which the detection circuit switches, can be programmed with
a resistive divider connected to the LBI pin. The resistive divider scales down the battery voltage to a voltage
level of 500 mV, which is then compared to the LBI threshold voltage. The LBI pin has a built-in hysteresis of
10 mV. See the application section for more details about the programming of the LBI threshold.
If the low-battery detection circuit is not used, the LBI pin should be connected to GND (or to VBAT) and the LBO
pin can be left unconnected. Do not let the LBI pin float.
low-EMI switch
The device integrates a circuit which removes the ringing that typically appears on the SW-node when the converter enters the discontinuous current mode. In this case, the current through the inductor ramps to zero and
the Schottky diode stops conducting. Due to remaining energy that is stored in parasitic components of the
diode, inductor, and switch, a ringing on the SW pin is induced. The integrated antiringing switch clamps this
voltage internally to VBAT and therefore dampens this ringing.
The antiringing switch is turned on by a comparator that monitors the voltage between SW and VOUT. This
voltage indicates when the diode is reverse biased. The ringing on the SW-node is damped to a large degree,
reducing the electromagnetic interference generated by the switching regulator to a very great extent.
adjustable output voltage (TPS61000 and TPS61007 only)
The accuracy of the internal voltage reference, the controller topology, and the accuracy of the external resistor
divider determine the accuracy of the adjustable output voltage versions. The reference voltage has an
accuracy of ± 4% over line, load, and temperature. The controller switches between fixed frequency and
pulse-skip mode, depending on load current. This adds an offset to the output voltage that is equivalent to 1%
of VO. Using 1% accurate resistors for the feedback divider, a total accuracy of ± 6% can be achieved over the
complete temperature and output current range. The TPS61007 is an improved adjustable output voltage
version. Ground shift in the feedback loop was eliminated by adding a separate ground pin for the feedback
resistor divider (FBGND).
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
5
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
absolute maximum ratings†
Input voltage range, VI (VBAT, VOUT, COMP, FB, LBO, EN, LBI) . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to 3.6 V
Input voltage, VI (SW) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VOUT +0.7 V
Peak current into SW . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1300 mA
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See dissipation rating table
Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to 85°C
Maximum junction temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
Lead temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
TA ≤ 25_C
POWER RATING
DGS
424 mW
DERATING FACTOR
ABOVE TA = 25_C
3.4 mW/_C
TA = 70_C
POWER RATING
TA = 85_C
POWER RATING
271 mW
220 mW
recommended operating conditions
MIN
Supply voltage at VBAT
Output current
NOM
0.8
VBAT = 0.8 V
VBAT = 1.8 V
MAX
VO
UNIT
V
100
mA
250
Inductor
10
33
µH
Input capacitor
10
µF
Output capacitor
22
µF
Operating junction temperature, TJ
6
–40
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
125
°C
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
electrical characteristics over recommended operating free-air temperature range, VBAT = 1.2 V, EN
= VBAT (unless otherwise noted)
PARAMETER
VI
Input voltage for start-up
start up
VI
Input voltage once started
VO
Programmable output voltage
TEST CONDITIONS
RL = 33 Ω
RL = 3 kΩ,
TPS61000,
TPS61007
1.5
0.8 V < VI < VO,
1.2 V,
TPS61003
0.8 V < VI < VO,
1.6 V < VI < VO,
1.2 V,
VO
Output voltage
TPS61004
0.8 V < VI < VO,
1.6 V < VI < VO,
1.2 V,
TPS61005
0.8 V < VI < VO,
1.6 V < VI < VO,
1.2 V,
TPS61006
0.8 V < VI < VO,
1.6 V < VI < VO,
IO
Switch current limit
V
3.3
1.44
1.5
1.55
1.45
1.5
1.55
IO = 1 mA
IO = 100 mA
1.72
1.8
1.86
1.74
1.8
1.86
IO = 1 mA
IO = 100 mA
2.40
2.5
2.58
2.42
2.5
2.58
IO = 200 mA
IO = 1 mA
2.42
2.5
2.58
2.68
2.8
2.89
IO = 100 mA
IO = 200 mA
2.72
2.8
2.89
2.72
2.8
2.89
IO = 1 mA
IO = 100 mA
2.88
3.0
3.1
2.9
3.0
3.1
IO = 200 mA
IO = 1 mA
2.9
3.0
3.1
3.16
3.3
3.4
3.2
3.3
3.4
3.2
3.3
3.4
IO = 100 mA
IO = 200 mA
V
mA
250
TPS61001
0.5
TPS61002
0.65
TPS61004
V
100
TPS61003
ILIM
UNIT
V
IO = 1 mA
IO = 100 mA
VI = 0.8 V
VI = 1.8 V
Maximum continuous output current
MAX
0.8
IO = 100 mA
1.2 V,
TPS61002
TA = 25°C
0.8
0.8 V < VI < VO,
TYP
0.9
IO = 100 mA
1.2 V,
TPS61001
MIN
0.9
0 8 V < VI < VO
0.8
A
0.95
TPS61005
1
TPS61006
1.1
TPS61000,
TPS61007
VFB
Feedback voltage
468
f
Oscillator frequency
DMAX
Maximum duty cycle
rDS(on)
Switch-on resistance
VO = 3.3 V
Line regulation (see Note 1)
VI = 0.8 V to 1.25 V, IO = 50 mA
Load regulation fixed output voltage versions
(see Note 1)
VI = 1.2 V,
360
500
515
mV
500
840
kHz
85%
0.18
IO = 10 mA to 90 mA
0.3
0.27
Ω
%/V
0.25%
NOTE 1: Line and load regulation is measured as a percentage deviation from the nominal value (i.e., as percentage deviation from the nominal
output voltage). For line regulation, x %/V stands for ± x% change of the nominal output voltage per 1-V change on the input/supply
voltage. For load regulation, y% stands for ± y% change of the nominal output voltage per the specified current change.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
7
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
electrical characteristics over recommended operating free-air temperature range, VBAT = 1.2 V, EN
= VBAT (unless otherwise noted) (continued)
PARAMETER
TEST CONDITIONS
IQ
Quiescent current drawn from power
ower source
(current into VBAT and into VOUT)
IO = 0 mA, VEN = VI,
VO = 3.4 V
ISD
Shutdown current from power source
(current into VBAT and into VOUT)
VEN = 0 V
VIL
EN low-level input voltage
VIH
EN high-level input voltage
VIL
MIN
IFB
MAX
44
6
0.2
EN input current
EN = GND or VBAT
LBI low-level input voltage threshold
VLBI voltage decreasing
470
0.2 ×
VBAT
V
V
0.1
1
µA
500
530
mV
mV
µA
0.01
0.1
0.04
0.2
V
LBO output leakage current
VLBI = 0 V, VO = 3.3 V, IOL = 50 µA
VLBI = 650 mV, VLBO = 3.3 V
0.01
1
µA
FB input bias current (TPS61000, TPS61007 only)
VFB = 500 mV
0.01
0.1
µA
LBO low-level output voltage
PARAMETER MEASUREMENT INFORMATION
L1
Ci
10 µF
D1
33 µH
6 VBAT
7
SW
9 LBI
R2
Low Battery
Warning
LBO 10
List of Components:
IC1: Only fixed output versions
(unless otherwise noted)
L1: Coilcraft DO3308P–333
D1: Motorola Schottky Diode
MBRM120LT3
CI:
Ceramic
CO: Ceramic
TPS6100x
8 NC/FBGND
FB 3
ON
OFF
Co
22 µF
VOUT 5
R3
R1
COMP 2
1 EN
GND
4
R4
10 kΩ
C1
100 pF
C2
33 nF
Figure 1. Circuit Used for Typical Characteristics Measurements
8
µA
A
µA
10
LBI input current
UNIT
5
0.8 ×
VBAT
LBI input hysteresis
II
VOL
TYP
VBAT
VOUT
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
η
Efficiency
vs Output Current
2, 3
vs Inductor Type
4
vs Input Voltage
5
IO
VO
Maximum Output Current
vs Input Voltage
6
Output Voltage
vs Output Current
7
VO
IQ
TPS61007 Output Voltage
vs Output Current
8
No-Load Supply Current
vs Input Voltage
9
ISD
VI
Shutdown Current
vs Input Voltage
10
Minimum Start-Up Input Voltage
vs Load Current
11
ILIM
Switch Current Limit
vs Output Voltage
12
Output Voltage Ripple Amplitude
13
Output Voltage Ripple Amplitude
14
Load Transient Response
15
Line Transient Response
16
Start-Up Timing
17
EFFICIENCY
vs
OUTPUT CURRENT
EFFICIENCY
vs
OUTPUT CURRENT
100
100
VI = 2.4 V
VI = 1.2 V
90
90
80
80
VO = 3.3 V
70
VO = 1.5 V
Efficiency – %
Efficiency – %
70
VO = 3.3 V
60
50
40
VO = 2.8 V
60
50
40
30
30
20
20
10
10
0
0
1
10
100
1000
1
10
100
1000
IO – Output Current – mA
IO – Output Current – mA
Figure 2
Figure 3
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
9
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
TYPICAL CHARACTERISTICS
EFFICIENCY
vs
INDUCTOR TYPE
100
VI = 1.2 V
VO = 3.3 V
IO = 100 mA
95
90
Efficiency – %
85
80
75
70
65
60
55
50
Coilcraft
DO1608C
Coilcraft
DS1608C
Coiltronics
Coiltronics
UP1B
UP2B
Inductor Type
Sumida
CD43
Sumida
CD54
Figure 4
EFFICIENCY
vs
INPUT VOLTAGE
MAXIMUM OUTPUT CURRENT
vs
INPUT VOLTAGE
95
1
IO = 50 mA
0.90
I O – Maximum Output Current – A
90
Efficiency – %
85
IO = 100 mA
80
75
70
65
VO = 3.2 V
0.80
0.70
VO = 2.42 V
VO = 1.75 V
0.60
0.50
VO = 1.45 V
0.40
0.30
0.20
0.10
60
0.80
1.30
1.80
2.30
VI – Input Voltage – V
2.80
3.30
0
0.8
1
Figure 5
10
1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.8
VI – Input Voltage – V
Figure 6
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
3
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
TYPICAL CHARACTERISTICS
TPS61002/3/6
TPS61007
OUTPUT VOLTAGE
vs
OUTPUT CURRENT
OUTPUT VOLTAGE
vs
OUTPUT CURRENT
3.60
3.60
VI = 1.2 V
3.3 V
3.40
VI = 1.2 V
3.20
3.20
VO – Output Voltage – V
VO – Output Voltage – V
VO = 3.3 V
3.40
3
2.80
2.5 V
2.60
2.40
2
2.00
3
2.80
VO = 2.5 V
2.60
2.40
2.20
2
VO = 1.8 V
1.8 V
1.80
1.80
1.60
0.1
1.60
1
10
100
1000
1
IO – Output Current – mA
NO-LOAD SUPPLY CURRENT
vs
INPUT VOLTAGE
1000
SHUTDOWN CURRENT
vs
INPUT VOLTAGE
45
1800
TA = 85°C
TA = 85°C
40
35
1600
TA = 25°C
30
1400
I SD – Shutdown Current – nA
I Q – No-Load Supply Current – µ A
100
Figure 8
Figure 7
TA = –40°C
25
20
15
10
5
0
0.80
10
IO – Output Current – mA
1.30
1.80
2.30
2.80
VI – Input Voltage – V
3.30
3.80
1200
1000
800
600
400
TA = 25°C
200
0
0.80
TA = –40°C
1.30
1.80
2.30
2.80
3.30
3.80
VI – Input Voltage – V
Figure 9
Figure 10
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TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
TYPICAL CHARACTERISTICS
TPS61000, TPS61007
MINIMUM START-UP INPUT VOLTAGE
vs
LOAD CURRENT
SWITCH CURRENT LIMIT
vs
OUTPUT VOLTAGE
1.5
VO = min 3.2 V
VI = 1.2 V
0.85
I LIM – Switch Current Limit – A
VI – Minimum Start-Up Input Voltage – V
0.90
0.80
0.75
0.70
0.65
0.60
0
10
20
30
40
50
60
70
80
1
0.5
0
1.5 1.7 1.9 2.1 2.3 2.5 2.7 2.9 3.1 3.3 3.5
90 100
IO – Load Current – mA
VO – Output Voltage – V
Figure 11
Figure 12
TPS61006
TPS61006
OUTPUT VOLTAGE RIPPLE AMPLITUDE
OUTPUT VOLTAGE RIPPLE AMPLITUDE
3.36
VO – Output Voltage – V
3.34
IO = 2 mA
VO – Output Voltage – V
3.32
3.30
3.28
3.26
VI = 1.2 V
3.32
3.30
VSW
2
3.24
VSW
3.22
0
3.20
3.18
0
1
2
3
4
5
0
1
2
Time – µs
Time – ms
Figure 14
Figure 13
12
VOUT
3.34
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4
5
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SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
TPS61006
LINE TRANSIENT RESPONSE
VO – Output Voltage – V
TPS61006
LOAD TRANSIENT RESPONSE
VI = 1.2 V
RC = 33 kΩ
3.3
60
50 mA
40
20
5 mA
0
0
1
2
3.45
3
4
5
6
Time – ms
7
8
9
VOUT
IO = 50 mA
RC = 33 kΩ
3.35
3.25
V I – Input Voltage – V
3.2
3.55
1.2
VBAT
1
0.8
0
10
1
2
Figure 15
3
4
5
6
Time – ms
7
8
9
10
Figure 16
TPS61006
START-UP TIMING INTO 33-Ω LOAD
140
VOUT
120
3
100
IOUT
2
80
60
40
1
20
EN
I O – Output Current – mA
3.4
VO – Output Voltage – V
I O– Output Current – mA
VO – Output Voltage – V
TYPICAL CHARACTERISTICS
0
0
0
2
4
6
8
10 12
Time – ms
14
16
18
20
Figure 17
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SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
APPLICATION INFORMATION
The TPS6100x boost converter family is intended for systems that are powered by a single-cell NiCd or NiMH
battery with a typical terminal voltage between 0.9 V to 1.6 V. It can also be used in systems that are powered
by two-cell NiCd or NiMH batteries with a typical stack voltage between 1.8 V and 3.2 V. Additionally, singleor dual-cell, primary and secondary alkaline battery cells can be the power source in systems where the
TPS6100x is used.
programming the TPS61000 and TPS61007 adjustable output voltage devices
The output voltage of the TPS61000 and TPS61007 can be adjusted with an external resistor divider. The typical
value of the voltage on the FB pin is 500 mV in fixed frequency operation and 485 mV in the power-save
operation mode. The maximum allowed value for the output voltage is 3.3 V. The current through the resistive
divider should be about 100 times greater than the current into the FB pin. The typical current into the FB pin
is 0.01 µA, and the voltage across R4 is typically 500 mV. Based on those two values, the recommended value
for R4 is in the range of 500 kΩ in order to set the divider current at 1 µA. From that, the value of resistor R3,
depending on the needed output voltage VOUT, can be calculated using the following equation:
R3 + R4
ǒ
Ǔ
V
O *1
V
FB
ǒ
+ 500 kΩ
Ǔ
V
O *1
500 mV
(1)
If, as an example, an output voltage of 2.5 V is needed, a 2-MΩ resistor should be chosen for R3.
D1
L1
VO
33 µH
7
SW
Ci
10 µF
10 V
VOUT
CO
22 µF
10 V
5
R5
6 V
BAT
R1
9
LBO
LBI
FB
Low Battery
Warning
3
TPS61007
R4
R2
FBGND
1
R3
10
8
EN
Alkaline Cell
4 COMP
GND
2
RC
10 kΩ
CC1
100 pF
CC2
33 nF
Figure 18. Typical Application Circuit for Adjustable Output Voltage Option
The TPS61007 is an improved version of the TPS61000 adjustable output voltage device. The FBGND pin is
internally connected to GND.
14
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APPLICATION INFORMATION
programming the low battery comparator threshold voltage
The current through the resistive divider should be about 100 times greater than the current into the LBI pin.
The typical current into the LBI pin is 0.01 µA. The voltage across R2 is equal to the reference voltage that is
generated on-chip, which has a value of 500 mV ±15 mV. The recommended value for R2 is therefore in the
range of 500 kΩ. From that, the value of resistor R1, depending on the desired minimum battery voltage (VBAT),
can be calculated using the following equation:
R1 + R2
ǒ
Ǔ
V
TRIP * 1
V
REF
+ 500 kΩ
ǒ
V
Ǔ
BAT * 1
0.5 V
(2)
For example, if the low-battery detection circuit should flag an error condition on the LBO output pin at a battery
voltage of 1.0 V, a resistor in the range of 500 kΩ should be chosen for R1.
The output of the low battery comparator is a simple open-drain output that goes active low if the battery voltage
drops below the programmed threshold voltage on LBI. The output requires a pullup resistor with a
recommended value of 1MΩ, and should only be pulled up to the VOUT. If not used, the LBO pin can be left
floating.
inductor selection
The output filter of inductive switching regulators is a low pass filter of second order. It consists of an inductor and
a capacitor, often referred to as storage inductor and output capacitor.
To select an inductor, keep the possible peak inductor current below the current limit threshold of the power
switch in your chosen configuration. For example, the current limit threshold of the TPS61006’s switch is
1100 mA at an output voltage of 3.3 V. The highest peak current through the inductor and the switch depends on
the output load, the input (VBAT), and the output voltage (VOUT). Estimation of the maximum average inductor
current can be done using the following equation:
I
L
+ I
V
OUT
x
V
OUT
x 0.8
BAT
(3)
For example, for an output current of 100 mA at 3.3 V, at least 515-mA current flows through the inductor at a
minimum input voltage of 0.8 V.
The second parameter for choosing the inductor is the desired current ripple in the inductor. Normally it is advisable to work with a ripple of less than 20% of the average inductor current. A smaller ripple reduces the magnetic
hysteresis losses in the inductor as well as output voltage ripple and EMI. But in the same way, the regulation
time at load change rises. In addition, a larger inductor increases the total system cost.
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SLVS279C – MARCH 2000 – REVISED APRIL 2003
APPLICATION INFORMATION
With those parameters it is possible to calculate the value for the inductor:
L +
V
ǒ
–V
x V
BAT
BAT
OUT
∆I x f x V
L
OUT
Ǔ
(4)
Parameter f is the switching frequency and ∆IL is the ripple current in the inductor, i.e., 20% x IL.
In this example, the desired inductor has the value of 12 µH. With this calculated value and the calculated currents, it is possible to chose a suitable inductor. Care has to be taken that load transients and losses in the circuit
can lead to higher currents as estimated in equation 3. Also, the losses in the inductor caused by magnetic hysteresis losses and copper losses are a major parameter for total circuit efficiency.
The following inductors from different suppliers were tested. All work with the TPS6100x converter within their
specified parameters:
Table 1. Recommended Inductors
VENDOR
PART NUMBER
Coilcraft
DO1608P Series
DS1608P Series
DO3308 Series
Coiltronics
UP1B Series
UP2B Series
Murata
LQH3N Series
Sumida
CD43 Series
CD54 Series
CDR74B Series
TDK
NLC453232T Series
capacitor selection
The major parameter necessary to define the output capacitor is the maximum allowed output voltage ripple of
the converter. This ripple is determined by two parameters of the capacitor, the capacitance and the ESR. It is
possible to calculate the minimum capacitance needed for the defined ripple, supposing that the ESR is zero.
C
min
+
I
ǒ
–V
x V
BAT
OUT
OUT
f x ∆V x V
OUT
Ǔ
(5)
Parameter f is the switching frequency and ∆V is the maximum allowed ripple.
With a chosen ripple voltage of 15 mV, a minimum capacitance of 10 µF is needed. The total ripple will be larger
due to the ESR of the output capacitor. This additional component of the ripple can be calculated using the following equation:
∆V
16
ESR
+ I
OUT
xR
(6)
ESR
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APPLICATION INFORMATION
An additional ripple of 30 mV is the result of using a tantalum capacitor with a low ESR of 300 mΩ. The total ripple
is the sum of the ripple caused by the capacitance and the ripple caused by the ESR of the capacitor. In this
example, the total ripple is 45 mV. It is possible to improve the design by enlarging the capacitor or using smaller
capacitors in parallel to reduce the ESR or by using better capacitors with lower ESR, like ceramics. For example, a 10-µF ceramic capacitor with an ESR of 50 mΩ is used on the evaluation module (EVM). Tradeoffs have to
be made between performance and costs of the converter circuit.
A 10-µF input capacitor is recommended to improve transient behavior of the regulator. A ceramic capacitor or a
tantalum capacitor with a 100-nF ceramic capacitor in parallel placed close to the IC is recommended.
rectifier selection
The rectifier diode has a major impact on the overall converter efficiency. Standard diodes are not suitable for
low-voltage switched mode power supplies. A Schottky diode with low forward voltage and fast reverse recovery
should be used as a rectifier to minimize overall losses of the dc-dc converter. The maximum current rating of the
diode must be high enough for the application. The maximum diode current is equal to the maximum current in
the inductor that was calculated in equation 3. The maximum reverse voltage is the output voltage. The chosen
diode should therefore have a reverse voltage rating higher than the output voltage.
Table 2. Recommended Diodes
VENDOR
PART NUMBER
Motorola Surface Mount
MBRM120LT3
MBR0520LT1
Motorola Axial Lead
1N1517
ROHM
RB520S-30
RB160L–40
The typical forward voltage of those diodes is in the range of 0.35 to 0.45 V assuming a peak diode current of
600 mA.
compensation of the control loop
An R/C/C network must be connected to the COMP pin in order to stabilize the control loop of the converter. Both
the pole generated by the inductor L1 and the zero caused by the ESR and capacitance of the output capacitor
must be compensated. The network shown in Figure 19 satisfies these requirements.
RC
10 kΩ
COMP
CC1
100 pF
CC2
33 nF
Figure 19. Compensation of the Control Loop
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SLVS279C – MARCH 2000 – REVISED APRIL 2003
APPLICATION INFORMATION
Resistor RC and capacitor CC2 depend on the chosen inductance. For a 33-µH inductor, the capacitance of CC2
should be chosen to 33 nF, or in other words, if the inductor is xx µH, the chosen compensation capacitor should
be xx nF, the same number value. The value of the compensation resistor is then chosen based on the requirement to have a time constant of 0.3 ms for the R/C network of RC and CC2; hence for a 33-nF capacitor, a 10-kΩ
resistor should be chosen for RC.
Capacitor CC1 is depending on the ESR and capacitance value of the output capacitor, and on the value chosen
for RC. Its value is calculated using following equation:
C
C1
+
C
O
x ESR
COUT
R
C
3
(7)
For a selected output capacitor of 22 µF with an ESR of 0.2 Ω, and RC of 33 kΩ, the value of CC1 is in the range of
100 pF.
Table 3. Recommended Compensation Components
OUTPUT CAPACITOR
INDUCTOR
[µH]
RC
[kΩ]
CC1
[pF]
CC2
[nF]
0.2
10
100
33
0.3
15
100
22
22
0.4
33
100
10
10
0.1
33
100
10
CAPACITANCE
[µF]
ESR
[Ω]
33
22
22
22
10
10
schematic of TPS6100x evaluation modules (TPS6100xEVM–156)
J1
LP1
R6
C5
TPS6100x
R5
EN
C6
LBO
COMP
FB
OUT
R4
LBI
NC/FBGND
L1
R3
GND
R2
SW
VOUT
R1
IN
VBAT
C2
C1
C3
D1
Evaluation modules are available for device types TPS61000, TPS61002, TPS61003, and TPS61006.
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SLVS279C – MARCH 2000 – REVISED APRIL 2003
APPLICATION INFORMATION
suggested board layout and component placement (21 mm x 21 mm board size)
Figure 20. Top Layer Layout and Component Placement
Figure 21. Bottom Layer Layout and Component Placement
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TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006, TPS61007
SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
THERMAL INFORMATION
Implementation of integrated circuits in low-profile and fine-pitch surface-mount packages typically requires
special attention to power dissipation. Many system-dependent issues such as thermal coupling, airflow, added
heat sinks and convection surfaces, and the presence of other heat-generating components affect the powerdissipation limits of a given component.
Three basic approaches for enhancing thermal performance are listed below:
•
•
•
Improving the power dissipation capability of the PWB design
Improving the thermal coupling of the component to the PWB
Introducing airflow in the system
The maximum junction temperature (TJ) of the TPS6100x devices is 125°C. The thermal resistance of the
10-pin MSOP package (DGS) is RθJA = 294°C/W. Specified regulator operation is assured to a maximum
ambient temperature (TA) of 85 °C. Therefore, the maximum power dissipation is about 130 mW. More power
can be dissipated if the maximum ambient temperature of the application is lower.
T
P
=
D ( MAX )
J ( MAX ) – A
R
=
125 ° C – 85 ° C
Θ JA
294 ° C / W
= 136 mW
(8)
Under normal operating conditions, the sum of all losses generated inside the converter IC is less than 50 mW,
which is well below the maximum allowed power dissipation of 136 mW as calculated in equation 8. Therefore,
power dissipation is given no special attention.
Table 4 shows where the losses inside the converter are generated.
Table 4. Losses Inside the Converter
20
LOSSES
AMOUNTS
Conduction losses in the switch
36 mW
Switching losses
8 mW
Gate drive losses
2.3 mW
Quiescent current losses
< 1 mW
TOTAL
< 50 mW
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SINGLE- AND DUAL-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279C – MARCH 2000 – REVISED APRIL 2003
MECHANICAL DATA
DGS (S-PDSO-G10)
PLASTIC SMALL-OUTLINE PACKAGE
0,27
0,17
0,50
10
0,25 M
6
0,15 NOM
3,05
2,95
4,98
4,78
Gage Plane
0,25
1
0°–ā6°
5
3,05
2,95
0,69
0,41
Seating Plane
1,07 MAX
0,15
0,05
0,10
4073272/A 03/98
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion.
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