® VCA610 VCA 610 VCA 610 WIDEBAND VOLTAGE CONTROLLED AMPLIFIER FEATURES APPLICATIONS ● WIDE GAIN CONTROL RANGE: 80dB ● SMALL PACKAGE: 8-pin SOIC or DIP ● OPTICAL DISTANCE MEASUREMENT ● AGC AMPLIFIER ● WIDE BANDWIDTH: 30MHz ● LOW VOLTAGE NOISE: 2.2nV/ √Hz ● FAST GAIN SLEW RATE: 300dB/µs ● ULTRASOUND ● SONAR ● ACTIVE FILTERS ● EASY TO USE ● LOG AMPLIFIER ● IF CIRCUITS ● CCD CAMERAS DESCRIPTION The VCA610 is a wideband, continuously variable, voltage controlled gain amplifier. It provides lineardB gain control with high impedance inputs. It is designed to be used as a flexible gain control element in a variety of electronic systems. The VCA610 has a gain control range of 80dB (–40dB to +40dB) providing both gain and attenuation for maximum flexibility in a small 8-lead SO-8 or plastic dual-in-line package. The broad attenuation range can be used for gradual or controlled channel turn-on and turn-off for applications in which abrupt gain changes can create artifacts or other errors. In addition, the output can be disabled to provide –80dB of attenuation. Group delay variation with gain is typically less than ±2ns across a bandwidth of 1 to 15MHz. The VCA610 is designed with a very fast overload recovery time of only 200ns. This allows a large signal transient to overload the output at high gain, without obscuring low-level signals following closely behind. The excellent overload recovery time and distortion specifications optimize this device for lowlevel doppler measurements. The VCA610 has a noise figure of 3.5dB (with an RS of 200Ω) including the effects of both current and voltage noise. Instantaneous output dynamic range is 70dB for gains of 0dB to +40dB with 1MHz noise bandwidth. The output is capable of driving 100Ω. The high speed, 300dB/µs, gain control signal is a unipolar (0 to –2V) voltage that varies the gain linearly in dB/V. +5V –5V 6 –In +In VC Gain Control 7 2 8 5 1 VOUT 3 VCA610 International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 • Twx: 910-952-1111 Internet: http://www.burr-brown.com/ • FAXLine: (800) 548-6133 (US/Canada Only) • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 ©1992 Burr-Brown Corporation PDS-1140D Printed in U.S.A. January, 1995 SPECIFICATIONS ELECTRICAL All specifications at VS = ±5V, RL = 500Ω, RS = 0Ω, and TA = +25°C, unless otherwise noted. VCA610PA, UA PARAMETER CONDITIONS INPUT NOISE Input Voltage Noise Input Current Noise Noise Figure INPUT Input Impedance Bias Current Offset Current Differential Voltage Range Common-Mode Voltage Range Common-Mode Rejection MIN TYP MAX UNITS G = +40dB, RS = 0Ω G = –40dB to +40dB G = +40dB, RS = 200Ω 2.2 1.4 3.5 * * * nV/√Hz pA/√Hz dB Common-Mode All Gains All Gains 1 || 1 6 2 * * * * * * MΩ || pF µA µA –40 –40dB ≤ G ≤ +40dB TA = –25°C to +85°C +0.1V ≤ VC ≤ +2.0V, f = 1MHz ±0.5 ±0.01 –80 –40dB ≤ G ≤ +40dB G = –40dB (VC = 0V) to +40dB (VC = –2V) –3dB 80dB Gain Step VIN = 10mVDC, ∆ G = 80dB OUTPUT Voltage Swing(1) G = +40dB G = 0dB Output Voltage Limit Short-Circuit Current Instantaneous Dynamic Range (IDR)(6) G = 0dB to +40dB Offset Output Resistance +40 ±2 * –2 * ±2 * * 40 0 V dB * ±4 * * dB/V V MHz dB/µs ns MΩ || pF µA mV All Gains ∆ G = 80dB –3dB, All Gains VO = 1Vp-p, G ≥ 0dB 30 25 * * MHz MHz f = 1 to 15MHz f = 1 to 15MHz VO = 1Vp-p Small-Signal Small-Signal ±1 ±2 60 200 –50 15 * * * * * * ns ns V/µs ns dBc dBm * * * * V V Continuous to Common ±1 ±1.6 ±0.5 ±0.75 Symmetrical to Ground (±10%) ±80 VO = 1.5Vp-p G = –40dB f = 1MHz, All Gains 70 ±2 10 Applies to Temperature Drift Specs ±4.5 40 ±75 –25 –40 * * * * ±125 mA * * * * ±32 +85 +125 100 125 * * ±30 ±5.5 50 –26/+30 * * * * * * dB dB dB/°C dB 1 300 800 1 || 1 2 ±30 ±5V Recommended G = 0dB TEMPERATURE Specification Operation Thermal Resistance, θJA P, PA U, UA VCA610P, U MAX (1) FREQUENCY RESPONSE Bandwidth, Small-Signal Bandwidth, Large-Signal Group Delay Unit-to-Unit Variation 0dB ≤ G ≤ +40dB –40dB ≤ G < 0dB Output Slew Rate Overload Recovery(4) Two-tone Intermodulation Distortion(5) Two-tone, 3rd Order IMD Intercept(5) POWER SUPPLY Specification PSR Quiescent Current TYP ±2.5 50 GAIN Specified Gain Range Gain Accuracy(2) Gain Accuracy Temperature Drift Gain with Output Disabled GAIN CONTROL Gain Scaling Factor Control Voltage (VC) Bandwidth Slew Rate Settling Time: 1% Input Impedance Input Bias Current Output Offset Change(3) MIN * * * * dB mV Ω * V dB mA * * °C °C °C/W °C/W NOTES: (1) See Input/Output Range discussion in Applications Information Section (Figure 2). (2) Gain is laser trimmed and tested at gains of –40dB, 0dB, +15dB, +25dB, and +40dB; VIN =1Vp-p for gains less than 0dB; VOUT = 1V for gains of 0dB to +40dB. (3) Output offset change from offset at G = –40dB. (4) Gain = +40dB; Input step of 2V to 2mV; time required for output to return from saturation to linear operation. (5) VIN = 7mVp-p, VOUT = 700mVp-p (250mVrms); Output Power = –10dBm/tone, equal amplitude tones of 5MHz ±500Hz, G = +40dB. See typical performance curves. (6) With RS = 0Ω, and noise bandwidth of 1MHz. IDR = 20 log (VORMS/(eORMS x √BW)); where VORMS is rms output voltage, eORMS is output noise spectral density, and BW is noise bandwidth. ® VCA610 2 PACKAGE/ORDERING INFORMATION PIN CONFIGURATION Top View –In 8 –VS 7 +VS VOUT 6 5 DIP SO-8 VCA610 PRODUCT PACKAGE PACKAGE DRAWING NUMBER(1) VCA610PA VCA610P VCA610UA VCA610U 8-Pin Plastic DIP 8-Pin Plastic DIP SO-8 Surface-Mount SO-8 Surface-Mount 006 006 182 182 NOTE:(1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. 1 +In 2 3 4 ELECTROSTATIC DISCHARGE SENSITIVITY GND Gain No Control, Internal VC Connection This integrated circuit can be damaged by ESD. Burr-Brown recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ABSOLUTE MAXIMUM RATINGS Supply ................................................................................................. ±7V Differential Input Voltage ............................................................... Total VS Input Voltage Range ..................................... See Input Protection Section Storage Temperature Range .......................................... –65°C to +150°C Lead Temperature (soldering, DIP, 10s) ........................................ +300°C Lead Temperature (soldering, SO-8, 3s) ....................................... +260°C Output Short-Circuit to Ground (+25°C) ................................... Continuous Junction Temperature (TJ) ............................................................. +175°C ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® 3 VCA610 TYPICAL PERFORMANCE CURVES At VS = ±5V, RL = 500Ω, RS = 0Ω, and TA = +25°C, unless otherwise noted. GAIN vs CONTROL VOLTAGE SMALL-SIGNAL RESPONSE vs GAIN 50 VC = –2.0V 40 Specified Operating Range 40 30 VC = –1.5V 10 20 Gain (dB) 20 Gain (dB) 60 VC = –1.0V 0 –10 VC = –0.5V –20 0 –20 Large Signal, VO = 1Vp-p –40 –30 VC = 0V –40 Output Disabled for +0.1V ≤ VC ≤ +2V –60 –80 0.5 –50 0.1 1.0 10 100 0 –0.5 Frequency (MHz) GAIN CONTROL RESPONSE –1.5 –2 –2.5 FEEDTHRU WITH OUTPUT DISABLED 3 0 0 –20 Disabled Gain (dB) Normalized Response (dB) –1 Control Voltage, VC (V) –3 –6 –9 Output Disabled for +0.1V ≤ VC ≤ +2V –40 –60 –80 –12 –15 –100 10k 100k 1M 10M 100M 0.1 1 Frequency (Hz) VOLTAGE AND CURRENT NOISE vs GAIN 10k 10 100 Frequency (MHz) NOISE FIGURE vs SOURCE RESISTANCE 1k 25 100 Output Voltage Noise 10 100 1 10 –40 –20 0 +20 15 en2 + (inRS)2 4kTRS 10 0 0.1 +40 10 Gain (dB) 100 1k Source Resistance (Ω) ® VCA610 NFdB = 10 log 1 + 5 Input Current Noise 1 20 NF (dB) 1k Current Noise (pA/√Hz) Voltage Noise (nV/√Hz) Input-Referred Voltage Noise 4 10k 100k TYPICAL PERFORMANCE CURVES (CONT) At VS = ±5V, RL = 500Ω, RS = 0Ω, and TA = +25°C, unless otherwise noted. 2-TONE, 3rd ORDER INTERMODULATION INTERCEPT vs GAIN GROUP DELAY vs GAIN 20 16 10 Intercept Point (dBm) Group Delay (ns) 14 1MHz 12 10MHz 10 8 15MHz 6 –40 0 –10 1MHz 10MHz –20 –30 –20 0 20 –40 –40 40 –20 0 Gain (dB) OUTPUT OFFSET CHANGE vs GAIN Change from Output Offset at G = –40dB 100 PSR and CMR vs FREQUENCY 50 High Grade 50 Specification Limit 0 –50 –PSR 40 30 CMR 20 +PSR –100 G = 0dB 10 –150 0 –40 –20 0 20 40 10k 100k Gain (dB) 1M 10M 100M Frequency (Hz) “DIAMOND PATTERN” RESPONSE LARGE SIGNAL RESPONSE +500 Output Voltage (mV) +500 Output Voltage (mV) 40 60 Low Grade Rejection (dB) Output Offset Change (mV) 150 20 Gain (dB) G = +40dB f = 5MHz RL = 500Ω 0 –500 0 100 0 –500 0 200 25 50 Time (µs) Time (µs) ® 5 VCA610 APPLICATIONS INFORMATION to +40dB range as VC varies from 0 to –2V. Optionally, making VC slightly positive, ≥ 0.1V, effectively disables the amplifier, giving 80dB of attenuation at low frequencies. CIRCUIT DESCRIPTION The VCA610 is a wideband voltage amplifier with a voltage-controlled gain, as modeled in Figure 1. The circuit’s basic voltage amplifier responds to the control of an internal gain control amplifier. At its input, the voltage amplifier presents the high impedance of a differential stage, permitting flexible input impedance matching. To preserve termination options, no internal circuitry connects to the input bases of this differential stage. For this reason, the user should provide DC paths for the input base currents either through a grounded termination resistor or a direct connection to ground. The differential input stage also permits rejection of commonmode signals to remove ground bounce effects. At its output, the voltage amplifier presents the low impedance of class A-B emitter-follower stage, again simplifying impedance matching. An open-loop design produces wide bandwidth at all gain levels and avoids the added overload-recovery and propagation delays of feedback designs. Repeated use of differential stages minimizes offset effects for reduced feedthrough of the gain control signal. A ground-sensing, differential to single-ended converter retains the low offset in the amplifier output stage. Internally, the gain control circuit varies the amplifier gain through a time-proven method which exploits the linear relationship between the transconductance, gm, of a bipolar transistor and the transistor’s bias current. Varying the bias currents of differential stages varies gm to control the voltage gain of the VCA610. Relying on transistor gm to set gain also avoids the need for a noise-producing gain-set resistor in the amplifier input circuit. This reliance normally introduces a high thermal sensitivity to the gain. However, the VCA610 employs specialized analog signal processing that removes this thermal effect. INPUT/OUTPUT RANGE The VCA610’s 80dB gain range allows the user to handle an exceptionally wide range of input signal levels. If the unit’s input and output voltage range specifications are exceeded, however, signal distortion and amplifier overloading will occur. The VCA610’s maximum input and output voltage range is best illustrated in Figure 2. OUTPUT POWER vs INPUT POWER +10 V– Output Power (dBm) GND Voltage Amplifier –In VO + –1dBm, Compression Points –10 –20 G = +40dB + 0.633 + G = +20dB –30 –40 Ideal — Actual G = –20dB + 0.02 +13.5dBm 6E-3 (3Vp-p) G = –40dB –60 –60 –50 –40 –30 –20 –10 0 +10 (6E-4) (0.002) (0.006) (0.02) (0.063) (0.2) (0.63) (2) Gain Control Circuit 0.20 0.063 G = 0dB +In –50 6.33 3.0 2.0 + VOUT (Vp-p) 0 V+ Rs = 50Ω RL = 500Ω f = 1MHz 2E-3 +20 (6.3) Input Power in dBm (Differential Input Voltage in Vp-p) FIGURE 2. Input and Output Range. VC Gain Control Amplifier Figure 2 plots output power vs input power for five voltage gains spaced at 20dB intervals. The 1dBm compression points occur where the actual output power (solid lines) deviates by –1dBm from the ideal output power (dashed lines). Compression is produced by different mechanisms depending on the selected gain. For example, at G = –40dB, 1dBm compression occurs when the input signal approaches approximately 3Vp-p (13.5dBm for RS = 50Ω). Input overloading is the compression mechanism for all gains from –40dB to about –5dB. For gains between –5dB and +5dB, the compression is due to internal gain stage overloading. Compression over this gain range occurs when the output signal becomes distorted as internal gain stages become overdriven. At G = 0dB, 1dBm compression occurs when the input exceeds approximately 1.5Vp-p (7.5dBm). At gains greater than about 5dB, the compression mechanism is due to output stage overloading. Output overloading occurs FIGURE 1. Block Diagram of the VCA610. A user-applied voltage, VC, controls the amplifier’s gain magnitude through a high-speed control circuit. Gain polarity can be either inverting or noninverting depending upon the amplifier input driven by the input signal. Use of the inverting input is recommended since this connection tends to minimize positive feedback from the output to the noninverting input. The gain control circuit presents the high input impedance of a noninverting op amp connection. Control voltage VC varies the amplifier gain according to the exponential relationship G(V/V) = 10 –2 (Vc +1). This translates to the linear, logarithmic relationship G(dB) = – 40 – 40VC. Thus, G(dB) varies linearly over the specified –40dB ® VCA610 6 when either the maximum output voltage swing or output current is exceeded. The VCA610’s high output current of ±80mA insures that virtually all output overloads will be limited by voltage swing rather than by current limiting. At G = +40dB, 1dBm compression occurs when the output voltage approaches 3Vp-p (3.5dBm for RL = 500Ω). Table I below summarizes these results. GAIN RANGE –40dB < G < –5dB –5dB < G < +5dB +5dB < G < +40dB OUTPUT COMPRESSION MECHANISM TO PREVENT OPERATE WITHIN Input Stage Overload Input Voltage Range V+ R1 VIN 10kΩ RV 100kΩ 1µF V– VCA610 R2 10Ω VO VC 3a) Optional Offset Adjustment. Internal Stages Overloading Output Voltage Range Output Stage Overload VO VCA610 Output Voltage Range Cp TABLE I. Output Signal Compression. f–3dB = 1 2π Rp Cp RP WIRING PRECAUTIONS Maximizing the VCA610’s capability requires some wiring precautions and high-frequency layout techniques. In general, printed circuit board conductors should be as short and as wide as possible to provide low resistance, low impedance signal paths. Stray signal coupling from the output or power supplies to the inputs should be minimized. Unused inputs should be grounded as close to the package as possible. VC 3b) Control Line Filtering. FIGURE 3. Optional Offset Adjustment and Control Line Filtering. one selected gain. Selecting the maximum gain optimizes offset performance for higher gains where high amplification of the offset effects produces the greatest output offset. Two features minimize the offset control circuit’s noise contribution to the amplifier input circuit. First, making the resistance of R2 a low value minimizes the noise directly introduced by the control circuit. This reduces both the thermal noise of the resistor and the noise produced by the resistor with the amplifier’s input noise current. A second noise reduction results from capacitive bypass of the potentiometer output. This filters out power supply noise that would otherwise couple to the amplifier input. Low impedance ground returns for signal and power are essential. Proper supply bypassing is also extremely critical and must always be used. Both power supply leads should be bypassed to ground as close as possible to the amplifier pins. Tantalum capacitors (1µF to 10µF) with very short leads are recommended. Surface mount bypass capacitors will provide excellent results due to their low lead inductance. OVERLOAD RECOVERY As shown in Figure 2, the onset of overload occurs whenever the actual output begins to deviate from the ideal expected output. If possible, the user should operate the VCA610 within the linear regions shown in order to minimize signal distortion and overload delay time. However, instances of amplifier overload are actually quite common in Automatic Gain Control (AGC) circuits which involve the application of variable gain to signals of varying levels. The VCA610’s design incorporates circuitry which allows it to recover from most overload conditions in 200ns or less. Overload recovery time is defined as the time required for the output to return from overload to linear operation following the removal of either an input or gain control overdrive signal. This filtering action would diminish as the wiper position approaches either end of the potentiometer but practical conditions prevent such settings. Over its full adjustment range, the offset control circuit produces a ±5mV offset correction for the values shown. However, the VCA610 only requires one tenth of this range for offset correction, assuring that the potentiometer wiper will always be near the potentiometer center. With this setting, the resistance seen at the wiper remains high and this stabilizes the filtering function. GAIN CONTROL The VCA610’s gain is controlled by means of a unipolar negative voltage applied between ground and the gain control input, pin 3. If use of the output disable feature is required, a ground-referenced bipolar voltage is needed. Output disable occurs for +0.1V ≤ VC ≤ +2V, and produces 80dB of attenuation. The control voltage should be limited to +2V in disable mode, and –2V in the gain mode in order to prevent saturation of internal circuitry. OFFSET ADJUSTMENT Where desired, the offset of the VCA610 can be removed as shown in Figure 3. This circuit simply presents a DC voltage to one of the amplifier’s inputs to counteract the offset error voltage. For best offset performance, the trim adjustment should be made with the amplifier set at the maximum gain of the intended application. The offset voltage of the VCA610 varies with gain, limiting the complete offset cancellation to ® 7 VCA610 ULTRASOUND TGC AMPLIFIER The Figure 5 block diagram illustrates the fundamental configuration common to pulse-echo imaging systems. A piezoelectric crystal serves as both the ultrasonic pulse generator and the echo monitor transducer. A transmit/ receive (T/R) switch isolates the monitor amplifier from the crystal during the pulse generation cycle and, then, connects the amplifier to the crystal during the echo monitor cycle. The VCA610’s gain control input has a –3dB bandwidth of 1MHz and varies with frequency as shown in the Typical Performance Curves. This wide bandwidth, although useful for many applications, can allow high frequency noise to modulate the gain control input. In practice, this can be easily avoided by filtering the control input as shown in Figure 3b. RP should be no greater than 100Ω so as not to introduce gain errors by interacting with the gain control’s input bias current of 2µA. INPUT PROTECTION Electrostatic damage (ESD) has been well recognized for MOSFET devices, but any semiconductor device deserves protection from this potentially damaging source. The VCA610 incorporates on-chip ESD protection diodes as shown in Figure 4. This eliminates the need for the user to add external protection diodes, which can add capacitance and degrade AC performance. T/R Switch ADC & DSP VCA610 Transducer VC VC Transmit Receive t 0 +VS ESD Protection diodes internally connected to all pins. External Pin –2V Internal Circuitry FIGURE 5. Typical Ultrasound Application. During the monitor (receive) cycle, the control voltage VC, varies the amplifier gain. The gain is varied for three basic signal processing requirements of a transducer array based beamformer: compensation for depth attenuation effects, sometimes called Time Gain Compensation (TGC); receive apodization or windowing for reducing side lobe energy; and dynamic aperture sizing for better near field resolution. –VS FIGURE 4. Internal ESD Protection. All pins on the VCA610 are internally protected from ESD by means of a pair of back-to-back reverse-biased diodes to either power supply as shown. These diodes will begin to conduct when the pin voltage exceeds either power supply by about 0.7V. This situation can occur with loss of the amplifier’s power supplies while a signal source is still present. The diodes can typically withstand a continuous current of 30mA without destruction. To insure long term reliability, however, diode current should be externally limited to 10mA whenever possible. The internal protection diodes are designed to withstand 2.5kV (using Human Body Model) and provides adequate ESD protection for most normal handling procedures. However, static protection is strongly recommended since static damage can cause subtle changes in amplifier operational characteristics without necessarily destroying the device. Time gain compensation increases the amplifier’s gain as the ultrasound signal moves through the material to compensate for signal attenuation versus material depth. For this purpose, a ramp signal applied to the VCA610 gain control input linearly increases the dB gain of the VCA610 with time. The gain control provides signal apodization or windowing with transducer arrays connected to amplifier arrays. Selective weighting of amplifier gains across the transducer aperture suppresses side lobe effects in the beamformer output to reduce image artifacts. Gain controlled attenuation or disabling the amplifier can be used to dynamically size the array aperture for better near field resolution. The controlled attenuation of the VCA610 minimizes switching artifacts and eliminates the bright radial rings that can result. The VCA610’s 80dB gain range accommodates these functions. APPLICATIONS The electronically variable gain of the VCA610 suits pulseecho imaging systems well. Such applications include medical imaging, non-destructive structural inspection and optical distance measurement. The amplifier’s variable gain also serves AGC amplifiers, amplitude-stabilized oscillators, log amplifiers and exponential amplifiers. The discussions below present examples of these applications. WIDE-RANGE LOW-NOISE VCA Figure 6 combines two VCA610s in series, extending the overall gain range and improving noise performance. This combination produces a gain equal to the sum of the two amplifier’s logarithmic gains for a composite range of ® VCA610 8 WIDE-RANGE AGC AMPLIFIER The voltage-controlled gain feature of the VCA610 makes this amplifier ideal for precision AGC applications with control ranges as large as 60dB. The AGC circuit of Figure 7 adds an op amp and diode for amplitude detection, a holding capacitor to store the control voltage and resistors R1 through R3 that determine attack and release times. Resistor R4 and capacitor CC phase compensate the AGC feedback loop. The op amp compares the positive peaks of output VO with a DC reference voltage VR. Whenever a VO peak exceeds VR, the OPA620 output swings positive, forward biasing the diode and charging the holding capacitor. This drives the capacitor voltage in a positive direction, reducing the amplifier gain. R3 and the CH largely determine the attack time of this AGC correction. –80dB to +80dB. Simply connecting VC1 and VC2 to the same 0 to –2V gain control voltage can produce this range, however, separate control voltages for the two amplifiers offer a noise performance improvement. In that configuration, each amplifier separately controls one half the gain range in a manner that always holds G1 at the maximum level possible. VCA1 VCA2 VIN G1 G2 VO VC1 VC2 GAIN –80dB to 0dB VIN 0dB to 80dB 2mV to 2V 100kHz VC1 0 to –2V –2V G1 –40dB to +40dB +40dB VC2 0V 0 to –2V G2 –40dB –40dB to +40dB VO VCA610 VC R3 1kΩ HP5082 OPA620 R2 50kΩ R1 50kΩ FIGURE 6. Two Series Connected VCA610s Expand the Gain Range and Improve Noise Performance. CH 0.1µF CC At higher gains, variation of VC2 alone makes VCA2 provide all of the gain control, leaving the gain of VCA1 fixed at its maximum of 40dB. This gain maximum corresponds to the maximum bias currents in VCA1, minimizing this amplifier’s noise. Thus, for composite circuit gains of 0dB to +80dB, VCA1 serves as a low-noise, fixed-gain preamp. Between gain corrections, resistor R1 charges the capacitor in a negative direction, increasing the amplifier gain. R1, R2 and CH determine the release time of this action. Resistor R2 forms a voltage divider with R1, limiting the maximum negative voltage developed on CH. This limit prevents input overload of the VCA610’s gain control circuit. CW2 RW2 300Ω 4700pF RW1 300Ω f = 1/2πRW1CW1 RW1 = RW2 CW1 = CW2 VCA610 VO VC VOPEAK = VR R3 HP5082 OPA620 R4 CC 100Ω 1kΩ R2 50kΩ 0.1 VDC FIGURE 7. This AGC Circuit Maintains a Constant Output Amplitude for a 1000:1 Input Range. For lower composite gains, VCA1 provides the gain control and VCA2 acts as a fixed attenuator. There, variation of VC1 varies G1 from –40dB to +40dB while VC2 remains fixed at 0V for G2 = –40dB. This mode produces the –80dB to 0dB segment of the composite gain range. R1 50kΩ R4 100Ω V R 50pF V– CW1 4700pF VOUT PEAK = VR CH 1µF VR 0.1 VDC 10pF V– FIGURE 8. Adding Wein-bridge Feedback to the AGC Circuit of Figure 7 Produces an Amplitude Stabilized Oscillator. ® 9 VCA610 STABILIZED WEIN-BRIDGE OSCILLATOR Adding Wein-bridge feedback to the above AGC amplifier produces an amplitude-stabilized oscillator. Shown in Figure 8, this alternative requires the addition of just two resistors (RW1, RW2) and two capacitors (CW1, CW2). reveals an option and a constraint. In Figure 9, VR represents a DC reference voltage. Optionally, making this voltage a second signal produces log-ratio operation. Either way, the Log term’s argument constrains the polarities of VR and VIN. These two voltages must be of opposite polarities to ensure a positive argument. This polarity combination results when VR connects to the inverting input of the VCA610. Alternately, switching VR to this amplifier’s noninverting input removes the minus sign of the log term’s argument. Then, both voltages must be of the same polarity to produce a positive argument. In either case, the positive polarity requirement of the argument restricts VIN to a unipolar range. Connecting the feedback network to the amplifier’s noninverting input introduces positive feedback to induce oscillation. The feedback factor displays a frequency dependence due to the changing impedances of the CW capacitors. As frequency increases, the decreasing impedance of the CW2 increases the feedback factor. Simultaneously, the decreasing impedance of the CW1 decreases this factor. The above VOL expression reflects a circuit gain introduced by the presence of R1 and R2. This feature adds a convenient scaling control to the circuit. However, a practical matter sets a minimum level for this gain. The voltage divider formed by R1 and R2 attenuates the voltage supplied to the VC terminal by the op amp. This attenuation must be great enough to prevent any possibility of an overload voltage at the VC terminal. Such an overload saturates the VCA610’s gain control circuitry, reducing the amplifier’s gain. For the feedback connection of Figure 9, this overload condition permits a circuit latch. To prevent this, choose R1 and R2 to ensure that the op amp can not possibly deliver more than 2.5V to the VC terminal. Analysis shows that the maximum factor occurs at f = 1/2πRWCW, making this the frequency most conducive to oscillation. At this frequency the impedance magnitude of CW equals RW and inspection of the circuit shows that this condition produces a feedback factor of 1/3. Thus, selfsustaining oscillation requires a gain of three through the amplifier. The AGC circuitry establishes this gain level. Following initial circuit turn on, R1 begins charging CH negative, increasing the amplifier gain from its minimum. When this gain reaches three, oscillation begins at f = 1/2πRWCW and R1’s continued charging effect makes the oscillation amplitude grow. This growth continues until that amplitude reaches a peak value equal to VR. Then, the AGC circuit counteracts the R1 effect, controlling the peak amplitude at VR by holding the amplifier gain at a level of three. Making VR an AC signal, rather than a DC reference, produces amplitude modulation of the oscillator output. VR –10mV VC R1 470Ω LOW-DRIFT WIDEBAND LOG AMP The VCA610 can be used to provide a 250kHz (–3dB) log amp with low offset voltage and low gain drift. ( VOL = – 1 + R2 330Ω The exponential gain control characteristic of the VCA610 permits simple generation of a temperature-compensated logarithmic response. Enclosing the exponential function in an op amp feedback path inverts this function, producing the log response. Figure 9 shows the practical implementation of this technique. A DC reference voltage, VR, sets the VCA610 inverting input voltage. This makes the amplifier’s output voltage VOA = – GVR where G = 10 -2 (Vc + 1). VOL OPA620 R1 R2 ) 1 + 0.5 Log (–VIN/VR) R3 100Ω VIN CC 50pF FIGURE 9. Driving the Gain Control Pin of the VCA610 with a Feedback Amplifier Produces a TemperatureCompensated Log Response. A second input voltage also influences VOA through control of gain G. The feedback op amp forces VOA to equal the input voltage VIN connected at the op amp inverting input. Any difference between these two signals drops across R3, producing a feedback current that charges CC. The resulting change in VOL adjusts the gain of the VCA610 to change VOA. At equilibrium, VOA = VIN = –VR10 -2 (Vc +1). The op amp forces this equality by supplying the gain control voltage VC = R1 VOL /(R1 + R2). Combining the last two expressions and solving for VOL yields the circuit’s logarithmic response. LOW-DRIFT WIDEBAND EXPONENTIAL AMP A common use of the Log amp above involves signal companding. The inverse function, signal expanding, requires an exponential transfer function. The VCA610 produces this latter response directly as shown in Figure 10. DC reference VR again sets the amplifier’s input voltage and the input signal VIN now drives the gain control point. Resistors R1 and R2 attenuate this drive to prevent overloading the gain control input. Setting these resistors at the same values as in the preceding Log amp produces an exponential amplifier with the inverse function of the Log amp. VOL = – (1 + R2/R1) [1 + 0.5LOG (–VIN /VR)] Examination of this result illustrates several circuit characteristics. First, the argument of the Log term, –VIN/VR, ® VCA610 VOA = –G VR VCA610 10 VR –10mV Finite loop gain and a signal swing limitation set performance boundaries for the circuit. Both limitations occur when the VCA610 attenuates rather than amplifies the feedback signal. These two limitations reduce the circuit’s utility at the lower extreme of the VCA610’s gain range. For –1 ≤ VC ≤ 0, this amplifier produces attenuating gains in the range from 0dB to –40dB. This directly reduces the net gain in the circuit’s feedback loop, increasing gain error effects. Also, this attenuation transfers an output swing limitation from the OPA620 output to the overall circuit’s output. Note that OPA620 output voltage, VOA, relates to VO through the expression VO = GVOA. Thus, a G < 1 limits the maximum VO swing to a value less than the maximum VOA swing. VCA610 VO = –VR10 –2 [R1 VIN /(R1 + R2) + 1] VC VIN R2 330Ω R1 470Ω FIGURE 10. Signal Drive of the VCA610 Gain Control Pin Produces and Exponential Response, Re-expanding Signal Companded by Figure 9. However, the circuit shown provides greater output swing than the more common multiplier implementation. The latter replaces the VCA610 of the figure with an analog multiplier having a response of VO = XY/10. Then, X = VOA and Y = VC, making the circuit output voltage VO = VOAVC/10. Thus, the multiplier implementation amplifies VOA by a gain of VC/ 10. Circuit constraints require that VC ≤ 10, making this gain ≤ 1. Thus, the multiplier performs only as a variable attenuator and never provides amplification. As a result, the voltage swing limitation of VOA restricts the VO swing throughout most of the circuit’s control range. Replacing the multiplier with the VCA610 shown permits equivalent gains greater > 1. Then, operating the VCA610 with gains in the range of one to 100 avoids the reduction in output swing capability. VOLTAGE-CONTROLLED LOW-PASS FILTER In the circuit of Figure 11, the VCA610 serves as the variable gain element of a voltage-controlled low-pass filter. As will be described, this implementation expands the circuit’s voltage swing capability over that normally achieved with the equivalent multiplier implementation. The circuit’s response pole responds to control voltage VC according to the relationship fP = G/2πR2C where G = 10–2 (VC + 1). With the components shown, the circuit provides a linear variation of the low-pass cutoff from 300Hz to 1MHz. R2 R1 C 330Ω 0.047µF 330Ω VI OPA620 VOLTAGE-CONTROLLED HIGH-PASS FILTER A circuit analogous to the above low-pass filter produces a voltage-controlled high-pass response. The gain control provided by the VCA610 of Figure 12 varies this circuit’s response zero from 1Hz to 10kHz according to the relationship FZ ≈ 1/2πGR1C where G = 10 –2 (VC + 1). VOA VCA610 VO VC VO 1 R2 –2(V + 1) C VI = – R1 1 + R2Cs/G fP = G/2πR2C G = 10 R1 R2 33kΩ 33kΩ VI FIGURE 11. This Voltage-Tuneable Low-Pass Filter Produces a Variable Cutoff Frequency with a 3,000:1 Range. VCA610 The response control results from amplification of the feedback voltage applied to R2. Consider first the case where the VCA610 produces G = 1. Then, the circuit performs as if this amplifier were replaced by a short circuit. Visually doing so leaves a simple voltage amplifier with a feedback resistor bypassed by a capacitor. This basic circuit produces a response pole at fP = 1/2πR2C. C R3 0.047µF 33Ω VOA OPA620 VO VC For R 3 << GR 1 and f << 1/2πR 3 Cs, R VO = – 2 (1 + GR 1 Cs), f Z = 1/2πGR 1 C VI R1 where G = 10 –2(V C + 1) For G > 1, the circuit applies a greater voltage to R2, increasing the feedback current this resistor supplies to the summing junction of the OPA620. The increased feedback current produces the same result as if R2 had been decreased in value in the basic circuit described above. Decreasing the effective R2 resistance moves the circuit’s pole to a higher frequency, producing the fP = G/2πR2C response control. FIGURE 12. A Voltage-Tunable High-Pass Filter Produces a Response Zero Variable from 1Hz to 10kHz. ® 11 VCA610 VOLTAGE-CONTROLLED BAND-PASS FILTER The VCA610’s variable gain also provides voltage control over the center frequency of a band-pass filter. Shown in Figure 13, this filter follows from the state-variable configuration with the VCA610 replacing the inverter common to that configuration. Variation of the VCA610 gain moves the filter’s center frequency through a 100:1 range following the relationship fO = [10 –(VC + 1) ]/2πRC. To visualize the circuit’s operation, consider a circuit condition and an approximation that permit replacing the VCA610 and R3 with short circuits. First consider the case where the VCA610 produces G = 1. Then, replacing this amplifier with short circuit leaves the operation unchanged. In this shorted state, the circuit is simply a voltage amplifier with an R–C bypass around R1. The resistance of this bypass, R3, serves only to phase compensate the circuit and practical factors make R3 << R1. Neglecting R3 for the moment, the circuit becomes just a voltage amplifier with capacitive bypass of R1. This circuit produces a response zero at fZ = 1/2πR1C. As before, variable gain controls a circuit time constant to vary the filter response. The gain of the VCA610 amplifies or attenuates the signal driving the lower integrator of the circuit. This alters the effective resistance of the integrator time constant producing the response Adding the VCA610 as shown permits amplification of the signal applied to capacitor C and produces voltage control of the frequency fZ. Amplified signal voltage on C increases the signal current conducted by the capacitor to the op amp feedback network. The result is the same as if C had been increased in value to GC. Replacing C with this effective capacitance value produces the circuit’s control expression fZ = 1/2πR1GC. VO –s/nRC = 2 VI s + s/nRC + G/R2C2 Evaluation of this response equation reveals a passband gain of AO = –1, a bandwidth of BW = 1/2πnRC and a selectivity of Q = n10 –(VC + 1). Note that variation of control voltage VC alters Q but not bandwidth. Two factors limit the high-frequency performance of the resulting high-pass filter. The finite bandwidth of the op amp and the circuit’s phase compensation produce response poles. These limit the frequency duration of the high-pass response. Selecting the R3 phase compensation with the equation R3 = √(R1/2πfCC) assures stability for all values of G and sets the circuit’s bandwidth at BW = √(fC/2πR1C). Here, fC is the unity-gain crossover frequency of the op amp used. With the components shown, BW = 100kHz. This bandwidth provides a high-pass response duration of five decades of frequency for fZ = 1Hz, dropping to one decade for fZ = 10kHz. The gain provided by the VCA610 restricts the output swing of the filter. Output signal VO must be constrained to a level that does not drive the VCA610 output, VOA, into its saturation limit. Note that these two outputs have voltage swings related by VOA = GVO. Thus, a swing limit VOAL imposes a circuit output limit of VOL ≤ VOAL/G. The output voltage limit of the VCA610 imposes an input voltage limit for the filter. The expression VOA = GVI relates these two voltages. Thus, an output voltage limit VOAL constrains the input voltage to VI ≤ VOAL/G. C –s/nRC VO = 2 s + s/nRC + G/R2C2 VI 0.047µF nR nR fO = VI 5kΩ R 5kΩ BW = VO OPA620 330Ω 10 –(VC + 1) 2πRC 1 2πnRC Q = n10 –(VC + 1) AO = –1 C 0.047µF OPA620 R VOA VCA610 330Ω VC FIGURE 13. Adding the VCA610 to a State-Variable Filter Produces a Voltage-Controlled BandPass Filter With a Center Frequency Variable Over a 100:1 Range. ® VCA610 12