TPS61020, TPS61024 TPS61025, TPS61027 (3,25 mm x 3,25 mm) www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 96% EFFICIENT SYNCHRONOUS BOOST CONVERTER WITH 1.5-A SWITCH FEATURES • • • • • • • • • • • DESCRIPTION 96% Efficient Synchronous Boost Converter – 200-mA Output Current From 0.9-V Input – 500-mA Output Current From 1.8-V Input Output Voltage Remains Regulated When Input Voltage Exceeds Nominal Output Voltage Device Quiescent Current: 25-µA (Typ) Input Voltage Range: 0.9-V to 6.5-V Fixed and Adjustable Output Voltage Options Up to 5.5-V Power Save Mode for Improved Efficiency at Low Output Power Low Battery Comparator Low EMI-Converter (Integrated Antiringing Switch) Load Disconnect During Shutdown Over-Temperature Protection Small 3 mm x 3 mm QFN-10 Package The TPS6102x devices provide a power supply solution for products powered by either a one-cell, two-cell, or three-cell alkaline, NiCd or NiMH, or one-cell Li-Ion or Li-polymer battery. Output currents can go as high as 200 mA while using a single-cell alkaline, and discharge it down to 0.9 V. It can also be used for generating 5 V at 500 mA from a 3.3-V rail or a Li-Ion battery. The boost converter is based on a fixed frequency, pulse-width-modulation (PWM) controller using a synchronous rectifier to obtain maximum efficiency. At low load currents the converter enters the Power Save mode to maintain a high efficiency over a wide load current range. The Power Save mode can be disabled, forcing the converter to operate at a fixed switching frequency. The maximum peak current in the boost switch is limited to a value of 1500 mA. The TPS6102x devices keep the output voltage regulated even when the input voltage exceeds the nominal output voltage. The output voltage can be programmed by an external resistor divider, or is fixed internally on the chip. The converter can be disabled to minimize battery drain. During shutdown, the load is completely disconnected from the battery. A low-EMI mode is implemented to reduce ringing and, in effect, lower radiated electromagnetic energy when the converter enters the discontinuous conduction mode. The device is packaged in a 10-pin QFN PowerPAD™package measuring 3 mm x 3 mm (DRC). APPLICATIONS • • • • • • All One-Cell, Two-Cell and Three-Cell Alkaline, NiCd or NiMH or Single-Cell Li Battery Powered Products Portable Audio Players PDAs Cellular Phones Personal Medical Products Camera White LED Flash Light L1 6.8 µH SW VOUT VBAT 0.9-V To 6.5-V Input C1 10 µF R1 R3 EN C2 2.2 µF C3 47 µF VO 3.3 V Up To 200 mA FB LBI R4 R5 R2 PS GND LBO Low Battery Output PGND TPS61020 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2003–2004, Texas Instruments Incorporated TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. AVAILABLE OUTPUT VOLTAGE OPTIONS (1) TA OUTPUT VOLTAGE DC/DC PACKAGE MARKING Adjustable BDR 3.0 V BDS 40°C to 85°C (1) (2) 3.3 V BDT 5V BDU PART NUMBER (2) PACKAGE TPS61020DRC TPS61024DRC 10-Pin QFN TPS61025DRC TPS61027DRC Contact the factory to check availability of other fixed output voltage versions. The DRC package is available taped and reeled. Add R suffix to device type (e.g., TPS61020DRCR) to order quantities of 3000 devices per reel. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) TPS6102x Input voltage range on SW, VOUT, LBO, VBAT, PS, EN, FB, LBI -0.3 V to 7 V Operating virtual junction temperature range, TJ -40°C to 150°C Storage temperature range Tstg -65°C to 150°C (1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATINGS TABLE PACKAGE THERMAL RESISTANCE ΘJA POWER RATING TA≤ 25°C DERATING FACTOR ABOVE TA = 25°C DRC 48.7 °C/W 2054 mW 21 mW/°C RECOMMENDED OPERATING CONDITIONS MIN NOM MAX UNIT Supply voltage at VBAT, VI 0.9 6.5 V Operating free air temperature range, TA -40 85 °C Operating virtual junction temperature range, TJ -40 125 °C 2 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 ELECTRICAL CHARACTERISTICS over recommended free-air temperature range and over recommended input voltage range (typical at an ambient temperature range of 25°C) (unless otherwise noted) DC/DC STAGE PARAMETER VI TEST CONDITIONS Minimum input voltage range for start-up MIN RL = 120 Ω Input voltage range, after start-up TYP MAX 0.9 1.2 0.9 VO TPS61020 output voltage range 1.8 VFB TPS61020 feedback voltage 490 f Oscillator frequency ISW Switch current limit VOUT= 3.3 V UNIT V 6.5 500 5.5 V 510 mV 480 600 720 kHz 1200 1500 1800 mA Start-up current limit 0.4 x ISW mA SWN switch on resistance VOUT= 3.3 V 260 mΩ SWP switch on resistance VOUT= 3.3 V 290 mΩ Total accuracy (including line and load regulation) -3% 3% Line regulation 0.6% Load regulation Quiescent current 0.6% VBAT VOUT Shutdown current IO = 0 mA, VEN = VBAT = 1.2 V, VOUT = 3.3 V, TA = 25°C 1 3 µA 25 45 µA VEN = 0 V, VBAT = 1.2 V, TA = 25°C 0.1 1 µA MIN TYP MAX UNIT 490 500 510 mV CONTROL STAGE PARAMETER TEST CONDITIONS VUVLO Under voltage lockout threshold VLBI voltage decreasing VIL LBI voltage threshold VLBI voltage decreasing 0.8 LBI input hysteresis VOL V 10 mV LBI input current EN = VBAT or GND 0.01 0.1 LBO output low voltage VO = 3.3 V, IOI = 100 µA 0.04 0.4 LBO output low current Vlkg LBO output leakage current VIL EN, PS input low voltage VIH EN, PS input high voltage EN, PS input current 100 VLBO = 7 V 0.01 V µA 0.1 µA 0.2 × VBAT V 0.8 × VBAT Clamped on GND or VBAT µA V 0.01 0.1 µA Overtemperature protection 140 °C Overtemperature hysteresis 20 °C 3 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 PIN ASSIGNMENTS DRC PACKAGE (TOP VIEW) EN VOUT FB LBO GND PGND SW PS LBI VBAT Terminal Functions TERMINAL NAME NO. I/O DESCRIPTION EN 1 I Enable input. (1/VBAT enabled, 0/GND disabled) FB 3 I Voltage feedback of adjustable versions GND 5 LBI 7 I Low battery comparator input (comparator enabled with EN) LBO 4 O Low battery comparator output (open drain) PS 8 I Enable/disable power save mode (1 / VBAT disabled, 0/ GND enabled) SW 9 I Boost and rectifying switch input PGND 10 VBAT 6 I Supply voltage VOUT 2 O Boost converter output PowerPAD™ 4 Control / logic ground Power ground Must be soldered to achieve appropriate power dissipation. Should be connected to PGND. TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 ELECTRICAL CHARACTERISTICS (continued) FUNCTIONAL BLOCK DIAGRAM (TPS61020) SW Backgate Control AntiRinging VBAT VOUT 10 kΩ VOUT Vmax Control 20 pF Gate Control PGND PGND Regulator PGND Error Amplifier _ FB + Vref = 0.5 V Control Logic + _ GND Oscillator Temperature Control EN PS GND Low Battery Comparator _ LBI LDO + + _ Vref = 0.5 V GND 5 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 PARAMETER MEASUREMENT INFORMATION L1 6.8 µH VOUT SW VBAT Power Supply C1 10 µF R1 R3 EN C2 2.2 µF C3 47 µF VCC Boost Output FB LBI R4 R5 R2 PS LBO GND List of Components: U1 = TPS6102xDRC L1 = EPCOS B82462−G4682 C1, C2 = X7R/X5R Ceramic C3 = Low ESR Tantalum Control Output PGND TPS6102x TYPICAL CHARACTERISTICS Table of Graphs FIGURE Maximum output current vs Input voltage 1 vs Output current (TPS61020) 2 vs Output current (TPS61025) 3 vs Output current (TPS61027) 4 vs Input voltage (TPS61025) 5 vs Input voltage (TPS61027) 6 vs Output current (TPS61025) 7 vs Output current (TPS61027) 8 No load supply current into VBAT vs Input voltage 9 No load supply current into VOUT vs Input voltage 10 Output voltage in continuous mode (TPS61025) 11 Output voltage in continuous mode (TPS61027) 12 Output voltage in power save mode (TPS61025) 13 Output voltage in power save mode (TPS61027) 14 Load transient response (TPS61025) 15 Load transient response (TPS61027) 16 Line transient response (TPS61025) 17 Line transient response (TPS61027) 18 Start-up after enable (TPS61025) 19 Start-up after enable (TPS61027) 20 Efficiency Output voltage Waveforms 6 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 TYPICAL CHARACTERISTICS TPS61020 EFFICIENCY vs OUTPUT CURRENT MAXIMUM OUTPUT CURRENT vs INPUT VOLTAGE 1400 100 VO = 3.3 V VO = 5 V 80 1000 70 Efficiency - % Maximum Output Current - mA 1200 800 600 VO = 1.8 V 400 VO = 1.8 V VBAT = 0.9 V 90 VBAT = 1.8 V 60 50 40 30 20 200 10 0 0.9 1.7 2.5 3.3 4.1 4.9 VI - Input Voltage - V 5.7 0 6.5 1 Figure 1. Figure 2. TPS61025 EFFICIENCY vs OUTPUT CURRENT TPS61027 EFFICIENCY vs OUTPUT CURRENT 100 100 90 90 80 1000 80 VBAT = 2.4 V 70 60 50 VBAT = 0.9 V 40 40 20 20 1 10 100 IO - Output Current - mA Figure 3. VO = 5 V 10 0 1000 VBAT = 3.6 V 50 30 VO = 3.3 V VBAT = 2.4 V VBAT = 1.8 V 60 30 10 VBAT = 1.2 V 70 VBAT = 1.8 V Efficiency - % Efficiency - % 10 100 IO - Output Current - mA 0 1 10 100 IO - Output Current - mA 1000 Figure 4. 7 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 TYPICAL CHARACTERISTICS (continued) TPS61025 EFFICIENCY vs INPUT VOLTAGE TPS61027 EFFICIENCY vs INPUT VOLTAGE 100 IO = 100 mA 90 90 85 85 IO = 10 mA 80 75 IO = 250 mA 70 70 60 60 55 55 1.9 2.4 2.9 3.4 3.9 IO = 250 mA 75 65 1.4 IO = 10 mA 80 65 50 0.9 IO = 100 mA 95 Efficiency - % Efficiency - % 95 100 VO = 3.3 V VO = 5 V 50 4.4 4.9 0.9 1.4 1.9 2.4 2.9 3.4 3.9 4.4 4.9 5.4 5.9 6.4 VI - Input Voltage - V VI - Input Voltage - V Figure 5. Figure 6. TPS61025 OUTPUT VOLTAGE vs OUTPUT CURRENT TPS61027 OUTPUT VOLTAGE vs OUTPUT CURRENT 3.35 5.10 VO = 3.3 V VO = 5 V VO - Output Voltage - V VO - Output Voltage - V 5.05 3.30 VBAT = 2.4 V 3.25 5 VBAT = 3.6 V 4.95 4.90 4.85 3.20 4.80 1 8 10 100 1000 IO - Output Current - mA 1 10 100 IO - Output Current - mA Figure 7. Figure 8. 1000 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 TYPICAL CHARACTERISTICS (continued) NO LOAD SUPPLY CURRENT INTO VBAT vs INPUT VOLTAGE NO LOAD SUPPLY CURRENT INTO VOUT vs INPUT VOLTAGE 1.6 34.8 TA = 85°C No Load Supply Current Into VOUT - µ A 1.4 1.2 1 0.8 TA = 25°C TA = -40°C 0.6 0.4 0.2 29.8 24.8 TA = -40°C TA = 25°C 19.8 14.8 9.8 4.8 -0.2 2 2.5 3 3.5 4 4.5 5 VI - Input Voltage - V 5.5 6 0.9 1.5 6.5 2 2.5 3 3.5 4 4.5 5 VI - Input Voltage - V 5.5 6 Figure 9. Figure 10. TPS61025 OUTPUT VOLTAGE IN CONTINUOUS MODE TPS61027 OUTPUT VOLTAGE IN CONTINUOUS MODE 6.5 Output Voltage 20 mV/div VI = 1.2 V, RL = 33 Ω, VO = 3.3 V Inductor Current 200 mA/div Output Voltage 20 mV/div 0 0.9 1.5 Inductor Current 200 mA/div No Load Supply Current Into VBAT - µ A TA = 85°C t - Time - 1 µs/div Figure 11. VI = 3.6 V, RL = 25 Ω, VO = 5 V t - Time - 1 µs/div Figure 12. 9 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 TYPICAL CHARACTERISTICS (continued) Inductor Current 200 mA/div, DC Figure 13. Figure 14. TPS61025 LOAD TRANSIENT RESPONSE TPS61027 LOAD TRANSIENT RESPONSE VI = 3.6 V, IL = 100 mA to 200 mA, VO = 5 V Output Voltage 20 mV/div, AC VI = 1.2 V, IL = 100 mA to 200 mA, VO = 3.3 V Output Current 100 mA/div, DC t - Time - 50 µs/div Output Voltage 20 mV/div, AC Output Current 100 mA/div, DC VI = 3.6 V, RL = 250 Ω, VO = 5 V t - Time - 50 µs/div t - Time - 2 ms/div Figure 15. 10 TPS61027 OUTPUT VOLTAGE IN POWER SAVE MODE Output Voltage 50 mV/div, AC VI = 1.2 V, RL = 330 Ω, VO = 3.3 V Inductor Current 100 mA/div, DC Output Voltage 20 mV/div, AC TPS61025 OUTPUT VOLTAGE IN POWER SAVE MODE t - Time - 2 ms/div Figure 16. TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 TYPICAL CHARACTERISTICS (continued) TPS61027 LINE TRANSIENT RESPONSE VI = 3 V to 3.6 V, RL = 25 Ω, VO = 5 V Output Voltage 20 mV/div, AC Input Voltage 500 mV/div, AC VI = 1.8 V to 2.4 V, RL = 33 Ω, VO = 3.3 V Output Voltage 20 mV/div, AC t - Time - 2 ms/div t - Time - 2 ms/div Figure 18. TPS61025 START-UP AFTER ENABLE TPS61027 START-UP AFTER ENABLE t - Time - 1 ms/div Figure 19. VI = 3.6 V, RL = 50 Ω, VO = 5 V Inductor Current 500 mA/div, DC Voltage At SW 2 V/div, DC Inductor Current 200 mA/div, DC Output Voltage 1 V/div, DC VI = 2.4V, RL = 33 Ω, VO = 3.3 V Output Voltage 2 V/div, DC Enable 5 V/div, DC Enable 5 V/div, DC Figure 17. Voltage At SW 2 V/div, DC Input Voltage 500 mV/div, AC TPS61025 LINE TRANSIENT RESPONSE t - Time - 500 µs/div Figure 20. 11 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 DETAILED DESCRIPTION CONTROLLER CIRCUIT The controller circuit of the device is based on a fixed frequency multiple feedforward controller topology. Input voltage, output voltage, and voltage drop on the NMOS switch are monitored and forwarded to the regulator. So changes in the operating conditions of the converter directly affect the duty cycle and must not take the indirect and slow way through the control loop and the error amplifier. The control loop, determined by the error amplifier, only has to handle small signal errors. The input for it is the feedback voltage on the FB pin or, at fixed output voltage versions, the voltage on the internal resistor divider. It is compared with the internal reference voltage to generate an accurate and stable output voltage. The peak current of the NMOS switch is also sensed to limit the maximum current flowing through the switch and the inductor. The typical peak current limit is set to 1500 mA. An internal temperature sensor prevents the device from getting overheated in case of excessive power dissipation. Synchronous Rectifier The device integrates an N-channel and a P-channel MOSFET transistor to realize a synchronous rectifier. Because the commonly used discrete Schottky rectifier is replaced with a low RDS(ON) PMOS switch, the power conversion efficiency reaches 96%. To avoid ground shift due to the high currents in the NMOS switch, two separate ground pins are used. The reference for all control functions is the GND pin. The source of the NMOS switch is connected to PGND. Both grounds must be connected on the PCB at only one point close to the GND pin. A special circuit is applied to disconnect the load from the input during shutdown of the converter. In conventional synchronous rectifier circuits, the backgate diode of the high-side PMOS is forward biased in shutdown and allows current flowing from the battery to the output. This device however uses a special circuit which takes the cathode of the backgate diode of the high-side PMOS and disconnects it from the source when the regulator is not enabled (EN = low). The benefit of this feature for the system design engineer is that the battery is not depleted during shutdown of the converter. No additional components have to be added to the design to make sure that the battery is disconnected from the output of the converter. Down Regulation In general, a boost converter only regulates output voltages which are higher than the input voltage. This device operates differently. For example, it is able to regulate 3.0 V at the output with two fresh alkaline cells at the input having a total cell voltage of 3.2 V. Another example is powering white LEDs with a forward voltage of 3.6 V from a fully charged Li-Ion cell with an output voltage of 4.2 V. To control these applications properly, a down conversion mode is implemented. If the input voltage reaches or exceeds the output voltage, the converter changes to a down conversion mode. In this mode, the control circuit changes the behavior of the rectifying PMOS. It sets the voltage drop across the PMOS as high as needed to regulate the output voltage. This means the power losses in the converter increase. This has to be taken into account for thermal consideration. Device Enable The device is put into operation when EN is set high. It is put into a shutdown mode when EN is set to GND. In shutdown mode, the regulator stops switching, all internal control circuitry including the low-battery comparator is switched off, and the load is isolated from the input (as described in the Synchronous Rectifier Section). This also means that the output voltage can drop below the input voltage during shutdown. During start-up of the converter, the duty cycle and the peak current are limited in order to avoid high peak currents drawn from the battery. Undervoltage Lockout An undervoltage lockout function prevents device start-up if the supply voltage on VBAT is lower than approximately 0.8 V. When in operation and the battery is being discharged, the device automatically enters the shutdown mode if the voltage on VBAT drops below approximately 0.8 V. This undervoltage lockout function is implemented in order to prevent the malfunctioning of the converter. 12 www.ti.com TPS61020, TPS61024 TPS61025, TPS61027 SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 DETAILED DESCRIPTION (continued) Softstart When the device enables, the internal startup cycle starts with the first step, the precharge phase. During precharge, the rectifying switch is turned on until the output capacitor is charged to a value close to the input voltage. The rectifying switch is current limited during that phase. This also limits the output current under short circuit conditions at the output. After charging the output capacitor to the input voltage, the device starts switching. If the input voltage is below 1.4 V the device works with a fixed duty cycle of 50% until the output voltage reaches 1.4V. After that the duty cycle is set depending on the input output voltage ratio. Until the output voltage reaches its nominal value, the boost switch current limit is set to 40% of its nominal value to avoid high peak currents at the battery during startup. As soon as the output voltage is reached, the regulator takes control and the switch current limit is set back to 100%. Power Save Mode The PS pin can be used to select different operation modes. To enable power save, PS must be set low. Power save mode is used to improve efficiency at light load. In power save mode the converter only operates when the output voltage trips below a set threshold voltage. It ramps up the output voltage with one or several pulses and goes again into power save mode once the output voltage exceeds the set threshold voltage. This power save mode can be disabled by setting the PS to VBAT. In down conversion mode, power save mode is always active and the device cannot be forced into fixed frequency operation at light loads. Low Battery Detector Circuit—LBI/LBO The low-battery detector circuit is typically used to supervise the battery voltage and to generate an error flag when the battery voltage drops below a user-set threshold voltage. The function is active only when the device is enabled. When the device is disabled, the LBO pin is high-impedance. The switching threshold is 500 mV at LBI. During normal operation, LBO stays at high impedance when the voltage, applied at LBI, is above the threshold. It is active low when the voltage at LBI goes below 500 mV. The battery voltage, at which the detection circuit switches, can be programmed with a resistive divider connected to the LBI pin. The resistive divider scales down the battery voltage to a voltage level of 500 mV, which is then compared to the LBI threshold voltage. The LBI pin has a built-in hysteresis of 10 mV. See the application section for more details about the programming of the LBI threshold. If the low-battery detection circuit is not used, the LBI pin should be connected to GND (or to VBAT) and the LBO pin can be left unconnected. Do not let the LBI pin float. Low-EMI Switch The device integrates a circuit that removes the ringing that typically appears on the SW node when the converter enters discontinuous current mode. In this case, the current through the inductor ramps to zero and the rectifying PMOS switch is turned off to prevent a reverse current flowing from the output capacitors back to the battery. Due to the remaining energy that is stored in parasitic components of the semiconductor and the inductor, a ringing on the SW pin is induced. The integrated antiringing switch clamps this voltage to VBAT and therefore dampens ringing. 13 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 APPLICATION INFORMATION DESIGN PROCEDURE The TPS6102x dc/dc converters are intended for systems powered by a single up to triple cell Alkaline, NiCd, NiMH battery with a typical terminal voltage between 0.9 V and 6.5 V. They can also be used in systems powered by one-cell Li-Ion or Li-Polymer with a typical voltage between 2.5 V and 4.2 V. Additionally, any other voltage source with a typical output voltage between 0.9 V and 6.5 V can power systems where the TPS6102x is used. Programming the Output Voltage The output voltage of the TPS61020 dc/dc converter can be adjusted with an external resistor divider. The typical value of the voltage at the FB pin is 500 mV. The maximum recommended value for the output voltage is 5.5 V. The current through the resistive divider should be about 100 times greater than the current into the FB pin. The typical current into the FB pin is 0.01 µA, and the voltage across R4 is typically 500 mV. Based on those two values, the recommended value for R4 should be lower than 500 kΩ, in order to set the divider current at 1 µA or higher. Because of internal compensation circuitry the value for this resistor should be in the range of 200 kΩ. From that, the value of resistor R3, depending on the needed output voltage (VO), can be calculated using Equation 1: R3 R4 V O 1 V FB 180 k V O 1 500 mV (1) If as an example, an output voltage of 3.3 V is needed, a 1.0-MΩ resistor should be chosen for R3. If for any reason the value for R4 is chosen significantly lower than 200kΩ additional capacitance in parallel to R3 is recommended, in case the device shows instable regulation of the output voltage. The required capacitance value can be easily calculated using Equation 2: C 20 pF 200 k 1 parR3 R4 (2) L1 VOUT SW C2 VBAT Power Supply C1 R1 C3 VCC Boost Output R3 EN FB LBI R4 R5 R2 PS LBO GND Control Output PGND TPS61020 Figure 21. Typical Application Circuit for Adjustable Output Voltage Option Programming the LBI/LBO Threshold Voltage The current through the resistive divider should be about 100 times greater than the current into the LBI pin. The typical current into the LBI pin is 0.01 µA, and the voltage across R2 is equal to the LBI voltage threshold that is generated on-chip, which has a value of 500 mV. The recommended value for R2 is therefore in the range of 500 kΩ. From that, the value of resistor R1, depending on the desired minimum battery voltage VBAT, can be calculated using Equation 3. 14 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com R1 R2 SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 V V BAT LBIthreshold 1 390 k V BAT 1 500 mV (3) The output of the low battery supervisor is a simple open-drain output that goes active low if the dedicated battery voltage drops below the programmed threshold voltage on LBI. The output requires a pullup resistor with a recommended value of 1 MΩ. If not used, the LBO pin can be left floating or tied to GND. Inductor Selection A boost converter normally requires two main passive components for storing energy during the conversion. A boost inductor and a storage capacitor at the output are required. To select the boost inductor, it is recommended to keep the possible peak inductor current below the current limit threshold of the power switch in the chosen configuration. For example, the current limit threshold of the TPS6102x's switch is 1800 mA at an output voltage of 5 V. The highest peak current through the inductor and the switch depends on the output load, the input (VBAT), and the output voltage (VOUT). Estimation of the maximum average inductor current can be done using Equation 4: V OUT I I L OUT V 0.8 BAT (4) For example, for an output current of 200 mA at 3.3 V, at least 920 mA of average current flows through the inductor at a minimum input voltage of 0.9 V. The second parameter for choosing the inductor is the desired current ripple in the inductor. Normally, it is advisable to work with a ripple of less than 20% of the average inductor current. A smaller ripple reduces the magnetic hysteresis losses in the inductor, as well as output voltage ripple and EMI. But in the same way, regulation time at load changes rises. In addition, a larger inductor increases the total system costs. With those parameters, it is possible to calculate the value for the inductor by using Equation 5: V V –V BAT OUT BAT L I ƒ V L OUT (5) Parameter f is the switching frequency and∆ IL is the ripple current in the inductor, i.e., 20% × IL. In this example, the desired inductor has the value of 5.5 µH. With this calculated value and the calculated currents, it is possible to choose a suitable inductor. In typical applications a 6.8 µH inductance is recommended. The device has been optimized to operate with inductance values between 2.2 µH and 22 µH. Nevertheless operation with higher inductance values may be possible in some applications. Detailed stability analysis is then recommended. Care has to be taken that load transients and losses in the circuit can lead to higher currents as estimated in Equation 5. Also, the losses in the inductor caused by magnetic hysteresis losses and copper losses are a major parameter for total circuit efficiency. The following inductor series from different suppliers have been used with the TPS6102x converters: Table 1. List of Inductors VENDOR Sumida Wurth Elektronik EPCOS Cooper Electronics Technologies INDUCTOR SERIES CDRH4D28 CDRH5D28 7447789 744042 B82462-G4 SD25 SD20 15 TPS61020, TPS61024 TPS61025, TPS61027 SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 www.ti.com Capacitor Selection Input Capacitor At least a 10-µF input capacitor is recommended to improve transient behavior of the regulator and EMI behavior of the total power supply circuit. A ceramic capacitor or a tantalum capacitor with a 100-nF ceramic capacitor in parallel, placed close to the IC, is recommended. Output Capacitor The major parameter necessary to define the output capacitor is the maximum allowed output voltage ripple of the converter. This ripple is determined by two parameters of the capacitor, the capacitance and the ESR. It is possible to calculate the minimum capacitance needed for the defined ripple, supposing that the ESR is zero, by using Equation 6: I C min V V OUT OUT BAT ƒ V V OUT (6) Parameter f is the switching frequency and ∆V is the maximum allowed ripple. With a chosen ripple voltage of 10 mV, a minimum capacitance of 24 µF is needed. The total ripple is larger due to the ESR of the output capacitor. This additional component of the ripple can be calculated using Equation 7: V I R ESR OUT ESR (7) An additional ripple of 16 mV is the result of using a tantalum capacitor with a low ESR of 80 mΩ. The total ripple is the sum of the ripple caused by the capacitance and the ripple caused by the ESR of the capacitor. In this example, the total ripple is 26 mV. Additional ripple is caused by load transients. This means that the output capacitor has to completely supply the load during the charging phase of the inductor. A reasonable value of the output capacitance depends on the speed of the load transients and the load current during the load change. With the calculated minimum value of 24 µF and load transient considerations the recommended output capacitance value is in a 47 to 100 µF range. For economical reasons, this is usually a tantalum capacitor. Therefore, the control loop has been optimized for using output capacitors with an ESR of above 30 mΩ. The minimum value for the output capacitor is 10 µF. Small Signal Stability When using output capacitors with lower ESR, like ceramics, the adjustable voltage version is recommended. The missing ESR can be compensated in the feedback divider. Typically a capacitor in the range of 4.7 pF in parallel to R3 helps to obtain small signal stability with lowest ESR output capacitors. For more detailed analysis, the small signal transfer function of the error amplifier and the regulator, which is given in Equation 8, can be used: 4 (R3 R4) A d REG V R4 (1 i 0.9 s) FB (8) Layout Considerations As for all switching power supplies, the layout is an important step in the design, especially at high peak currents and high switching frequencies. If the layout is not carefully done, the regulator could show stability problems as well as EMI problems. Therefore, use wide and short traces for the main current path and for the power ground tracks. The input capacitor, output capacitor, and the inductor should be placed as close as possible to the IC. Use a common ground node for power ground and a different one for control ground to minimize the effects of ground noise. Connect these ground nodes at any place close to one of the ground pins of the IC. The feedback divider should be placed as close as possible to the control ground pin of the IC. To lay out the control ground, it is recommended to use short traces as well, separated from the power ground traces. This avoids ground shift problems, which can occur due to superimposition of power ground current and control ground current. 16 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 APPLICATION EXAMPLES L1 6.8 µH Battery Input SW VOUT C2 2.2 µF VBAT R1 C1 10 µF EN C3 100 µF VCC 5 V Boost Output FB R5 LBI R2 PS GND LBO LBO PGND TPS61027 List of Components: U1 = TPS61027DRC L1 = EPCOS B82462-G4682 C1, C2 = X7R,X5R Ceramic C3 = Low ESR Tantalum Figure 22. Power Supply Solution for Maximum Output Power Operating from a Single Alkaline Cell L1 6.8 µH Battery Input SW VOUT C2 2.2 µF VBAT C1 10 µF R1 EN C3 47 µF VCC 5 V Boost Output FB R5 LBI R2 PS GND LBO LBO PGND TPS61027 List of Components: U1 = TPS61027DRC L1 = EPCOS B82462-G4682 C1, C2 = X7R,X5R Ceramic C3 = Low ESR Tantalum Figure 23. Power Supply Solution for Maximum Output Power Operating from a Dual/Triple Alkaline Cell or Single Li-Ion Cell 17 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 TPS61020 L1 SW 4.7 µH D1 C2 VBAT C1 Input 0.9 V to 6.5 V VOUT 10 µF 22 µF EN LBI FB PS LBO C3 R1 10 nF PGND GND List of Components: U1 = TPS61020DRC L1 = Sumida CDRH2D16-4R7 C1, C2, C3 = X7R, X5R Ceramic D1 = White LED Figure 24. Power Supply Solution for Powering White LED´s in Lighting Applications TPS61020 L1 4.7 µH Input 1.8 V to 6.5 V C1 10 µF SW VOUT C2 VBAT 22 µF EN LBI FB PS LBO GND C3 22 µF R1 1.5 MΩ D1 R2 200 kΩ PGND Flashlight Comtrol Q1 List of Components: U1 = TPS61020DRC L1 = TDK VLF3010AT 4R7MR70 C1, C2, C3 = X7R, X5R Ceramic D1 = OSRAM LWW57G Q1 = Vishay SI1012R Figure 25. Simple Power Supply Solution for Powering White LED Flashlights 18 TPS61020, TPS61024 TPS61025, TPS61027 www.ti.com SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 C5 VCC2 10 V Unregulated Auxiliary Output DS1 C6 1 µF 0.1 µF L1 6.8 µH Battery Input SW C2 2.2 µF VBAT R1 C1 10 µF VCC1 5 V Boost Main Output VOUT C3 47 µF EN R5 FB LBI R2 PS LBO LBO PGND GND List of Components: U1 = TPS61027DRC1 L1 = EPCOS B82462-G4682 C3, C5, C6, = X7R,X5R Ceramic C3 = Low ESR Tantalum DS1 = BAT54S TPS61027 Figure 26. Power Supply Solution With Auxiliary Positive Output Voltage C5 DS1 VCC2 -5 V Unregulated Auxiliary Output C6 1 µF 0.1 µF L1 6.8 µH Battery Input SW C2 2.2 µF VBAT C1 10 µF R1 VCC1 5 V Boost Main Output VOUT C3 47 µF EN FB LBI R5 R2 PS GND List of Components: U1 = TPS61027DRC L1 = EPCOS B82462-G4682 C1, C2, C5, C6 = X7R,X5R Ceramic C3 = Low ESR Tantalum DS1 = BAT54S LBO LBO PGND TPS61027 Figure 27. Power Supply Solution With Auxiliary Negative Output Voltage 19 TPS61020, TPS61024 TPS61025, TPS61027 SLVS451A – SEPTEMBER 2003 – REVISED APRIL 2004 www.ti.com THERMAL INFORMATION Implementation of integrated circuits in low-profile and fine-pitch surface-mount packages typically requires special attention to power dissipation. Many system-dependent issues such as thermal coupling, airflow, added heat sinks and convection surfaces, and the presence of other heat-generating components affect the power-dissipation limits of a given component. Three basic approaches for enhancing thermal performance are listed below. • Improving the power dissipation capability of the PCB design • Improving the thermal coupling of the component to the PCB • Introducing airflow in the system The maximum recommended junction temperature (TJ) of the TPS6102x devices is 125°C. The thermal resistance of the 10-pin QFN 3 x 3 package (DRC) is RΘJA = 48.7 °C/W, if the PowerPAD is soldered. Specified regulator operation is assured to a maximum ambient temperature TA of 85°C. Therefore, the maximum power dissipation is about 820 mW. More power can be dissipated if the maximum ambient temperature of the application is lower. T T J(MAX) A P 125°C 85°C 820 mW D(MAX) R 48.7 °CW JA (9) 20 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All products are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its hardware products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. 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