LTC1473L Dual Low Voltage PowerPathTM Switch Driver U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ The LTC®1473L provides reliable and efficient switching between two DC power sources. This device drives two external sets of back-to-back N-channel MOSFET switches to route power to the input of a low voltage system. An internal boost regulator provides the voltage to fully enhance the logic-level N-channel MOSFET switches while an internal undervoltage lock-out circuit keeps the system alive down to 2.8V. Power Path Management for Systems with Multiple DC Sources Switches and Isolates Sources from 3.3V to 10V All N-Channel Switching to Reduce Power Losses and System Cost Built-In Step-Up Regulator for N-Channel Gate Drive Capacitor Inrush and Short-Circuit Current Limited User-Programmable Timer Prevents Overdissipation During Current Limiting Undervoltage Lockout Prevents Operation with Low Inputs Small Footprint: 16-Pin Narrow SSOP The LTC1473L senses current to limit inrush between the batteries and the system supply capacitor during switchover transitions or during fault conditions. A user-programmable timer monitors the time the MOSFET switches are in current limit and latches them off when the programmed time is exceeded. U APPLICATIO S ■ ■ ■ ■ A unique “2-diode” logic mode ensures system start-up regardless of which input receives power first. Portable Computers Portable Instruments Fault Tolerant Computers Battery-Backup Systems 3.3V/5V Power Management , LTC and LT are registered trademarks of Linear Technology Corporation. PowerPath is a trademark of Linear Technology Corporation. U ■ TYPICAL APPLICATIO 3.3V to 4-Cell NiMH Backup Switch Si9926DY DCIN 3.3V 1 BAT54C LOGIC DRIVEN VBAT1 4× NiMH 2 3 4 5 6 CTIMER 2000pF 1mH* 1µF 1µF 7 8 LTC1473L IN1 GA1 IN2 SAB1 16 15 14 DIODE GB1 TIMER 13 SENSE + V+ VGG SW GND RSENSE 0.04Ω 3.3V OR VBAT1 + 12 SENSE – 11 GA2 10 SAB2 9 GB2 COUT 1473 TA01 * COILCRAFT 1812LS-105XKBC Si9926DY 1 LTC1473L U W U PACKAGE/ORDER I FOR ATIO U W W W ABSOLUTE AXI U RATI GS (Note 1) TOP VIEW SENSE +, SENSE –, V + .................................. – 0.3 to 10V GA1, GB1, GA2, GB2 ................................... – 0.3 to 20V SAB1, SAB2 ................................................. – 0.3 to 10V SW, VGG ...................................................... – 0.3 to 20V IN1, IN2, DIODE ...........................................– 0.3V to 7V Junction Temperature (Note 2) ............................. 125°C Operating Temperature Range ..................... 0°C to 70°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C IN1 1 16 GA1 IN2 2 15 SAB1 ORDER PART NUMBER DIODE 3 14 GB1 TIMER 4 13 SENSE + V+ 5 12 SENSE – VGG 6 11 GA2 SW 7 10 SAB2 GND 8 9 LTC1473LCGN GN PART MARKING GB2 1473L GN PACKAGE 16-LEAD NARROW PLASTIC SSOP TJMAX = 125°C, θJA = 150°C/ W Consult factory for Military and Industrial grade parts. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Test circuit, V+ = 5V, unless otherwise specified. SYMBOL PARAMETER V+ Supply Operating Range CONDITIONS MIN TYP 2.8 MAX UNITS 9 V IS Supply Current VIN1 = VDIODE = 5V, VIN2 = 0V, VSENSE = VSENSE = 5V ● 100 200 µA VGS VGS Gate Supply Voltage VGS = VGG – V +, 2.8V ≤ V + ≤ 10V (Note 3) ● 7.5 8.5 9.5 V V+ UVLO V + Undervoltage Lockout Threshold V + Ramping ● 2.3 2.5 2.8 V UVLOHYS V + Undervoltage Lockout Hysteresis 70 mV 2 0.9 V V+ + – Down VHIDIGIN Digital Input Logic High (Note 4) ● VLODIGIN Digital Input Logic Low (Note 4) ● IIN Input Current VIN1 = VIN2 = VDIODE = 5V VGS(ON) Gate-to-Source ON Voltage IGA1 = IGA2 = IGB1 = IGB2 = – 1µA, VSAB1 = VSAB2 = 5V ● VGS(OFF) Gate-to-Source OFF Voltage IGA1 = IGA2 = IGB1 = IGB2 = 100µA, VSAB1 = VSAB2 = 5V ● 4.5 0.6 0.4 V ±1 µA 5.6 7.0 V 0 0.4 V Input Bias Current VSENSE VSENSE 10V (Note 3) VSENSE + = VSENSE – = 0V (Note 5) ● ● 2 – 300 4.5 – 175 10 – 75 µA µA IBSENSE – SENSE – Input Bias Current VSENSE + = VSENSE – = 10V (Note 3) VSENSE + = VSENSE – = 0V (Note 5) ● ● 2 – 300 4.5 – 175 10 – 75 µA µA VSENSE Inrush Current Limit Sense Voltage VSENSE – = 10V (VSENSE + – VSENSE –) (Note 3) VSENSE – = 0V (VSENSE + – VSENSE –) 0.15 0.10 0.20 0.20 0.25 0.30 V V IPDSAB SAB1, SAB2 Pull-Down Current VIN1 = VIN2 = VDIODE = 0.4V, V + = 10V (Note 3) VIN1 = VIN2 = 0.4V, VDIODE = 2V 5 30 20 140 35 300 µA µA ITIMER Timer Source Current VIN1 = 0.4V, VIN2 = VDIODE = 2V, VTIMER = 0V, VSENSE + – VSENSE – = 300mV ● 3 6 9 µA VTIMER Timer Latch Threshold Voltage VIN1 = 0.4V, VIN2 = VDIODE = 2V ● 1.05 1.16 1.25 t ON Gate Drive Rise Time CGS = 1000pF, VSAB1 = VSAB2 = 0V (Note 6) 33 µs t OFF Gate Drive Fall Time CGS = 1000pF, VSAB1 = VSAB2 = 5V (Note 6) 2 µs t D1 Gate Drive Turn-On Delay CGS = 1000pF, VSAB1 = VSAB2 = 0V (Note 6) 22 µs t D2 Gate Drive Turn-Off Delay CGS = 1000pF, VSAB1 = VSAB2 = 5V (Note 6) 1 µs fOVGG VGS Regulator Operating Frequency 30 kHz IBSENSE 2 + SENSE + += –= V LTC1473L ELECTRICAL CHARACTERISTICS Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD)(150°C/W) Note 3: Some tests are performed under more stringent conditions to ensure reliable operation over the entire supply voltage range. Note 4: Digital inputs include: IN1, IN2 and DIODE. Note 5: IS increases by the same amount as IBSENSE+ + IBSENSE– when their common mode falls below 5V. Note 6: Gate turn-on and turn-off times are measured with no inrush current limiting, i.e., VSENSE = 0V. Gate rise times are measured from 1V to 4.5V and fall times are measured from 4.5V to 1V. Delay times are measured from the input transition to when the gate voltage has risen or fallen to 3V. Results are not tested, but guaranteed by design. U W TYPICAL PERFOR A CE CHARACTERISTICS DC Supply Current vs Supply Voltage 250 DC Supply Current vs Temperature 140 VSENSE + = VSENSE – = V + DC Supply Current vs VSENSE 400 V + = 5V 130 100 VDIODE = 5V VIN1 = VIN2 = 0V 50 0 0 1 2 3 4 5 6 7 8 9 VDIODE = VIN1 = 5V VIN2 = 0V 100 90 80 0 – 25 25 50 0 TEMPERATURE (°C) 5.7 5.6 5.5 5.4 5.3 5.2 5.1 – 60 – 40 –20 0 20 40 60 TEMPERATURE (°C) 80 100 1473 G04 75 0 100 1 2 3 4 5 6 7 8 VSENSE COMMON MODE (V) Undervoltage Lockout Threshold (V +) vs Temperature 2.75 9.0 8.9 2.60 START-UP THRESHOLD 2.55 2.50 2.45 SHUTDOWN THRESHOLD 2.40 2.35 V + = 5V VGS = VGG – V + 8.8 8.7 8.6 8.5 8.4 8.3 8.2 2.30 2.25 – 60 –40 –20 0 20 40 60 TEMPERATURE (°C) 10 VGS Gate Supply Voltage vs Temperature 2.70 2.65 9 1473 G03 1473 G02 UNDERVOLTAGE LOCKOUT THRESHOLD (V) VGS GATE-TO-SOURCE ON VOLTAGE (V) 5.8 150 50 VGS Gate-to-Source ON Voltage vs Temperature 5.9 200 60 1473 G01 V + = VSAB = 10V 250 100 SUPPLY VOLTAGE (V) 6.0 300 70 50 – 50 10 SUPPLY CURRENT (µA) 150 120 110 VGS GATE SUPPLY VOLTAGE (V) VDIODE = VIN1 = 5V VIN2 = 0V SUPPLY CURRENT (µA) SUPPLY CURRENT (µA) 200 V+ = 5V VDIODE = VIN1 = 5V VIN2 = 0V VSENSE+ – VSENSE– = 0V 350 80 100 1473 G05 8.1 20 40 60 – 60 – 40 – 20 0 TEMPERATURE (°C) 80 100 1473 G06 3 LTC1473L U W TYPICAL PERFOR A CE CHARACTERISTICS 2.0 1.8 45 V + = 5V CLOAD = 1000pF VSAB = 5V GATE FALL TIME 1.6 1.4 1.2 TURN-OFF DELAY 1.0 0.8 0.6 0.4 20 40 60 –60 –40 – 20 0 TEMPERATURE (°C) 80 V + = 5V CLOAD = 1000pF VSAB = 0V 40 35 40 35 GATE RISE TIME 30 TURN-ON DELAY 25 20 15 10 10 FALL TIME VSAB = 5V 10 100 100 1000 GATE CAPACITIVE LOADING (pF) TIMER LATCH THRESHOLD VOLTAGE (V) 1.28 1.6 1.4 V + = 10V 1.0 V + = 2.8V 0.6 0.4 0.2 80 V + = 5V 1.26 1.24 1.22 1.20 1.18 1.16 1.14 1.12 1.10 – 50 100 – 25 25 50 75 0 TEMPERATURE (°C) 100 1473 G10 SENSE Pin Source Current (IBSENSE) vs VSENSE 300 V + = 5V TIMER = 0V 7.5 7.0 6.5 6.0 5.5 5.0 150 100 50 – 50 – 25 25 50 75 0 TEMPERATURE (°C) 100 125 1473 G12 4 200 0 4.5 4.0 – 50 V+ = 5V VDIODE = VIN1 = 5V VIN2 = 0V VSENSE+ – VSENSE– = 0V 250 SENSE PIN CURRENT (µA) TIMER SOURCE CURRENT (µA) 8.0 125 1473 G11 Timer Source Current vs Temperature 8.5 10000 1473 G08 Timer Latch Threshold Voltage vs Temperature 2.0 0 – 60 –40 –20 0 20 40 60 TEMPERATURE (°C) 15 1473 G08 1.8 0.8 20 0 80 Logic Input Threshold Voltage vs Temperature 1.2 25 5 0 20 40 60 – 60 – 40 – 20 0 TEMPERATURE (°C) 100 RISE TIME VSAB = 0V 30 5 1473 G07 INPUT THRESHOLD VOLTAGE (V) Rise and Fall Time vs Gate Capacitive Loading RISE AND FALL TIME (µs) 2.2 Turn-On Delay and Gate Rise Time vs Temperature TURN-ON DELAY AND GATE RISE TIME (µs) TURN-OFF DELAY AND GATE FALL TIME (µs) Turn-Off Delay and Gate Fall Time vs Temperature 0 1 2 3 4 5 6 VSENSE (V) 7 8 9 10 1473 G13 LTC1473L U U U PI FU CTIO S IN1 (Pin 1): Logic Input of Gate Drivers GA1 and GB1. IN1 is disabled when IN2 is high or DIODE is low. During 2-diode mode, asserting IN1 disables the fault timer function. SW (Pin 7): Open Drain of an Internal N-Channel MOSFET Switch. This pin drives the bottom of the VGG switching regulator inductor which is connected between this pin and the V+ pin. IN2 (Pin 2): Logic Input of Gate Drivers GA2 and GB2. IN2 is disabled when IN1 is high or DIODE is low. During 2-diode mode, asserting IN2 disables the fault timer function. GND (Pin 8): Ground. DIODE (Pin 3): “2-Diode Mode” Logic Input. Diode overrides IN1 and IN2 by forcing the two back-to-back external N-channel MOSFET switches to mimic two diodes. SAB2 (Pin 10): Source Return. The SAB2 pin is connected to the sources of SW A2 and SW B2. A small pull-down current source returns this node to 0V when the switches are turned off. TIMER (Pin 4): Fault Timer. A capacitor connected from this pin to GND programs the time the MOSFET switches are allowed to be in current limit. To disable this function, Pin 4 can be grounded. SENSE – (Pin 12): Inrush Current Input. This pin should be connected directly to the bottom (output side) of the low valued resistor in series with the two input power selector switch pairs, SW A1/B1 and SW A2/B2, for detecting and controlling the inrush current into and out of the power supply sources and the output capacitor. V+ (Pin 5): Power Supply. Bypass this pin with at least a 1µF capacitor. VGG (Pin 6): Gate Driver Supply. This high voltage supply is intended only for driving the internal micropower gate drive circuitry. Do not load this pin with any external circuitry. Bypass this pin with at least 1µF. GB2, GA2 (Pins 9, 11): Switch Gate Drivers. GA2 and GB2 drive the gates of the second back-to-back external N-channel switches. SENSE + (Pin 13): Inrush Current Input. This pin should be connected directly to the top (switch side) of the low valued resistor in series with the two input power selector switch pairs, SW A1/B1 and SW A2/B2, for detecting and controlling the inrush current into and out of the power supply sources and the output capacitor. Current limit is invoked when (VSENSE + – VSENSE –) exceeds ±0.2V. Pin Function Table PIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 NAME IN1 IN2 DIODE TIMER V+ VGG SW GND GB2 SAB2 GA2 SENSE – SENSE + GB1 SAB1 GA1 DESCRIPTION Logic Input of Gate Drivers GA1 and GB1 Logic Input of Gate Drivers GA2 and GB2 “2-Diode Mode” Logic Input Fault Timer Programs Time in Current Limit Power Supply Gate Driver Supply Switch Node of Internal Boost Switching Regulator Ground Switch Gate Driver for Switch B2 Source Return of Switch 2 Switch Gate Driver for Switch A2 Inrush Current Input, Low Side Inrush Current Input, High Side Switch Gate Driver for Switch B1 Source Return of Switch 1 Switch Gate Driver for Switch A1 NOMINAL (V) TYP MAX 1 2 1 2 1 2 1.16 2.8 9 10.2 20 0 20 0 0 17 0 10 0 17 0 10 0 10 0 17 0 10 0 17 MIN 0.4 0.4 0.4 ABSOLUTE MAX (V) MIN MAX – 0.3 7 – 0.3 7 – 0.3 7 – 0.3 5 – 0.3 10 – 0.3 20 – 0.3 20 0 0 – 0.3 20 – 0.3 10 – 0.3 20 – 0.3 10 – 0.3 10 – 0.3 20 – 0.3 10 – 0.3 20 5 LTC1473L U U U PI FU CTIO S GB1, GA1 (Pins 14, 16): Switch Gate Drivers. GA1 and GB1 drive the gates of the first back-to-back external N-channel switches. SAB1 (Pin 15): Source Return. The SAB1 pin is connected to the sources of SW A1 and SW B1. A small pull-down current source returns this node to 0V when the switches are turned off. W FU CTIO AL DIAGRA U U 16 GA1 SW A1/B1 GATE DRIVERS IN1 1 IN2 2 DIODE 15 SAB1 14 GB1 13 SENSE + INRUSH CURRENT SENSE 12 SENSE – 3 11 GA2 V+ SW A2/B2 GATE DRIVERS 6µA 10 SAB2 9 GB2 TIMER 4 TO GATE DRIVERS V+ 5 VGG 6 SW 7 R S 900k VGG SWITCHING REGULATOR GND LATCH + – 1.16V 8 1473 FD 6 LTC1473L U OPERATIO The LTC1473L is responsible for low-loss switching and isolation for a dual supply system, where during a power backup situation, a battery pack can be connected or disconnected seamlessly. Smooth switching between input power sources is accomplished with the help of low-loss N-channel switches. They are driven by special gate drive circuitry which limits the inrush current in and out of the battery packs and the system power supply capacitors. Figure 2 shows a block diagram of a switch driver pair, SW A1/B1. A bidirectional current sensing and limiting circuit determines when the voltage drop across RSENSE reaches ±200mV. The gate-to-source voltage, VGS, of the appropriate switch is limited during the transition period until the inrush current subsides. This scheme allows capacitors and MOSFET switches of differing sizes and current ratings to be used in the same system without circuit modifications. All N-Channel Switching DCIN The LTC1473L drives external back-to-back N-channel MOSFET switches to direct power from two sources: the primary battery and the secondary battery, or a battery and a DC power supply. (N-channel MOSFET switches are more cost effective and provide lower voltage drops than their P-channel counterparts.) LTC1473L V+ L1 1mH (8.5V + V +) TO GATE DRIVERS VGG Gate Drive (VGG) Power Supply VGG SWITCHING REGULATOR C2 1µF 25V GND 1473 F01 Figure 1. VGG Switching Regulator SW B1 SW A1 RSENSE OUTPUT LOAD BAT1 + COUT GA1 6V SAB1 6V VGG SW A/B GATE DRIVERS Inrush and Short-Circuit Current Limiting The LTC1473L uses an adaptive inrush current limiting scheme to reduce current flowing in and out of the battery and the following system’s input capacitor during switchover transitions. The voltage across a single small valued resistor, RSENSE, is measured to ascertain the instantaneous current flowing through either of the two switch pairs, SW A1/B1 and SW A2/B2, during the transitions. C1 1µF 25V SW The gate drive for the low-loss N-channel switches is supplied by an internal micropower boost regulator which is regulated at approximately 8.5V above V +, up to 20V maximum. In a DC supply and backup battery system, the LTC1473L V + pin is diode ORed through two external Schottky diodes connected to the two main power sources, DCIN and BAT1. Thus, VGG is regulated at 8.5V above the higher power source and will provide the overdrive required to fully enhance the MOSFET switches. For maximum efficiency the input to the boost regulator inductor is connected to V + as shown in Figure 1. C1 provides filtering to the input of the 1mH switched inductor, L1, which is housed in a small surface mount package. An internal diode directs the current from the 1mH inductor to the VGG output capacitor C2. BAT1 GB1 VSENSE + VSENSE – ± 200mV THRESHOLD BIDIRECTIONAL INRUSH CURRENT SENSING AND LIMITING LTC1473L 1473 F02 Figure 2. SW A1/B1 Inrush Current Limiting 7 LTC1473L U W U U APPLICATIO S I FOR ATIO After the transition period, the VGS of both MOSFETs in the selected switch pair rises to approximately 5.6V. The gate drive is set at 5.6V to provide ample overdrive for logiclevel MOSFET switches without exceeding their maximum VGS rating. In the event of a fault condition, the current limit loop limits the inrush of current into the short. At the instant the MOSFET switch is in current limit, i.e., when the voltage drop across RSENSE is ±200mV, a fault timer starts timing. It will continue to time as long as the MOSFET switch is in current limit. Eventually the preset time will lapse and the MOSFET switch will latch off. The latch is reset by deselecting the gate drive input. Fault time-out is programmed by an external capacitor connected between the TIMER pin and ground. Each of the switches is controlled by a logic compatible input that can interface directly with a digital pin. Using Tantalum Capacitors The inrush (and “outrush”) current of the load capacitor is limited by the LTC1473L, i.e., the current flowing both in and out of the capacitor during transitions from one input power source to another is limited. In many applications, this inrush current limiting makes it feasible to use lower cost/size tantalum surface mount capacitors in place of more expensive/larger aluminum electrolytics. Note: The capacitor manufacturer should be consulted for specific inrush current specifications and limitations and some experimentation may be required to ensure compliance with these limitations under all possible operating conditions. POWER PATH SWITCHING CONCEPTS Back-to-Back Switch Topology Power Source Selection The LTC1473L drives low-loss switches to direct power from either the battery pack or the DC supply during power backup situations. Figure 3 is a conceptual block diagram that illustrates the main features of an LTC1473L dual supply power management system starting with a 4 NiMH battery pack and a 5V/ 3.3V DC supply and ending with an uninterrupted output load. Switches SW A1/B1 and SW A2/B2 direct power from either the DC supply or the battery to the output load. The simple SPST switches shown in Figure 3 actually consist of two back-to-back N-channel switches. These low-loss N-channel switch pairs are housed in 8-pin SO or SSOP packaging and are available from a number of manufacturers. The back-to-back topology eliminates the problems associated with the inherent body diodes in power MOSFET switches and allows each switch pair to block current flow in either direction when the two switches are turned off. SW A1/B1 INRUSH CURRENT LIMITING DCIN 5V/3.3V SW A2/B2 BAT1 4 NiMH + CLOAD LTC1473L 1473 F03 Figure 3. LTC1473L PowerPath Conceptual Diagram 8 LTC1473L U U W U APPLICATIO S I FOR ATIO sources, DCIN and BAT1 through two common cathode Schottky diodes as shown in Figure 1.) The back-to-back topology also allows for independent control of each half of the switch pair which facilitates bidirectional inrush current limiting and the so-called “2-diode mode” described in the following section. The 2-diode mode is asserted by applying an active low to the DIODE input. The 2-Diode Mode COMPONENT SELECTION Under normal operating conditions, both halves of each switch pair are turned on and off simultaneously. For example, when the input power source is switched from BAT1 to DCIN in Figure 4, both gates of switch pair SW A1/B1 are normally turned off and both gates of switch pair SW A2/B2 are turned on. The back-to-back body diodes in switch pair, SW A1/B1, block current flow in or out of the BAT1 input connector. N-Channel Switches The LTC1473L adaptive inrush limiting circuitry permits the use of a wide range of logic-level N-Channel MOSFET switches. A number of dual low RDS(ON) N-channel switches in 8-lead surface mount packages are available that are well suited for LTC1473L applications. The maximum allowable drain-source voltage, VDS(MAX), of the two switch pairs, SW A1/B1 and SW A2/B2 must be high enough to withstand the maximum input DC supply voltage. Since the DC supply is in the 3.3V to 10V range, 12V MOSFET switches will suffice. In the “2-diode mode,” only the first half of each power path switch pair, i.e., SW A1 and SW A2, is turned on; and the second half, i.e., SW B1 and SW B2, is turned off. These two switch pairs now act simply as two diodes connected to the two main input power sources as illustrated in Figure 4. The power path diode with the highest input voltage passes current through to the output load to ensure that the output is powered even under start-up or abnormal operating conditions. (An undervoltage lockout circuit defeats this mode when the V + pin drops below 2.5V. The supply to V + comes from the main power As a general rule, select the switch with the lowest RDS(ON) at the maximum allowable VDS. This will minimize the heat dissipated in the switches while increasing the overall system efficiency. Higher switch resistances can be tolerated in some systems with lower current requirements, but care should be taken to ensure that the SW B1 RSENSE SW A1 OUTPUT LOAD BAT1 ON SW B2 + CIN OFF SW A2 DCIN ON OFF LTC1473L 1473 F04 Figure 4. LTC1473L PowerPath Switches in 2-Diode Mode 9 LTC1473L U W U U APPLICATIO S I FOR ATIO power dissipated in the switches is never allowed to rise above the manufacturers’ recommended level. Inrush Current Sense Resistor, RSENSE A small valued sense resistor (current shunt) is used by the two switch pair drivers to measure and limit the inrush or short-circuit current flowing through the conducting switch pair. The inrush current limit should be set at approximately 2× or 3× the maximum required output current. For example, if the maximum current required by the DC/DC converter is 2A, an inrush current limit of 6A is set by selecting a 0.033Ω sense resistor, RSENSE, using the following formula: RSENSE = (200mV)/IINRUSH Note that the voltage drop across the resistor in this example is only 66mV under normal operating conditions. Therefore, the power dissipated in the resistor is extremely small (132mW), and a small 1/4W surface mount resistor can be used in this application (the resistor will tolerate the higher power dissipation during current limit for the duration of the fault time-out). A number of small valued surface mount resistors are available that have been specifically designed for high efficiency current sensing applications. Programmable Fault Timer Capacitor, CTIMER A fault timer capacitor, CTIMER, is used to program the time duration the MOSFET switches are allowed to be in current limit continuously. This feature can be disabled by either grounding the TIMER pin or asserting DIODE low and asserting either IN1 or IN2 high. In the event of a fault condition, the MOSFET switch is driven into current limit by the inrush current limit loop. The MOSFET switch operating in current limit is in a high dissipation mode and can fail catastrophically if not promptly terminated. The fault time delay is programmed with an external capacitor connected between the TIMER pin and GND. At 10 the instant the MOSFET switch enters current limit, a 6µA current source starts charging CTIMER through the TIMER pin. When the voltage across CTIMER reaches 1.16V an internal latch is set and the MOSFET switch is turned off. To reset the latch, the logic input of the MOSFET gate driver must be deselected. The fault time delay should be programmed as large as possible, at least 3× to 5× the maximum switching transition period, to avoid prematurely tripping the protection circuit. Conversely, for the protection circuit to be effective, the fault time delay must be within the safe operating area of the MOSFET switches as stated in the manufacturer’s data sheet. The maximum switching transition period happens during a cold start, when a fully charged battery is connected to an unpowered system. The inrush current charging up the system supply capacitor to the battery voltage determines the switching transition period. The following example illustrates the calculation of CTIMER. Assume the maximum battery voltage is 10V, the system supply capacitor is 100µF, the inrush current limit is 6A and the maximum current required by the following system is 2A. Then, the maximum switching transition period is calculated using the following formula: tSW(MAX) = (VBAT(MAX) )(CIN(SYSTEM) ) IINRUSH − ILOAD (10)(100µF ) = 250µs tSW(MAX) = 6 A − 2A Multiplying 3 by 250µs gives 0.75ms, the minimum fault delay time. Make sure this delay time does not fall outside of the safe operating area of the MOSFET switch dissipating 30W (6A • 10V/2). Using this delay time the CTIMER can be calculated using the following formula: 6µA CITMER = 0.75ms = 3879pF 1.16V Therefore, CTIMER can be 3900pF. LTC1473L U W U U APPLICATIO S I FOR ATIO VGG Regulator Inductor and Capacitors The VGG regulator provides a power supply voltage significantly higher than either of the two main power source voltages to allow the control of N-channel MOSFET switches. This micropower, step-up voltage regulator is powered by the higher potential available from the two main power sources for maximum regulator efficiency. Three external components are required by the VGG regulator: L1, C1 and C2, as shown in Figure 5. L1 is a small, low current, 1mH surface mount inductor. C1 provides filtering to the input of the 1mH switched inductor and should be at least 1µF to filter switching transients. The VGG output capacitor, C2, provides storage and filtering for the VGG output and should be at least 1µF and rated for 25V operation. C1 and C2 can be ceramic capacitors. DCIN LTC1473L BAT1 V+ L1* 1mH TO GATE DRIVERS (8.5V + V +) VGG C1 1µF 25V SW VGG SWITCHING REGULATOR C2 1µF 25V GND *COILCRAFT 1812LS-105 XKBC. (708) 639-6400 1473 F05 Figure 5. VGG Step-Up Switching Regulator 11 LTC1473L U TYPICAL APPLICATIO S LTC1473L with Battery Charger DCIN 3.3V C3 22µF 25V C2** 22µF VIN SYNC AND/OR SHDN S/S Si9926DY L1A* C5 0.1µF R5 1k 100mA VSW LT®1512 GND GND D1 MBRS130LT3 BAT54C FB VC IFB 1 L1B* R4 24Ω C4 0.22µF R1 47.55k R3 1Ω C1 22µF 25V LOGIC DRIVEN R2 12.45k 2 3 4 5 CTIMER 2000pF 6 1mH *L1A, L1B ARE TWO 33µH WINDINGS ON A SINGLE INDUCTOR: COILTRONICS CTX33-3 **TOKIN CERAMIC 1E22ZY5U-C203-F 1µF 7 8 1µF LTC1473L IN1 GA1 IN2 SAB1 DIODE GB1 TIMER + V + SENSE SENSE – VGG GA2 SW SAB2 GND GB2 16 15 14 RSENSE 0.04Ω 13 + 12 COUT 11 10 9 1473 TA03 BAT1 4 NiMH 12 3.3V OR VBAT1 Si9926DY LTC1473L U TYPICAL APPLICATIO S 2-Cell Li-Ion to 5V/3.5A DC/DC Converter with Battery Charger and Automatic Switchover Between Battery and DCIN C2, 0.1µF COSC 51pF 1 CSS, 0.1µF RC, 33k TG RUN/SS 3 CC2, 220pF C1 100pF COSC 2 BOOST ITH SW 16 15 14 SFB C4 0.1µF D1 CMDSH-3 13 LTC1735 12 INTVCC SGND 11 6 BG VOSENSE 10 7 – SENSE PGND 9 8 SENSE + EXTVCC 5V 4 CC 470pF VIN L1* 10µH 5 C5 1000pF CIN 22µF 30V OS-CON + M1 Si4412DY + C3 4.7µF 16V M2 Si4412DY RSENSE 0.015Ω COUT 100µF 10V ×3 D2 MBRS140T3 VOUT 5V/3.5A + R1 105k 1% C6 100pF SGND R2 20k 1% Si9926DY 74C00 13 11 BAT54C 1 12 2 3 10 7 8 4 6 3 9 1 2 5 2600pF CTIMER 5 4 14 R5 500k 6 C7 1µF C8 1µF 7 L2** 1mH 8 LTC1473L IN1 GA1 IN2 SAB1 16 15 14 DIODE GB1 TIMER SENSE + 13 V+ SENSE – 12 VGG GA2 SW SAB2 GND GB2 RSENSE 0.033Ω 11 10 9 DCIN D4 MBRD340 D3 6.8V Si9926DY RSENSE 0.033Ω D5 MBRD340 R14 510Ω 1 R6 900k 1% R7 130k 1% R8 427k 1% R9 113k 1% OUT A OUT B 2 V– 3 IN + A REF IN – B HYST 4 LTC1442 C10 1µF 8 5 2,3 L3*** 20µH 7 V+ 6 1 R10 50k 1% R11 1132k 1% 1,4 R12 3k 1% C9 0.1µF C11 0.47µF D6 MBR0540T 2 3 4 5 6 7 R13 5.1k 1% 8 9 *SUMIDA CDRH125-10 **COILCRAFT 1812LS-105XKBC ***COILTRONICS CTX20-4 10 11 12 C16 220pF GND GND SW GND BOOST VCC1 GND VCC2 GND VCC3 UV GND PROG LT1511 VC OVP UVOUT CLP GND CLN COMP2 COMP1 BAT SENSE SPIN 24 23 C12 10µF 22 21 C13 10µF 20 19 18 17 R15 1k 16 C15 0.33µF 15 + C17 10µF 13 R19 200Ω 1% R18, 200Ω, 1% R20 395k 0.1% R21 164k 0.1% R17 4.93k 14 RSENSE 0.033Ω 8.4V Li-Ion BATTERY R16 300Ω C14 1µF 1473 TA04 13 LTC1473L U TYPICAL APPLICATIO S Automatic PowerPath Switching for 3.3V Applications DCIN 3.3V Si4966DY R1 1.65M 1% R2 1.13M 1% LTC1442 3 7 + 1 6 – BAT54C 5 1 2 + 3 8 4 4 – 5 1.182V CTIMER 4700pF 2 6 1µF 1mH* 1µF 7 8 BAT1 4 NiMH LTC1473L IN1 GA1 IN2 SAB1 16 15 14 DIODE GB1 TIMER 13 SENSE + RSENSE 0.04Ω + 12 SENSE – 11 GA2 10 SAB2 9 GB2 V+ VGG SW GND COUT 1473 TA05 * COILCRAFT 18126S-105XKBC Si4966DY 3.3V or 5V, 6A, PowerPath Switch Si4966DY DCIN 3.3V 1 BAT54C LOGIC DRIVEN 2 3 4 5 6 CTIMER 550pF DCIN 5V 14 1mH 1µF 1µF 7 8 LTC1473L IN1 GA1 IN2 SAB1 16 15 14 DIODE GB1 TIMER 13 SENSE + V+ VGG SW GND RSENSE 0.015Ω 3.3V OR 5V 6A + 12 SENSE – 11 GA2 10 SAB2 9 GB2 COUT 1473 TA06 Si4966DY 3.3V OR VBAT1 LTC1473L U TYPICAL APPLICATIO S Protected Hot SwapTM Switchover Between Two Supplies for Portable PC DOCKING CONNECTOR 5V LONG PIN 100k Q1 Si9926DY SUPPLY V1 5V D1 MMBD2838LT1 1 2 3 4 100k 5 C6 4700pF 6 C7 1µF L1*, 1mH C5 1µF 7 8 LTC1473L IN1 GA1 IN2 SAB1 GB1 DIODE 15 14 R3 0.1Ω + 13 TIMER SENSE V+ SENSE – GA2 VGG 16 SW SAB2 GND GB2 OUT 12 LONG PIN 11 10 9 SUPPLY V2 3.3V Q2 Si9926DY *1812LS-105XKBC, COILCRAFT ON SHORT PIN U PACKAGE DESCRIPTIO 1473 • TA07 Dimensions in inches (millimeters) unless otherwise noted. GN Package 16-Lead Plastic SSOP (Narrow 0.150) (LTC DWG # 05-08-1641) 0.189 – 0.196* (4.801 – 4.978) 0.009 (0.229) REF 16 15 14 13 12 11 10 9 0.229 – 0.244 (5.817 – 6.198) 0.150 – 0.157** (3.810 – 3.988) 1 0.015 ± 0.004 × 45° (0.38 ± 0.10) 0.007 – 0.0098 (0.178 – 0.249) 0.053 – 0.068 (1.351 – 1.727) 2 3 4 5 6 7 8 0.004 – 0.0098 (0.102 – 0.249) 0° – 8° TYP 0.016 – 0.050 (0.406 – 1.270) * DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.008 – 0.012 (0.203 – 0.305) 0.0250 (0.635) BSC GN16 (SSOP) 1098 Hot Swap is a trademark of Linear Technology Corporation Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC1473L U TYPICAL APPLICATIO Protected Automatic Switchover Between Two Supplies 1 5V LT1121-5 8 Q1 Si9926DY 3 SUPPLY V1 10k 1µF 1M BAT54C 1M 3 + 8 LT1490 1 2 – 1 5 4 2 + 3 7 6 4 – 1M 5 C6 2600pF 1M 6 C7 1µF 10k L1*, 1mH C5 1µF 7 8 LTC1473L IN1 GA1 IN2 SAB1 DIODE GB1 16 15 14 R3 0.033Ω + 13 TIMER SENSE V+ SENSE – VGG GA2 SW SAB2 GND GB2 OUT 12 11 10 9 SUPPLY V2 Q2 Si9926DY *1812LS-105XKBC, COILCRAFT 1473 • TA02 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1155 Dual High Side Micropower MOSFET Driver Internal Charge Pump Requires No External Components LTC1161 Quad Protected High Side MOSFET Driver Rugged, Designed for Harsh Environment LTC1735 Single High Efficiency Synchronous DC/DC Controller Constant Frequency, 3.5 ≤ VIN ≤ 36V, Fault Protection LTC1473 Dual PowerPath Switch Driver V + Range from 4.75V to 30V LTC1479 PowerPath Controller for Dual Battery Systems Designed to Interface with a Power Management µP LT1505 Synchronous Battery Charger with Adapter Current Limit High Efficiency, Up to 8A Charge Current, End-of-Charge Flag, 28-Pin SSOP, 0.5V Dropout Voltage LT1510 Constant-Voltage/Constant-Current Battery Charger Up to 1.5A Charge Current for Lithium-Ion, NiCd and NiMH Batteries LT1511 3A Constant-Voltage/Constant-Current Battery Charger High Efficiency, Minimal External Components to Fast Charge Lithium, NiMH and NiCd Batteries LTC1558/LTC1559 Backup Battery Controller with Programmable Output Power Supply Backup Using a Single NiCd Cell LTC1622 Current Mode Step-Down DC/DC Converter 550kHz Operation, 100% Duty Cycle, VIN from 2V to 10V LTC1628 Dual High Efficiency Synchronous Buck DC/DC Controller 2-Phase Switching, 5V Standby in Shutdown, Fault Protection LT1769 2A Constant-Voltage/Constant-Current Battery Charger Charges Lithium, NiCd and NiMH Batteries, 28-Lead SSOP 16 Linear Technology Corporation 1473lf LT/TP 1099 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1999