Using the ISL8104 PWM Controller Evaluation Board ® Application Note Introduction The ISL8104 is a simple single-phase PWM controller for a synchronous buck converter with integrated MOSFET driver that operates from +8V to +14.4V bias supply voltage. The ISL8104 employs voltage-mode control with dual-edge modulation to achieve fast transient response. The controller features the ability to safely start-up into prebiased output loads and provides protection against overcurrent fault events. Overcurrent protection is implemented using topside MOSFET rDS(ON) sensing, eliminating the need for a current sensing resistor. The ISL8104 evaluation board highlights the operations of the controller in a DC/DC application. ISL8104 Reference Design The evaluation board is designed to optimize for the output voltage and current specifications shown in Table 1. TABLE 1. ISL8104 EVALUATION BOARD DESIGN PARAMETERS PARAMETER Input Voltage (VIN) Output Voltage (VOUT) ESR and capacitor charge are major contributions to the output voltage ripple. Assuming that the total output capacitance is sufficient, then the output voltage ripple is dominated by the ESR, which can be calculated using Equation 2. (EQ. 2) V RIPPLE = ΔI L ⋅ ESR To meet the 30mVP-P output voltage ripple requirement, the effective ESR should be less than 4mΩ. The output voltage response to a transient load is contributed from ESL, ESR and the amount of output capacitance. With VIN>>VOUT, the amplitude of the voltage excursions can be approximated using Equation 3: 2 L ⋅ I tran ΔV = ------------------------------------C OUT ⋅ V OUT (EQ. 3) With 0.68µH inductor and 0A to 20A step load, the total output capacitance of 1900µF is required for 80mV output voltage transient. In the evaluation board, four of Fujitsu’s FP-4R0RE561M-L8 are employed. TYP MAX Input Capacitor Selection 8V 12V 14.4V The input bulk capacitors selection criteria are based on the capacitance and RMS current capability. The RMS current rating requirement for the input capacitor is approximated in Equation 4: 30mVP-P Continuous Load Current 20A Switching Frequency AN1416.1 MIN 1.8V Output Voltage Ripple (VRIPPLE) June 9, 2009 I IN ( RMS ) = 300kHz ΔI 2 I O 2 ( D – D 2 ) + -------- D 12 VO D = ---------VIN (EQ. 4) In this application, the RMS current for the input capacitor is 7.2A; therefore, three of Sanyo’s 35ME330AX are used. Design Procedure The following sections illustrate simple design steps and component selections for a converter using the ISL8104. Output Inductor Selection The output inductor is chosen by the desired inductor ripple current, which is typically set to be approximately 40% of the rated output current. The desired output inductor can be calculated using Equation 1: V IN – V OUT V OUT 1 L = -------------------------------- × ---------------- × -----------V IN ΔI F SW (EQ. 1) 14.4 – 1.8 1.8 1 = -------------------------- × ----------- × ---------------------3 0.4 ⋅ 20 14.4 300 ×10 Small ceramic capacitors for high frequency decoupling are also required to control the voltage overshoot across the MOSFETs. MOSFET Selection The ISL8104 requires two N-Channel power MOSFETs as the main and the synchronous switches. These should be selected based in rDS(ON), gate supply requirements and thermal management requirements. The total power loss on MOSFET consists of conduction loss and switching loss, as shown in Equation 5: P MOSFET ( TOT ) = P cond + P sw = 0.66μH In the evaluation board, a 0.68µH inductor with 1.6mΩ DCR (Vishay’s IHLP5050FD-R68) is employed. This yields approximately 0.64W conduction loss in the inductor. Output Capacitor Selection The output capacitors are generally selected by the output voltage ripple and load transient response requirements. 1 (EQ. 5) In this relatively small duty cycle design, the low-side MOSFET conducts current most of the time. To optimize the converter efficiency, select the high-side MOSFET with low gate charge for fast switching transition and low-side MOSFET with low rDS(ON). The budget power losses in each high-side and low-side MOSFETs is 1W. CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2008, 2009. All Rights Reserved All other trademarks mentioned are the property of their respective owners. Application Note 1416 LOW-SIDE MOSFET SELECTION Overcurrent Protection Setting The low-side MOSFET’s RMS current is approximated in Equation 6: The OCP function is enabled with the drivers at start-up. OCP is implemented via a resistor (RTSOC) and a capacitor (CTSOC) connecting the TSOC pin and the drain of the topside MOSFET. An internal 200µA current source develops a voltage across RTSOC, which is then compared with the voltage developed across the top-side MOSFET at turn on, as measured at the LX pin. When the voltage drop across the MOSFET exceeds the voltage drop across the resistor, a sourcing OCP event occurs. CTSOC is placed in parallel with RTSOC to smooth the voltage across RTSOC in the presence of switching noise on the input bus. 2 1 ⎛ ΔI L ⎞ I L ( RMS ) = I OUT ⋅ 1 – D ⋅ 1 + ------ ⋅ ⎜ -------------⎟ ≈ 18.6A 12 ⎝ I OUT⎠ (EQ. 6) Assuming a target conduction loss of 0.5W in the low-side MOSFET, the ON-resistance of the low-side MOSFET must be less than 1.5mΩ. Two of Infineon’s BSC030N03LS are employed in the evaluation board. The conduction loss in the low-side MOSFETs is calculated using Equation 7: 2 P LFET ( cond ) = I L ( RMS ) ⋅ r DS ( ON ) = 0.52W LFET (EQ. 7) The switching loss in the low-side MOSFETs is dominated by the loss in body diode, which can be calculated using Equation 8: P diode = I O ⋅ t D ⋅ V F ⋅ F SW = 0.4W (EQ. 8) Where tD is the total dead time in each switching period (~60ns) and VF is the forward voltage drop of MOSFET’s body diode. The total power dissipation in the low-side MOSFETs is calculated using Equation 9: P LFET ( TOT ) = 0.92W (EQ. 9) HIGH-SIDE MOSFET SELECTION For the high-side MOSFET selection, first we assume that the conduction loss and the switching loss contribute evenly to the total power dissipation. The high-side MOSFET’s RMS current is approximated using Equation 10: 2 1 ⎛ ΔI L ⎞ I H ( RMS ) = I OUT ⋅ D ⋅ 1 + ------ ⋅ ⎜ -------------⎟ ≈ 7.8A 12 ⎝ I OUT⎠ (EQ. 10) Hence, the required ON-resistance of the high-side MOSFET is 8.2mΩ. Infineon’s BSC080N03LS is selected. The conduction loss in the high-side MOSFET is calculated using Equation 11: 2 P HFET ( cond ) = I H ( RMS ) ⋅ r DS ( ON ) = 0.49W (EQ. 11) HFET The switching loss in the high-side MOSFET can be approximated using Equation 12: 2 1 1 P HFET ( SW ) = --- ⋅ I O ⋅ V IN ⋅ t tr ⋅ F SW + --- ⋅ C OSS ⋅ V IN ⋅ F SW 2 2 = 0.21W A 120ns blanking period is used to reduce the current sampling error due to leading-edge switching noise. An additional simultaneous 120ns low pass filter is used to further reduce measurement error due to noise. The OCP trip point varies mainly due to MOSFET rDS(ON) variations and layout noise concerns. To avoid overcurrent tripping in the normal operating load range, find the ROCSET resistor from Equation 14 with: 1. The maximum rDS(ON) at the highest junction temperature 2. The minimum ITSOC from the specification table in datasheet Determine the overcurrent trip point greater than the maximum output continuous current at maximum inductor ripple current. Simple OCP Equation I OC_SOURCE • r DS ( ON ) R TSOC = ---------------------------------------------------------------200μA Detailed OCP Equation ΔI ⎛I + -----⎞ • r ⎝ OC_SOURCE 2 ⎠ DS ( ON ) R TSOC = ---------------------------------------------------------------------------------I TSOC • N T (EQ. 14) N T = NUMBER OF TOP-SIDE MOSFETs V IN - V OUT V OUT ΔI = --------------------------------- • ---------------F SW • L OUT V IN With Infineon’s BSC080N03LS as the top-side MOSFET and RTSOC of 1.15kΩ, the overcurrent trip point on the evaluation board has been approximately set to 25A. (EQ. 12) where ttr is the combined ON and OFF MOSFET transition times. The total power dissipation in high-side MOSFET is shown in Equation 13: (EQ. 13) P HFET ( TOT ) = 0.7W 2 AN1416.1 June 9, 2009 Application Note 1416 : V OSC ⋅ R 1 ⋅ F BW R 2 = --------------------------------------------d max ⋅ V IN ⋅ F 0 C2 R2 COMP C1 = 44.2kΩ C3 R3 3. Select C1 such that FZ1 is located at 1.5kHz: FB E/A (EQ. 17) + 1 C 1 = -------------------------------------------3 2π ⋅ R 2 ⋅ 1.5 ×10 R1 (EQ. 18) ≈ 2.2nF VREF 4. Select C2 such that FP1 is located at FESR: GND C1 C 2 = -----------------------------------------------------------2π ⋅ R 2 ⋅ C 1 ⋅ F ESR – 1 (EQ. 19) ≈ 82pF VOUT OSCILLATOR R1 R 3 = -------------------------------- ≈ 655Ω 3 150 ×10 ---------------------- – 1 F0 VIN VOSC PWM CIRCUIT TGATE HALF-BRIDGE DRIVE L DCR LX BGATE ISL8104 5. Select R3 such that FZ2 is located at FLC and FP2 is located at 150kHz: (EQ. 20) 1 C 3 = ---------------------------------------------- ≈ 1.5nF 3 2π ⋅ R 3 ⋅ 150 ×10 C ESR A more detailed explanation of designing compensation networks for buck converters with voltage mode control can be found in TB417 entitled “Designing Stable Compensation Networks for Single Phase Voltage Mode Buck Regulators”. EXTERNAL CIRCUIT Evaluation Board Performance FIGURE 1. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN Figures 2 and 3 show photographs of the ISL8104EVAL1Z and ISL8104EVAL2Z boards, respectively. Feedback Compensator Type-III network is recommended for compensating the feedback loop. Figure 1 shows Type-III compensation configuration for the ISL8104. With the inductor and output capacitor selected as described in the previous sections, the poles and zero of the power stage can be summarized in Equation 15: 1 F 0 = ------------------------------------- = 4.1kHz 2×π× L×C 1 F ESR = ------------------------------------------- = 47.3kHz 2 × π × C × ESR (EQ. 15) 1. With a value of 23.2kΩ for R1, select R4 for the target output voltage of 1.8V using Equation 16: V REF R 4 = R 1 × ------------------------------------V –V OUT FIGURE 2. ISL8104EVAL1Z (EQ. 16) REF = 11.5kΩ 2. With the desired feedback loop bandwidth at approximately 50kHz, R2 can be calculated using Equation 17: 3 AN1416.1 June 9, 2009 Application Note 1416 . VIN = 12V, VOUT = 1.8V, IOUT = 20A EN VOUT SS FIGURE 3. ISL8104EVAL2Z Power and Load Connections Terminals J1 and J2 are connected to the input of the power stage. For single rail supply, the IC bias supply can be tied to the converter input supply through pin 1 and 2 of the Jumper J5. When using separate supplies, provide the IC bias voltage to terminal J6 with pin 2 and pin 3 of J5 connected together. The load can be connected to terminal J4 and J5. TP6 and TP3 can be used for DMM to measure output voltage. The toggle switch, SW1, can be used to disable the controller. Start-up When the voltages at VCC and PVCC of ISL8104 exceed their rising POR thresholds, a 30µA current source driving the SS pin is enabled. Upon the SS pin exceeding 1V, the ISL8104 begins ramping the non-inverting input of the error amplifier from GND to the System Reference. During initialization, the MOSFET drivers pull TGATE to LX and BGATE to PGND. If the ISL8104 is utilizing the internal reference, then as the SS pin’s voltage ramps from 1V to 3V, the soft-start function scales the reference input (positive terminal of error amp) from GND to VREF (0.597V nominal). Figure 4 shows the start-up profile of the ISL8104 in relation to the start-up of the 12V input supply and the bias supply. FIGURE 4. SOFT-START Soft-Start with Pre-Biased Output If the output is pre-biased to a voltage less than the expected value, the ISL8104 will detect that condition. Drivers are held in tri-state (TGATE pulled to LX, BGATE pulled to PGND) at the beginning of a soft-start cycle until two PWM pulses are detected. The bottom-side MOSFET is turned on first to provide for charging of the bootstrap capacitor. This method of driver activation provides support for start-up into prebiased loads by not activating the drivers until the control loop has entered its linear region, thereby substantially reducing output transients that would otherwise occur had the drivers been activated at the beginning of the soft-start cycle. VIN = 12V, VOUT = 1.8V, IOUT = 1A EN VOUT SS FIGURE 5. SOFT-START WITH PRE-BIASED OUTPUT 4 AN1416.1 June 9, 2009 Application Note 1416 Output Ripple Figure 6 shows the ripple voltage on the output of the regulator. IOUT VOUT FIGURE 9. TRANSIENT RESPONSE Efficiency ISL8104 based regulators enable the design of highly efficient systems. The efficiency of the evaluation board using a 12V input supply is shown in Figure 10. 95 FIGURE 6. OUTPUT RIPPLE (20MHz BW) 90 Transient Performance Figures 7, 8 and 9 show the response of the output voltage when subjected to transient loading from 0A to 15A at 1A/µs. EFFICIENCY (%) 85 80 75 70 65 60 55 IOUT 50 VOUT 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 OUTPUT CURRENT (A) FIGURE 10. EVALUATION BOARD EFFICIENCY (VOUT = 1.8V) References FIGURE 7. TRANSIENT RESPONSE For Intersil documents available on the web, go to http://www.intersil.com/. 1. ISL8104 Data Sheet “8V to 14V, Single-Phase Synchronous Buck Pulse-Width Modulation (PWM) Controller With Integrated Gate Drivers”, Intersil Corporation IOUT 2. Tech Brief TB417 , “Designing Stable Compensation Networks for Single Phase Voltage Mode Buck Regulators”, Intersil Corporation VOUT FIGURE 8. TRANSIENT RESPONSE 5 AN1416.1 June 9, 2009 ISL8104EVAL1Z Schematic J5 J6 1 2 3 VBIAS J7 R9 4.7 GND R11 4.7 J1 TP1 1 1 1 2 2 2 2 2 1 2 1 C19 330µF 1 2 TP3 L1 0.68µH 1 J3 VOUT 2 R19 0 1 R3 655 C3 1.5nF R4 11.5k R1 23.2k C13 0.01µF R20 0 1.8V 2 1 C27 2 0.10µF 1 2 R17 DNP 2 1 C5 DNP 2 1 C26 0.10µF 1 TP8 C25 2 0.10µF 2 1 C8 1000pF Q4 1 1 C9 DNP 2 Q2 2 C24 560µF R10 DNP 1 0 1 2 R18 2.2 1 2 1 BG 20A R21 DNP C28 0.10µF TP5 1 R22 (DNP) R12 C16 DNP Q3 C23 560µF Q1 C10 0.1µF ISL8104 C14 DNP TG C22 560µF 0 2 TP4 R13 1 C7 82pF C15 DNP TP2 J4 Application Note 1416 TP6 R14 0 10 BOOT FB 5 FB BOOT 9 TGATE EN 6 TGATE EN 7 8 LX GND LX 2 10k DNP 1 2 R16 R6 10k VOUT C17 TSOC 2 C1 2.2nF 1 ADDITIONAL SMD FOOTPRINT FOR OUTPUT CAPACITORS J2 PGND C21 560µF SW1 C20 330µF TP7 C18 330µF 2 R2 44.2k R8 20k VCC 14 VCC FSET TSOC PVCC 13 PVCC 3 12 BGATE SS SS BGATE COMP 4 COMP PGND 11 PGND 2 C29 1µF U1 2 1 FSET 1 1 C2 82pF 1 2 D1 1 2 6 C12 1µF 1 C4 1µF BAT54C/SOT VCC C30 1µF 2 2 C11 1000pF 2 1 1 C6 0.1µF PVIN 1.15k 2 R7 DNP R5 63.4k 1 R15 AN1416.1 June 9, 2009 Application Note 1416 ISL8104EVAL1Z Bill of Materials ID REFERENCE QTY PART NUMBER PART TYPE DESCRIPTION PACKAGE VENDOR 1 U1 1 ISL8104IBZ IC, Linear IC, Single PWM Controller 14 Ld SOIC Intersil 2 Q1 1 BSC080N30LS G MOSFET 30V N-Channel MOSFET TDSON-08 Infineon 3 Q2, Q4 2 BSC030N03LS G MOSFET 30V N-Channel MOSFET TDSON-08 Infineon 4 Q3 DNP 5 D1 1 BAT54C Schottky Diode 30V, 200mA Schottky Diode SOT23 Onsemi 6 L1 1 IHLP5050FD-R68 Inductor 0.68µH, High Current Inductor SMD Vishay 7 SW1 1 GT11MSCKE Toggle Switch Switchtoggle, SMD, Ultramini,1P, SPST Mini C&K 8 C1 1 Capacitor, Ceramic, 2200pF, 50V, 10%, ROHS X7R SM_0603 Generic 9 C2,C7 2 Capacitor, Ceramic, 82pF, 50V, 10%, ROHS COG SM_0603 Generic 10 C3 1 Capacitor, Ceramic, 1500pF, 50V, 10%, ROHS X7R SM_0603 Generic 11 C4, C12, C29,C30 4 Capacitor, Ceramic, 1µF, 25V, 10%, ROHS X5R SM_0603 Generic 12 C6, C10, C25, C26, C27, C28 6 Capacitor, Ceramic, 0.1µF, 50V, 10%, ROHS X7R SM_0603 Generic 13 C8, C11 2 Capacitor, Ceramic, 1000pF, 50V, 10%, ROHS X7R SM_0603 Generic 14 C13 1 Capacitor, Ceramic, 0.01µF, 50V, 10%, ROHS X7R SM_0603 Generic 15 C18, C19, C20 3 35ME330AX Aluminum Capacitor 330µF, 35V RAD 10x20 Sanyo 16 C21, C22, C23, C24 4 FP-4R0RE561M-L8R Polymer Alumium Capacitor MOSFET CAPACITORS 560µF, 4V, 20%, ROHS RAD 8x8 Fujistu 23.2kΩ, 1%, 1/16W SM_0603 Generic 17 C5, C9, C14, C15, C16, C17 DNP RESISTORS 18 R1 1 Resistor, Film 19 R2 1 Resistor, Film 44.2kΩ, 1%, 1/16W SM_0603 Generic 20 R3 1 Resistor, Film 665Ω, 1%, 1/16W SM_0603 Generic 21 R4 1 Resistor, Film 11.5kΩ, 1%, 1/16W SM_0603 Generic 22 R5 1 Resistor, Film 63.4kΩ, 1%, 1/16W SM_0603 Generic 23 R6 1 Resistor, Film 10kΩ, 1%, 1/16W SM_0603 Generic 24 R8 1 Resistor, Film 20kΩ, 1%, 1/16W SM_0603 Generic 25 R9, R11 2 Resistor, Film 4.7Ω, 1%, 1/16W SM_0603 Generic 26 R12, R13, R14, R19, R20 Resistor, Film 0Ω, 1/16W SM_0603 Generic 27 R15 1 Resistor, Film 1.15kΩ, 1%, 1/16W SM_0603 Generic 28 R16 1 Resistor, Film 10Ω, 1%, 1/16W SM_0603 Generic Resistor, Film 2.2Ω, 1%, 1/16W SM_0603 Generic 29 R18 1 30 R7, R10, R17, R21, R22 DNP SM_0603 OTHERS 31 J1, J3 2 111-0702-001 Blinding Post Conn-Gen, Bind. Post, Red, Thmbnut-Gnd Johnson Components 32 J2, J4 2 111-0703-001 Blinding Post Conn-Gen, Bind. Post, Black, Thmbnut-Gnd Johnson Components 7 AN1416.1 June 9, 2009 Application Note 1416 ISL8104EVAL1Z Bill of Materials (Continued) ID REFERENCE QTY 33 J6, J7 2 1514-2 34 J5 1 68000-236-1X3 35 TP1 through TP7 7 5002 36 TP8 DNP 8 PART NUMBER PART TYPE Turrett Post Test Point DESCRIPTION PACKAGE VENDOR Conn-Turret, Terminal Post, TH, ROHS Keystone 3-pin Jumper Berg/FCI Conn-Mini Test Point, Vertical, White, ROHS Keystone AN1416.1 June 9, 2009 Application Note 1416 ISL8104EVAL1Z Printed Circuit Board Layers FIGURE 11. ISL8104EVAL1Z - TOP LAYER (SILKSCREEN) FIGURE 12. ISL8104EVAL1Z - TOP LAYER (COMPONENT SIDE) FIGURE 13. ISL8104EVAL1Z - LAYER 2 FIGURE 14. ISL8104EVAL1Z - LAYER 3 FIGURE 15. ISL8104EVAL1Z - BOTTOM LAYER 9 FIGURE 16. ISL8104EVAL1Z - BOTTOM LAYER (SOLDER SIDE) AN1416.1 June 9, 2009 ISL8104EVAL2Z Schematic J5 J6 1 2 3 VBIAS R9 4.7 J7 R7 DNP R23 10k 1 1 1 2 2 2 1 2 2 2 2 TG Q3 TP5 TP3 J3 C8 1000pF C5 R17 DNP 1 DNP 2 R19 0 C3 1.5nF R3 665 R1 23.2K 1 1 C13 0.01uF 2 2 R4 11.5K 2 2 1 C26 0.10µF C24 560µF R20 0 R21 DNP C27 0.10uF 0.10µF C28 0.10uF 0.10µF TP8 TP2 J4 1 2 Q4 2 Q2 C22 560µF 2 2 1 R22 DNP C25 0.10µF C23 560µF C21 560µF C9 DNP R10 DNP 1 0 2 2 1 R12 C31 DNP 1 1 1 R18 2.2 1.8V 20A VOUT 2 2 1 L1 0.68µH 1 1 Q1 2 0 1 R13 TGATE BG 2 C14 DNP C16 DNP R14 0 0.1µF 1 C15 DNP C17 DNP 9 BOOT TP9 TP10 PGND 10 PGND 2 8 LX 7 6 5 REFIN 2 TP4 C7 82pF VOUT 1 2 1 BOOT 1 2 1 PGND EN J2 Application Note 1416 TP6 R6 10k 2 2 14 FSET 13 VCC 15 TSOC FB ADDTIONAL SMD FOOTPRINT FOR OUTPUT CAPACITORS C20 330µF TP7 330uF 11 BGATE BGATE SW1 R16 10 C19 330µF C18 330µF 12 PVCC PVCC ISL8104 D1 BAT54C/SOT 1 2 C30 1µF C29 1µF 2 EN 4 C12 1uF 1µF 1 C1 2.2nF COMP 3 PVIN 1 1 COMP 2 FB 1 SS 2 C11 1000pF VCC TGATE R2 44.2k R8 20k 2 C2 82pF 1 SS REFIN 1 2 LX 1 FSET 16 EPAD C6 0.1µF SSDONE 17 U1 TSOC SSDONE 1 GND 10 2 J1 1 C4 1µF 1uF D2 VCC R15 1.15k TP1 R5 63.4k 1 GND R11 4.7 AN1416.1 June 9, 2009 Application Note 1416 ISL8104EVAL2Z Bill of Materials ID REFERENCE QTY PART NUMBER PART TYPE DESCRIPTION PACKAGE VENDOR 1 U1 1 ISL8104IRZ IC, Linear IC, Single PWM Controller 16 Ld 4x4 MLFP Intersil 2 Q1 1 BSC080N30LS G MOSFET 30V N-Channel MOSFET TDSON-08 Infineon 3 Q2, Q4 2 BSC030N03LS G MOSFET 30V N-Channel MOSFET TDSON-08 Infineon 4 Q3 DNP 5 D1 1 30V, 200mA Schottky diode SOT23 Onsemi MOSFET BAT54C Schottky Diode 6 L1 1 IHLP5050FD-R68 Inductor 0.68µH, high current inductor SMD Vishay 7 SW1 1 GT11MSCKE Toggle Switch Switchtoggle, SMD, Ultramini,1P, SPST Mini C&K 8 D2 1 597-3311-407F LED Green LED 9 C1 1 10 C2, C7 11 12 SMD 1206 Dialight Capacitor, Ceramic, 2200pF, 50V, 10%, ROHS X7R SM_0603 Generic 2 Capacitor, Ceramic, 82pF, 50V, 10%, ROHS COG SM_0603 Generic C3 1 Capacitor, Ceramic, 1500pF, 50V, 10%, ROHS X7R SM_0603 Generic C4,C12,C29,C30 4 Capacitor, Ceramic, 1µF, 25V, 10%, ROHS X5R SM_0603 Generic 13 C6, C10, C25, C26, C27, C28 6 Capacitor, Ceramic, 0.1µF, 50V, 10%, ROHS X7R SM_0603 Generic 14 C8, C11 2 Capacitor, Ceramic, 1000pF, 50V, 10%, ROHS X7R SM_0603 Generic 15 C13 1 Capacitor, Ceramic, 0.01µF, 50V, 10%, ROHS X7R SM_0603 Generic 16 C18,C19,C20 3 35ME330AX Aluminum Capacitor 330µF, 35V RAD 10x20 Sanyo 17 C21,C22,C23,C24 4 FP-4R0RE561ML8R Polymer Alumium Capacitor 560µF, 4V, 20%, ROHS RAD 8x8 Fujistu 18 C5, C9, C14, C15, C16, C17, C31 DNP CAPACITORS RESISTORS 19 R1 1 Resistor, Film 23.2kΩ, 1%, 1/16W SM_0603 Generic 20 R2 1 Resistor, Film 44.2kΩ, 1%, 1/16W SM_0603 Generic 21 R3 1 Resistor, Film 665Ω, 1%, 1/16W SM_0603 Generic 22 R4 1 Resistor, Film 11.5kΩ, 1%, 1/16W SM_0603 Generic 23 R5 1 Resistor, Film 63.4kΩ, 1%, 1/16W SM_0603 Generic 24 R6, R23 2 Resistor, Film 10kΩ, 1%, 1/16W SM_0603 Generic 25 R8 1 Resistor, Film 20kΩ, 1%, 1/16W SM_0603 Generic 26 R9, R11 2 Resistor, Film 4.7Ω, 1%, 1/16W SM_0603 Generic Resistor, Film 0Ω, 1/16W SM_0603 Generic 27 R12, R13, R14, R19, R20 28 R15 1 Resistor, Film 1.15kΩ, 1%, 1/16W SM_0603 Generic 29 R16 1 Resistor, Film 10Ω, 1%, 1/16W SM_0603 Generic 30 R18 1 Resistor, Film 2.2Ω, 1%, 1/16W SM_0603 Generic 31 R7, R10, R17, R21, R22 DNP SM_0603 OTHERS 32 J1, J3 2 11 111-0702-001 Blinding Post Conn-Gen, Bind. Post, RED, Thmbnut-Gnd Johnson Components AN1416.1 June 9, 2009 Application Note 1416 ISL8104EVAL2Z Bill of Materials (Continued) ID REFERENCE QTY 33 J2, J4 2 111-0703-001 Blinding Post Conn-Gen, Bind. Post, Black, Thmbnut-Gnd Johnson Components 34 J6, J7 2 1514-2 Turrett Post Conn-Turret, Terminal Post, TH, ROHS Keystone 35 J5 1 68000-236-1x3 3-pin Jumper Berg/FCI 36 TP1 through TP7, TP9,TP10 9 5002 Conn-Mini Test Point, Vertical, White, ROHS Keystone 37 TP8 DNP 12 PART NUMBER PART TYPE Test Point DESCRIPTION PACKAGE VENDOR AN1416.1 June 9, 2009 Application Note 1416 ISL8104EVAL2Z Printed Circuit Board Layers FIGURE 17. ISL8104EVAL2Z - TOP LAYER (SILKSCREEN) FIGURE 18. ISL8104EVAL2Z - TOP LAYER (COMPONENT SIDE) FIGURE 19. ISL8104EVAL2Z - LAYER 2 FIGURE 20. ISL8104EVAL2Z - LAYER 3 FIGURE 21. ISL8104EVAL2Z - BOTTOM LAYER FIGURE 22. ISL8104EVAL1Z - BOTTOM LAYER (SOLDER SIDE) Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that the Application Note or Technical Brief is current before proceeding. For information regarding Intersil Corporation and its products, see www.intersil.com 13 AN1416.1 June 9, 2009