Si9961
Vishay Siliconix
12-V Voice Coil Motor Driver
FEATURES
D 1.8-A H-Bridge Output
D Class B Linear Operation
D Externally Programmable Gain and
Bandwidth
D Undervoltage Head Retract
D Rail-to-Rail Output Swing
D Programmable Retract Current D Single 12-V Supply
D Low Standby Current
D System Voltage Monitor with Fault Output
DESCRIPTION
The Si9961 is a linear actuator (voice coil motor) driver suitable
for use in disk drive head positioning systems. The Si9961
contains all of the power and control circuitry necessary to
drive the VCM that is typically found in 31/2-inch hard disk
drives and optical disk drives. The driver is capable of
delivering 1.8 A at a nominal supply of 12 V.
operation during linear tracking. Externally programmable
gain switch at the input summing junction increases the
resolution and dynamic range for a given DAC. The head
retract circuitry can be activated by either an undervoltage
condition or an external command. An external resistor is
required to set the VCM current during retract.
The Si9961 provides all necessary functions including a motor
current sense amplifier, a loop compensation amplifier and a
power amplifier featuring four complementary MOSFETs in a
H-bridge configuration. The output crossover protection
ensures no cross-conducting current and true Class B
The Si9961 is constructed on a self-isolated BiC/DMOS power
IC process. The IC is available in 24-pin SO package for
operation over the commercial, C suffix (0 to 70_C)
temperature range.
FUNCTIONAL BLOCK DIAGRAM
FAULT
VCC
V+
EXT
VREF
VREF–
VDD
8
7
12
18
Q1
4
5
Q3
VR
Voltage
Monitor
8R
IA2–
23
–
A2
R
A4
+
+
–
Q2
17
OUTPUT
A
19
OUTPUT
B
Q4
VR
RETRACT
IRET
Enable
OA2
VR
9
Retract
Control
6
11
22
A5
–
Acceleration Error
+
R
VR
7R
GAIN
SELECT
–
+
1
RINH
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
A3
10
2
RINL
24
RFB
3
ISENSE
OUT
13
ISENSE
IN+
21
ISENSE
IN–
15 14
SA
GND
16
20
SB
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Si9961
Vishay Siliconix
ABSOLUTE MAXIMUM RATINGS
Operating Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 70_C
Voltages Referenced to Common Pin
V+ Supply Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to 16 V
Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150_C
Pin (FAULT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VCC + 0.3 V
Power Dissipation (Package)a
24-Pin SOICb . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.125 W
Pin (Output A & B, Source A & B) . . . . . . . . . . . . . . . . –0.3 V to VDD + 0.3 V
Pin (All Others) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to V+ + 0.3 V
Maximum Clamp Current
Output A, Output B (Pulsed 10 ms at 10% duty cycle) . . . . . . . . . . . . "1.8 A
Pin (All Others) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . "20 mA
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65 to 150_C
Thermal Impedance (JA)a
24-Pin SOIC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40_C/W
Notes
a. Device mounted with all leads soldered or welded to PC board.
b. Derate 25 mW/_C above 25_C.
SPECIFICATIONS
Test Conditions
Unless Otherwise Specified
Limits
C Suffix 0 to 70_C
Symbol
V+ = 12 V "
"10%, VDD = 11.6 V "
"10%
VCC = 5 V "10%, VREF– = GND = 0 V
VREF = 5 V "5%
Minb
Typa
High Level Output Voltage
VOH
IOH = 1.0 A, VDD = 10.2 V, OA2= VREF "1 V
8.0
9.1
Low Level Output Voltage
VOL
IOL = –1.0 A, OA2 = VREF "1 V
Clamp Diode Voltage
VCL
IF = 1.0 A, ENABLE = High
Parameter
Maxb
Unit
1.1
V
Bridge Outputs (A4, A5)
Output VRANGE = VREF "2 V
Amplifier Gain
Dynamic Crossover Current
Slew Rate
0.6
2.5
12
Measured at VDD
SR
16
18
10
mA
1
Small Signal Bandwidth (–3 dB)
V/S
0.2
Input Deadband
V/V
–60
MHz
60
mV
–8
8
mV
–50
50
A2, Loop Compensation Amplifier
Input Offset Voltage
Input Bias Current
VOS
IB
Unity Gain Bandwidth
Slew Rate
RLOAD = 10 k, CLOAD = 100 pF to VREF
SR
Power Supply Rejection Ratio
PSRR
Open Loop Voltage Gain
AVOL
Output Voltage Swing
Gain Select = High, IA2– = 5 V
VO
1
1
@ 10 kHz
V/s
50
dB
80
RLOAD = 10 k to VREF
nA
MHz
VREF –2
VREF +2
V
–5
5
mV
A3, Current Sense Amplifier
Input Offset Voltage
VOS
Input Impedance
RIN
Small Signal Bandwidth (–3 dB)
Common Mode Rejection Ratio
Slew Rate
CMRR
ISENSEIN+ to ISENSEIN–
5
k
RLOAD = 10 k, CLOAD = 100 pF to VREF
1
MHz
@ 5 kHz
SR
Gain
Input Common-Mode Voltage Range
Output Voltage Swing
50
dB
2
3.9
V/s
4
4.1
VCM
To GND
–0.3
2
VO
RLOAD = 10 k, CLOAD = 100 pF to VREF
VREF –2
VREF +2
ICC
Static, No Load
IV+
RETRACT = High
2
5
IDD
ENABLE = Low
5
13
V/V
V
Supply
Supply Current (Normal)
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2
0.01
mA
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
Si9961
Vishay Siliconix
SPECIFICATIONS
Test Conditions
Unless Otherwise Specified
Parameter
Limits
C Suffix 0 to 70_C
Symbol
V+ = 12 V "10%, VDD = 11.6 V "10%
VCC = 5 V "10%, VREF– = GND = 0 V
VREF = 5 V "5%
ICC
Static, No Load
IV+
RETRACT = High
0.2
0.4
IDD
ENABLE = High
0.8
1.6
11.6
13.2
Minb
Typa
Maxb
Unit
Supply
Supply Current (Standby)
0.01
Normal Mode
10.2
Retract Mode
2.0
VDD Range
VDD
VCC Range
VCC
4.5
5
5.5
V+ Range
V+
10.8
12
13.2
108
240
mA
14
V
Gain Select Switch
RFB Switch Resistance
IA2– = 5 V
RINH Switch Resistance
135
300
810
1800
0.15
0.40
0.65
mA
4.75
5
5.25
V
3.82
4.12
4.42
RINL Switch Resistance
VREF (EXT)
Input Current
IREF
External Voltage Range
VREF
OA2 = VREF
Power Supply Monitor
VCC Undervoltage Threshold
VREF = 5.0 V
Hysteresis
40
V+ Undervoltage Threshold
VREF = 5.0 V
9.1
Hysteresis
9.8
V
mV
10.6
100
V
mV
Gain Select, RETRACT, ENABLE Input
Input High Voltage
VIH
3.5
Input Low Voltage
VIL
Input High Current
IIH
VIN = 5 V
–1
1
Input Low Current
IIL
VIN = 0 V
–1
1
VOH
IOH = –100 A
VCC
–0.8
Output Low Voltage
VOL
IOL = 1.6 mA
0.25
0.50
Output High Sourcing Current
IOHS
VOUT = 0 V
400
1100
1.5
V
A
FAULT Output
Output High Voltage
VCC
–0.33
V
A
RETRACT Current Control (RETRACT = Low, Output Current from A to B)
IRET Bias Voltage
V(IRET)
VDD = 10 V, RRET = 3.74 k
Retract Output Pull-Up Voltage
VOUT A
VDD = 2.5 V to 14 V, IOUTA = 30 mA
VDD –1
IOUTB
VDD = 10 V, VOUTB = 5 V RRET = 3.74 k
RSB = 0.5 , TA = 25_C
22
IOUTB (Max)
VDD = 2 V, VOUTB = 0.7 V RRET = < 10 , RSB = 0.5 ,
40
Retract Output Pull-Down Current
Maximum Emergency Retract Current
Retract Current VDD Supply Rejection Ratio
Retract Current Temperature Coefficient
0.66
V
30
38
mA
VDD = 2 V to 14 V, RRET = 3.74 k
3.0
%/V
VDD = 10 V, RRET = 3.74 k
–0.3
%/_C
Notes
a. Typical values are for DESIGN AID ONLY, not guaranteed nor subject to production testing.
b. The algebraic convention whereby the most negative value is a minimum and the most positive a maximum.
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
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Si9961
Vishay Siliconix
PIN CONFIGURATION
24-Pin SOIC
(Wide Body)
RINH
1
24
RFB
RINL
2
23
IA2–
ISENSE OUT
3
22
OA2
FAULT
4
21
ISENSE (IN)–
VCC
5
20
SOURCE B
IRET
6
19
OUTPUT B
EXT VREF
7
18
VDD (Spindle Supply)
V+
8
17
OUTPUT A
RETRACT
9
16
GND
GAIN SELECT
10
15
SOURCE A
ENABLE
11
14
GND
VREF–
12
13
ISENSE (IN)+
Order Number: Si9961ACY
Top View
APPLICATIONS
Introduction
User-Programmable Gains
The Si9961 Voice Coil Motor (VCM) driver integrates the active
feedback and drive components of a head-positioning servo
loop for high-performance hard-disk applications. The Si9961
operates from a 12-V ("10%) power supply and delivers 1 A
of steady-state output current. This device is made possible by
a power IC process which combines bipolar, CMOS and
complimentary DMOS technologies. CMOS logic and linear
components minimize power consumption, bipolar front-ends
on critical amplifiers provide necessary accuracy, and
complimentary (p- and n-channel) DMOS devices allow the
transconductance output amplifier to operate from ground to
VDD. Two user-programmable, current feedback/input voltage
ratios may be digitally selected to optimize gain for both seek
and track following modes, to maximize system accuracy for
a given DAC resolution. An undervoltage lockout circuit
monitors the V+ supply and generates a fault signal to trigger
an orderly head-retract sequence at a voltage level sufficient
to allow the spindle motor’s back EMF-generated voltage to
supply the necessary head parking energy. Head retract can
also be commanded via a separate RETRACT input. VCM
current during retract can be user programmed with a single
external resistor. External components are limited to R/C filter
components for loop compensation and the resistors that are
required to program gain, retract current, and the load current
sense.
During linear operation, the transconductance amplifiers’
gains (input voltage at VIN vs. VCM current, in Figure 1) are set
by external resistors R3 R5, RSA, and RSB and selected by
gain input. After selecting a value for RSA and RSB that will yield
the desired VCM current level, the High and Low feedback
gain ratios may be determined by the following:
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4
High Gain +
Low Gain +
ǒǓ
ǒǓ
R5
R3
R5
R4
1
4 RS
(GAIN SELECT Input = High)
1
4 RS
(GAIN SELECT Input = Low)
Where RS = RSA = RSB
Input offset current may then be calculated as:
1
IOS + 4 R
S
ǒ
ǒ
ǒRS
Ǔ
) RINǓ
V
) 5 VIAS3Ǔ
RIN OSA2
Where RIN = R3 or R4
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
Si9961
Vishay Siliconix
Back EMF Supply
12 V
System
Supply
5-V Ref
V+
4
mP
EXT
VREF
8
7
VREF– 12
VDD 18
FAULT
Voltage
Monitor
5
5V
VCC
VR
Q1
VCM
8R
23
IA2–
A2
–
17
R
OUTPUT
C2 R2
A
19
A4
+
CL
+
–
RL
VR
9
6
IOUT
Q3
Q2
Q4
OUTPUT
B
VR
RETRACT
IRET
Retract
Control
ENABLE
11
OA2
A5
–
Acceleration Error
22
+
RRET
R
7R
VR
A3
GAIN
SELECT
–
10
+
mP
RINH RINL
RFB
ISENSE ISENSE ISENSE
OUT
IN+
IN–
1
24
3
2
R3
R4
13
GND
A
21
B
15 14
16 20
R5
VIN
RSA
RSB
FIGURE 1. Si9961 Typical Application
Head Retract
A low on the RETRACT input pin turns output devices Q1 and
Q4 on, and output devices Q2 and Q3 off. Maximum VCM
current can be set during head retract by adding an external
resistor between the IRET pin and ground. Maximum retract
current may be calculated as:
IOUT + 175 x Iret + 175 x
0.66 V
Rret
Head retract can be initiated automatically by an undervoltage
condition (either the 12-V or 5-V supplies on the Si9961) by
connecting the FAULT output to the RETRACT input.
A high ENABLE input puts both driver outputs in a
high-impedance state. The ENABLE function can be used to
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
eliminate quiescent output current when power is applied but
the head has been parked, such as a sleep mode. A
sleep-mode power down sequence should be preceded by a
retract signal since a power failure during this state may not
provide adequate spindle-motor back EMF to permit head
retraction.
Transconductance Amplifier Compensation
The Si9961CY features an integrated transconductance
amplifier to drive the voice coil motor (VCM). To ensure proper
operation, this amplifier must be compensated specifically for
the VCM being driven. As a first approximation, the torque
constant and inertia of the VCM may be ignored, although they
will have some influence on the final results, especially if large
values are involved. (See Figure 1.)
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5
Si9961
Vishay Siliconix
A = 16 x RL/10000
CL = Lv/(Rv x RL) = 100 x 10–6/RL farads
Frequency Compensation:
The VCM transconductance (in siemens) of this simplified
case may be expressed in the s (Laplace) plane as:
1
Lv
gv +
Rv
Lv
s )
Where Rv = VCM resistance in ohms
LV = VCM inductance in henrys
s is the Laplace operator
In this case, the transconductance pole is at –Rv/Lv. It is
desirable to cancel this pole in the interest of stability. To do
this, a compensation amplifier is cascaded with the VCM and
its driver. The transfer function of this amplifier is:
ǒs )
Hc + A
1
RL
CL
Ǔ
s
Where RL = Compensation amplifier feedback
resistor in ohms
CL = Compensation amplifier feedback
capacitor in farads
A = Compensation amplifier and driver
voltage gain at high frequency
If RL x CL is set equal to Lv/Rv, then the combined open loop
transconductance in siemens becomes:
gto +
The first two problems can be considered together. Let us
assume a disk drive with a spindle RPM of 4400 and with
50 servo sectors per track. The sample rate is therefore:
f s + 50
A
Lv
A
s )
B
Lv
The entire transconductance now contains only a single pole
at –A*B/Lv. A and B are chosen to be considerably higher than
the servo bandwidth, to avoid undue phase margin reduction.
As a typical example, in the referenced schematic, assume
that Rsa and Rsb = 0.5 , R5= R3 = 10 k, VCM inductance
(Lv) = 1.5 mH, VCM resistance (Rv) = 15 . Hence:
Rv = 15 Lv = 1.5 mH
B = 2
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440
60
This is a sample frequency of 3667 Hz
As a rule of thumb, the open loop unity gain crossover
frequency of the entire servo (mechanical + electrical +
firmware) loop should be less than 1/10 of the sample
frequency. In this example, the servo open loop unity gain
crossover frequency would be less than 367 Hz. If we allow
only a 10_ degradation in phase margin due to the
transconductance amplifier, then a phase lag of 10_ at 367 Hz
is acceptable. This results in a 3-dB point in the
transconductance at :
f3db +
Lv
Where B = Current feedback transimpedence amplifier gain in
ohms.
6
There are three things to consider when optimizing the gain (A)
above. The first is servo bandwidth. The main criterion here is
to avoid having the transconductance amplifier cause an
undue loss of phase margin in the overall servo (mechanical
+ electrical + firmware) loop. The second is to avoid confirguing
a bandwidth that is more than required in view of noise and
stability considerations. The third is to keep the voltage output
waveform overshoot to a level that will not cause
cross-conduction of the output FETs.
A
s
In this case, the transconductance has a single pole at the
origin. If this open loop transfer is closed with a
transimpedance amplifier having a gain of B ohms, the
resultant closed loop transconducatance stage has the
transfer function (in siemens) of:
gtc +
Gain Optimization:
367
tan (10)
or a 3-dB point in the transconductance at 2081 Hz.
The pole in the closed loop transconductance (–A * B / Lv)
should then be 2081 * 2 * = 13075. This means that A = 9.8.
From the above equation for A, RL = 6.2 k. This sets the
minimum gain limit governed by the servo bandwidth
requirements. The gain should not be much greater than this,
since increased noise will degrade the servo response.
The third problem, keeping the transconductance amplifier
voltage output wave form overshoot to a level that will not
cause the wrong output FETs to conduct, can be evaluated by
deriving the voltage transfer function of the closed loop
transconductance amplifier from input voltage to output
voltage (Vin to output A and B on the reference schematic).
This is :
Hto + A
Where
s ) p
s ) x
p = 1/RL x CL) or Rv/Lv Comp amplifier
zero/VCM pole
x = A x B/Lv closed loop pole
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
Si9961
Vishay Siliconix
If a unit step voltage is applied to the above transfer function
and the inverse Laplace transform is taken, the output result is:
p ) (x * p) x e*x
x
VO + A
t = time
As we can see, if x = p (i.e. if the VCM pole and compensation
amplifier zero = the transconductance closed loop pole), then
Vo reduces to A. In other words, a step input results in a step
output without overshoot. If x < p then a step input results in an
increased rise time output and no overshoot. If x > p, a step
input results in a step output with an overshoot.
If this overshoot is large enough, there may be a
cross-conduction condition in the output FETs.
Let us look at the above equation at t = 0 and t >> 0, expressed
in terms of the open loop high frequency voltage gain, A.
VO + A
VO +
In the example for the 2081-Hz roll-off case with 31%
overshoot and proper pole cancellation, the compensation
values are:
RL = 6.2 k
CL = 0.016 F
In the example for the 1592-Hz roll-off case with no overshoot
and proper pole cancellation, the compensation values are:
RL = 4.7 k
CL = 0.022 F
The linearity of the transconductance amplifier (around a
center value of 500 mA/volt) is shown in Figure 2. In this case,
the output current sense resistors (RSA and RSB) were "5%
tolerance, 0.5 . Any mismatch between RSA and RSB
contribute directly to mismatch between the positive and
negative “full-scale”. Including the external resistor mismatch,
the overall loop nonlinearity is approximately 1% maximum
over a "250-mV input voltage range.
At t = 0
p
Lv
B
At t uu 0
In the example shown above, p = 10,000 and A = 9.8. This
means that there is some overshoot. At t = 0, the output voltage
is 9.8 V per volt of input. At some later time, it has dropped to
7.5 V per volt of input. An overshoot of 31 % is thus produced.
The maximum overshoot voltage requires careful
consideration, since it constitutes a potentially catastrophic
problem area. If we had decided to optimize for no overshoot,
A would equal 7.5, and hence the closed loop pole (A * B / Lv)
would be 10,000, which is a frequency of 1.592 kHz. This
would have resulted in a phase margin degradation of 13_ at
the 367-Hz frequency desired. This may or may not be
acceptable. One must weigh the servo bandwidth, phase
margin degradation, and maximum voltage at the VCM for
each individual case.
Document Number: 70014
S-20883—Rev. G, 24-Jun-02
5
4
Error in Percent of Full Scale
Where
t
Result:
3
2
1
0
VDD = 12 V
RSA = RSB = 0.5 ”5%
Rm = 52 Gm = 500 mA/V
–1
–2
–3
–4
–5
–300
–200
–100
0
100
200
300
VIN in mV
FIGURE 2. Si9961 Transconductance
End Point Non-Linearity
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Si9961
Vishay Siliconix
0.016 F
CL
6.2 k
RL
8R
VDD
RIN
–
VIN
A2
R
A4
+
+
VR
IOUT
Cross-Over
Protection
–
10 k
VR
VCM
1.5 mH
15 VDD
A5
Gain +
VS
R5
10 k
VIN
–
+
Cross-Over
Protection
R
7R
VR
A3
–
VS
+
(4 x Gain)
RSA
0.5 RSB
0.5 VR
FIGURE 3. Transconductance Amplifier
RL = 6.2 k , CL = 0.016 F
RL = 6.2 k , CL = 0.016 F
0
–5
–8
PHASE (in degrees)
GAIN (in dB)
–20
–11
–14
–40
–60
–17
–20
1
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8
10
100
1000
10000
–80
1
10
100
Frequency (Hz)
Frequency (Hz)
FIGURE 4.
FIGURE 5.
1000
10000
Document Number: 70014
S-20883—Rev. G, 24-Jun-02