OPA OPA 468 4 4 68 OPA4684 4 ® SBOS240B – JUNE 2002 – REVISED AUGUST 2002 Quad, Low-Power, Current-Feedback OPERATIONAL AMPLIFIER FEATURES APPLICATIONS ● ● ● ● ● ● ● ● ● ● ● ● ● ● MINIMAL BANDWIDTH CHANGE VERSUS GAIN 170MHz BANDWIDTH AT G = +2 > 120MHz BANDWIDTH TO GAIN > +10 LOW DISTORTION: < –78dBc at 5MHz HIGH OUTPUT CURRENT: 120mA SINGLE +5V TO +12V SUPPLY OPERATION DUAL ±2.5 TO ±6.0V SUPPLY OPERATION LOW SUPPLY CURRENT: 6.8mA Total DESCRIPTION The OPA4684 provides a new level of performance in lowpower, wideband, current-feedback (CFB) amplifiers. This CFBPLUS amplifier is among the first to use an internally closed-loop input buffer stage that enhances performance significantly over earlier low-power CFB amplifiers. While retaining the benefits of very low power operation, this new architecture provides many of the benefits of a more ideal CFB amplifier. The closed-loop input stage buffer gives a very low and linearized impedance path at the inverting input to sense the feedback error current. This improved inverting input impedance retains exceptional bandwidth to much higher gains and improves harmonic distortion over earlier solutions limited by inverting input linearity. Beyond simple high-gain applications, the OPA4684 CFBPLUS amplifier permits the gain setting element to be set with considerable LOW-POWER BROADCAST VIDEO DRIVERS EQUALIZING FILTERS SAW FILTER HIGH-GAIN POST AMPLIFIERS MULTICHANNEL SUMMING AMPLIFIERS WIDEBAND DIFFERENTIAL CHANNELS ANALOG-TO-DIGITAL CONVERTERS (ADC) INPUT DRIVERS ● MULTIPLE POLE ACTIVE FILTERS ● OPA4658 LOW-POWER UPGRADE freedom from amplifier bandwidth interaction. This allows frequency response peaking elements to be added, multiple input inverting summing circuits to have greater bandwidth, and low-power line drivers to meet the demanding requirements of studio cameras and broadcast video. The output capability of the OPA4684 also sets a new mark in performance for low-power current-feedback amplifiers. Delivering a full ±4Vp-p swing on ±5V supplies, the OPA4684 also has the output current to support > ±3V swing into 50Ω. This minimal output headroom requirement is complemented by a similar 1.2V input stage headroom giving exceptional capability for single +5V operation. The OPA4684’s low 6.8mA supply current is precisely trimmed at 25°C. This trim, along with low shift over temperature and supply voltage, gives a very robust design over a wide range of operating conditions. BW (MHz) vs GAIN 6 V+ + Z(S) IERR V– IERR RF VO Normalized Gain (3dB/div) 1 of 4 Channels 0 –3 Low-Power G=5 –6 –9 –12 G = 10 G = 20 –15 –18 –21 RG G=1 G=2 3 G = 50 G = 100 RF = 800Ω –24 10 Amplifier 100 200 MHz Patent Pending Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. Copyright © 2002, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. www.ti.com ELECTROSTATIC DISCHARGE SENSITIVITY ABSOLUTE MAXIMUM RATINGS(1) Power Supply ............................................................................... ±6.5VDC Internal Power Dissipation ................................. See Thermal Information Differential Input Voltage .................................................................. ±1.2V Input Voltage Range ............................................................................ ±VS Storage Temperature Range: ID, IDBV ......................... –40°C to +125°C Lead Temperature (soldering, 10s) .............................................. +300°C Junction Temperature (TJ ) ........................................................... +175°C ESD Rating: HBM ........................................................................... 2000V CDM ........................................................................... 1500V This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. NOTE: (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. OPA4684 RELATED PRODUCTS SINGLES OPA683 OPA684 OPA691 OPA685 DUALS TRIPLES FEATURES OPA2683 OPA2684 OPA2691 — — OPA3684 OPA3691 — Very Low-Power CFBPLUS Low-Power CFBPLUS High Slew Rate CFB > 500MHz CFB PACKAGE/ORDERING INFORMATION PACKAGE-LEAD PACKAGE DESIGNATOR(1) SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER TRANSPORT MEDIA, QUANTITY OPA4684 SO-14 D –40°C to +85°C OPA4684 " " " " " OPA4684ID OPA4684IDR Rails, 58 Tape and Reel, 2500 OPA4684 TSSOP-14 PW –40°C to +85°C OPA4684 OPA4684IPWT Tape and Reel, 250 " " " " " OPA4684IPWR Tape and Reel, 2500 PRODUCT NOTE: (1) For the most current specifications, and package information, refer to our web site at www.ti.com. PIN CONFIGURATION Top View SO, TSSOP Output A 1 –Input A 2 A 14 Output A 13 –Input A +Input A D +Input A 3 12 +V 4 11 +Input B 5 B –V 10 +Input B C –Input B 6 9 –Input B Output B 7 8 Output B OPA4684 2 OPA4684 www.ti.com SBOS240B ELECTRICAL CHARACTERISTICS: VS = ±5V Boldface limits are tested at +25°C. RF = 800Ω, RL = 100Ω, and G = +2, (see Figure 1 for AC performance only), unless otherwise noted. OPA4684ID, IPW TYP PARAMETER AC PERFORMANCE (See Figure 1) Small-Signal Bandwidth (VO = 0.5Vp-p) Bandwidth for 0.1dB Gain Flatness Peaking at a Gain of +1 Large-Signal Bandwidth Slew Rate Rise-and-Fall Time Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Noninverting Input Current Noise Inverting Input Current Noise Differential Gain Differential Phase All Hostile Crosstalk, Input Referred DC PERFORMANCE(4) Open-Loop Transimpedance Gain (ZOL) Input Offset Voltage Average Offset Voltage Drift Noninverting Input Bias Current Average Noninverting Input Bias Current Drift Inverting Input Bias Current Average Inverting Input Bias Current Drift INPUT Common-Mode Input Range(5) (CMIR) Common-Mode Rejection Ratio (CMRR) Noninverting Input Impedance Inverting Input Resistance (RI) OUTPUT Voltage Output Swing Current Output, Sourcing Current Output, Sinking Closed-Loop Output Impedance POWER SUPPLY Specified Operating Voltage Maximum Operating Voltage Range Max Quiescent Current Min Quiescent Current Power-Supply Rejection Ratio (–PSRR) TEMPERATURE RANGE Specification: ID, IPW Thermal Resistance, θJA D SO-14 PW TSSOP-14 CONDITIONS +25°C G = +1, RF = 800Ω G = +2, RF = 800Ω G = +5, RF = 800Ω G = +10, RF = 800Ω G = +20, RF = 800Ω G = +2, VO = 0.5Vp-p, RF = 800Ω RF = 800Ω, VO = 0.5Vp-p G = +2, VO = 4Vp-p G = –1, VO = 4V Step G = +2, VO = 4V Step G = +2, VO = 0.5V Step G = +2, VO = 4VStep G = +2, f = 5MHz, VO = 2Vp-p RL = 100Ω RL ≥ 1kΩ RL = 100Ω RL ≥ 1kΩ f > 1MHz f > 1MHz f > 1MHz G = +2, NTSC, VO = 1.4Vp, RL = 150Ω G = +2, NTSC, VO = 1.4Vp, RL = 150Ω 3 Channels Driven at 5MHz, 1Vp-p 4th Channel Measured 250 170 138 120 95 19 1.4 90 780 750 3 6.8 VO = 0V, RL = 1kΩ VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to 70°C(2) –40°C to +85°C(2) 120 118 117 16 4.8 14 5.9 14 6.3 675 680 650 660 575 650 –59 –66 –66 –82 4.1 11 18 –59 –65 –65 –81 4.2 12 18.5 160 typ min typ typ typ min max typ min min typ typ C B C C C B B C B B C C –58 –65 –65 –81 4.4 12.5 19 dBc dBc dBc dBc nV/√Hz pA/√Hz pA/√Hz % deg dB max max max max max max max typ typ typ B B B B B B B C C C 155 ±4.6 ±12 ±14.5 ±25 ±18.5 ±35 153 ±4.8 ±12 ±15 ±30 ±19.5 ±40 kΩ mV µV/°C µA nA/°C µA nA°/C min max max max max max max A A B A B A B ±3.65 52 ±3.6 52 V dB kΩ || pF Ω min min typ typ A A C C ±3.9 115 –95 ±3.8 110 –90 V mA mA Ω min min min typ A A A C ±6 7.6 6.2 53 ±6 7.8 5.8 53 V V mA mA dB typ max max min typ C A A A A –40 to +85 °C typ C 100 110 °C/W °C/W typ typ C C –67 –82 –70 –84 3.7 9.4 17 0.04 0.02 –52 ±4.0 ±5.0 ±13 ±5.0 ±17 ±3.65 Open-Loop, DC ±3.75 60 50 || 2 4.0 1kΩ Load VO = 0 VO = 0 G = +2, f = 100kHz ±4.1 160 –120 0.006 ±3.9 ±5 VS = ±5V VS = ±5V Input Referred MIN/ TEST MAX LEVEL(3) MHz MHz MHz MHz MHz MHz dB MHz V/µs V/µs ns ns 355 ±1.5 VCM = 0V UNITS 6.8 6.8 60 Junction-to-Ambient 53 120 –100 ±6 7.2 6.4 54 NOTES: (1) Junction temperature = ambient for 25°C tested specifications. (2) Junction temperature = ambient at low temperature limit, junction temperature = ambient +7°C at high temperature limit for over temperature tested specifications. (3) Test levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out-of-node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ±CMIR limits. OPA4684 SBOS240B www.ti.com 3 ELECTRICAL CHARACTERISTICS: VS = +5V Boldface limits are tested at +25°C. RF = 1kΩ, RL = 100Ω, and G = +2, (see Figure 3 for AC performance only), unless otherwise noted. OPA4684ID, IPW TYP PARAMETER AC PERFORMANCE (See Figure 3) Small-Signal Bandwidth (VO = 0.5Vp-p) Bandwidth for 0.1dB Gain Flatness Peaking at a Gain of +1 Large-Signal Bandwidth Slew Rate Rise-and-Fall Time Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Noninverting Input Current Noise Inverting Input Current Noise Differential Gain Differential Phase All Hostile X-Talk, Input Referred DC PERFORMANCE(4) Open-Loop Transimpedance Gain (ZOL) Input Offset Voltage Average Offset Voltage Drift Noninverting Input Bias Current Average Noninverting Input Bias Current Drift Inverting Input Bias Current Average Inverting Input Bias Current Drift INPUT Least Positive Input Voltage(5) Most Positive Input Voltage(5) Common-Mode Refection Ratio (CMRR) Noninverting Input Impedance Inverting Input Resistance (RI) OUTPUT Most Positive Output Voltage Least Positive Output Voltage Current Output, Sourcing Current Output, Sinking Closed-Loop Output Impedance POWER SUPPLY Specified Single-Supply Operating Voltage Max Single-Supply Operating Voltage Range Max Quiescent Current Min Quiescent Current Power-Supply Rejection Ratio (+PSRR) CONDITIONS +25°C G = +1, RF = 1.3kΩ G = +2, RF = 1.3kΩ G = +5, RF = 1.3kΩ G = +10, RF = 1.3kΩ G = +20, RF = 1.3kΩ G = +2, VO < 0.5Vp-p, RF = 1.3kΩ RF = 1.3kΩ, VO < 0.5Vp-p G = 2, VO = 2Vp-p G = 2, VO = 2V Step G = 2, VO = 0.5V Step G = 2, VO = 2VStep G = 2, f = 5MHz, VO = 2Vp-p RL = 100Ω to VS/2 RL ≥ 1kΩ to VS/2 RL = 100Ω to VS/2 RL ≥ 1kΩ to VS/2 f > 1MHz f > 1MHz f > 1MHz G = +2, NTSC, VO = 1.4Vp, RL = 150Ω G = +2, NTSC, VO = 1.4Vp, RL = 150Ω 3-Channels Driven at 5MHz, 1Vp-p 4th Channel Measured 140 110 100 90 75 21 0.5 86 300 4.3 5.3 VO = VS/2, RL = 1kΩ to VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 VCM = VS/2 –65 –84 –65 –74 3.7 9.4 17 0.04 0.07 –52 MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to 70°C(2) –40°C to +85°C(2) 86 85 82 12 2.6 11 3.4 10 3.7 300 290 280 –60 –62 –64 –70 4.1 11 18 –59 –61 –63 –70 4.2 12 18.5 160 UNITS MIN/ TEST MAX LEVEL(3) MHz MHz MHz MHz MHz MHz dB MHz V/µs ns ns typ min min typ typ min max typ min typ typ C B C C C B B C B C C –59 –61 –63 –69 4.4 12.5 19 dBc dBc dBc dBc nV/ √Hz pA/ √Hz pA/ √Hz % deg dB max max max max max max max typ typ typ B B B B B B B C C C 155 ±4.1 ±12 ±14.5 ±25 ±14.5 ±25 153 ±4.3 ±12 ±15 ±30 ±16 ±30 kΩ mV µV/°C µA nA/°C µA nA°/C min max max max max max max A A B A B A B 355 ±1.0 ±3.5 ±5 ±13 ±5 ±13 1.32 3.68 51 1.35 3.65 50 1.38 3.62 50 Open-Loop 1.25 3.75 58 50 || 1 5 V V dB kΩ || pF Ω max min min typ typ A A A C C RL = 1kΩ to VS/2 RL = 1kΩ to VS/2 VO = VS/2 VO = VS/2 G = +2, f = 100kHz 4.10 0.9 80 70 0.006 3.9 1.1 65 55 3.9 1.1 60 50 3.8 1.2 55 45 V V mA mA Ω min max min min typ A A A A C 12 6.2 5.2 12 6.2 4.8 12 6.2 4.6 V V mA mA dB typ max max min typ C A A A C –40 to +85 °C typ C 100 110 °C/W °C/W typ typ C C VCM = VS/2 5 VS = +5V VS = +5V Input Referred TEMPERATURE RANGE Specification: ID, IPW Thermal Resistance, θJA Junction-to-Ambient D SO-14 PW TSSOP-14 5.8 5.8 58 NOTES: (1) Junction temperature = ambient for 25°C tested specifications. (2) Junction temperature = ambient at low temperature limit, junction temperature = ambient +3°C at high temperature limit for over temperature tested specifications. (3) Test levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out-of-node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ± CMIR limits. 4 OPA4684 www.ti.com SBOS240B ELECTRICAL CHARACTERISTICS: VS = ±5V At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. NONINVERTING SMALL-SIGNAL FREQUENCY RESPONSE VO = 0.5Vp-p RF = 800Ω 3 G=1 G=2 0 –3 –6 G=5 G = 10 –9 G = 20 –12 G = 50 –15 See Figure 1 VO = 0.5Vp-p RF = 800Ω 0 –3 –6 G = 100 –18 See Figure 2 –12 1 10 100 200 1 10 Frequency (MHz) VO = 0.5Vp-p G = –1 RL = 100Ω VO = 0.5Vp-p 0 Gain (dB) Gain (dB) 200 INVERTING LARGE-SIGNAL FREQUENCY RESPONSE 3 G = +2 RL = 100Ω 6 VO = 1Vp-p 3 1Vp-p –3 2Vp-p 5Vp-p –6 VO = 2Vp-p 0 VO = 5Vp-p –9 See Figure 1 –3 See Figure 2 –12 1 10 100 200 1 10 Frequency (MHz) 0.8 G = +2 1.6 Large-Signal Right Scale 0.2 0.8 0.4 Small-Signal Left Scale 0 0 –0.2 –0.4 –0.4 –0.8 –0.6 Output Voltage (200mV/div) 1.2 Output Voltage (400mV/div) G = –1 0.6 –1.2 0.6 1.2 0.4 0.8 0.2 0.4 0 0 Small-Signal Left Scale –0.2 –0.4 Large-Signal Right Scale –0.4 –0.6 See Figure 1 –0.8 –1.2 See Figure 2 –0.8 –1.6 Time (10ns/div) –0.8 –1.6 Time (10ns/div) OPA4684 SBOS240B 200 INVERTING PULSE RESPONSE 1.6 0.4 100 Frequency (MHz) NONINVERTING PULSE RESPONSE 0.8 Output Voltage (200mV/div) 100 Frequency (MHz) NONINVERTING LARGE-SIGNAL FREQUENCY RESPONSE 9 G = –1 G = –2 G = –5 G = –10 G = –16 –9 www.ti.com 5 Output Voltage (400mV/div) Normalized Gain (3dB/div) INVERTING SMALL-SIGNAL FREQUENCY RESPONSE 3 Normalized Gain (3dB/div) 6 ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. HARMONIC DISTORTION vs LOAD RESISTANCE HARMONIC DISTORTION vs FREQUENCY –50 –50 VO = 2Vp-p f = 5MHz G = +2 –60 VO = 2Vp-p RL = 100Ω Harmonic Distortion (dBc) Harmonic Distortion (dBc) –55 –65 2nd-Harmonic –70 –75 3rd-Harmonic –80 –60 2nd-Harmonic –70 3rd-Harmonic –80 –85 See Figure 1 –90 See Figure 1 –90 100 0.1 1k 1 HARMONIC DISTORTION vs OUTPUT VOLTAGE f = 5MHz RL = 100Ω 2nd-Harmonic Harmonic Distortion (dBc) Harmonic Distortion (dBc) –50 –60 –70 3rd-Harmonic –80 –90 0.5 1 VO = 2Vp-p RL = 100Ω –70 3rd-Harmonic –80 –90 ±2.5 5 2nd-Harmonic –60 Output Voltage (Vp-p) ±3 ±3.5 ±4 ±4.5 ±5 Supply Voltage (±V) ±5.5 ±6 HARMONIC DISTORTION vs INVERTING GAIN HARMONIC DISTORTION vs NONINVERTING GAIN –50 –50 –55 –55 2nd-Harmonic Harmonic Distortion (dBc) Harmonic Distortion (dBc) 20 5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE –50 –60 –65 –70 –75 3rd-Harmonic –80 2nd-Harmonic –60 –65 –70 3rd-Harmonic –75 –80 –85 –85 –90 –90 1 10 1 20 10 20 Inverting Gain (V/V) Gain (V/V) 6 10 Frequency (MHz) Load Resistance (Ω) OPA4684 www.ti.com SBOS240B ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. 2-TONE, 3RD-ORDER INTERMODULATION DISTORTION INPUT VOLTAGE AND CURRENT NOISE DENSITY –50 100 20MHz 3rd-Order Spurious Level (dBc) Noninverting Current Noise 9.4pA/√Hz 10 Voltage Noise 3.7nV/√Hz +5V –60 PI 50Ω 50Ω PO OPA4684 10MHz 50Ω –5V 800Ω –70 800Ω 5MHz –80 1MHz –90 1 100 1k 10k 100k 1M 10M –8 –7 –6 –5 –4 –3 –2 –1 0 1 2 3 4 5 Power at Load (each tone, dBm) Frequency (Hz) 9 12pF 0.5dB Peaking 8 5pF 6 Normalized Gain (dB) RS (Ω) 40 30 20 100pF 3 +5V 75pF RS VI 0 VO 50Ω OPA4684 CL 1kΩ –5V 800Ω –3 10 20pF –6 0 1 10 1 100 10 CMRR and PSRR vs FREQUENCY Open-Loop Transimpedance Gain (dBΩ) CMRR 60 50 +PSRR 40 –PSRR 30 20 10 0 104 105 106 Frequency (Hz) 107 108 OPA4684 SBOS240B 300 OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE 70 103 100 Frequency (MHz) CLOAD (pF) 102 50pF 33pF 800Ω Common-Mode Rejection Ratio (dB) 7 SMALL-SIGNAL BANDWIDTH vs CLOAD RS vs CLOAD 50 6 www.ti.com 120 0 20log (ZOL) 100 –30 80 –60 60 –90 ∠ ZOL 40 –120 20 –150 0 –180 102 103 104 105 106 Frequency (Hz) 107 108 109 7 Open-Loop Phase (°) Voltage Noise (nV/√Hz) Current Noise (pA/√Hz) Inverting Current Noise 17pA/√Hz ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. OUTPUT CURRENT AND VOLTAGE LIMITATIONS COMPOSITE VIDEO DIFFERENTIAL GAIN/PHASE 0.07 2 0.06 dG 0.05 =1 00Ω 3 VO (V) 50 Ω 1 0 –1 0.04 –2 0.03 dP 0.02 –3 0.01 –4 0 –5 1 2 3 4 Each Channel –150 1W Power Limit Number of 150Ω Video Loads 0 IO (MA) TYPICAL DC DRIFT OVER AMBIENT TEMPERATURE SUPPLY AND OUTPUT CURRENT vs AMBIENT TEMPERATURE 4 –100 –50 50 100 150 7.6 200 3 2 1 Noninverting Input Bias Current Input Offset Voltage 0 –1 –2 Inverting Input Bias Current 7.2 175 Supply Current 6.8 150 Sinking Output Current 125 6.4 –3 –4 6.0 100 –50 –25 0 25 50 75 100 125 –25 Ambient Temperature (°C) 2V Step See Figure 1 Crosstalk (Input referred) (dB) % Error to Final Value 100 125 –20 0.03 0.02 0.01 0 –0.01 –0.02 –0.03 1Vp-p Output 3-Channels, 100Ω Load –25 –30 –35 –40 –45 –50 –55 –60 –65 –0.04 –70 –0.05 0 8 25 50 75 Ambient Temperature (°C) ALL HOSTILE CROSSTALK SETTLING TIME 0.05 0.04 0 10 20 30 Time (ns) 40 50 0.1 60 1 10 100 Frequency (MHz) OPA4684 www.ti.com SBOS240B Supply Current (mA) Sourcing Output Current Output Current (mA) Input Bias Currents (µA) and Offset Voltage (mV) = RL RL = 500Ω Differential Gain (%) Differential Phase (°) 4 0.08 L Gain = +2 NTSC, Positive Video 0.09 1W Power Limit R 5 0.10 ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.) At TA = +25°C, G = +2, RF = 800Ω, and RL = 100Ω, unless otherwise noted. 8.0 3.2 6.4 6.4 6.4 2.4 4.8 4.8 4.8 1.6 3.2 0.8 1.6 0 0 Output Voltage Right Scale –0.8 –1.6 See Figure 1 –1.6 –2.4 –3.2 –4.8 Input Voltage Left Scale –3.2 –4.0 Input Voltage (1.6V/div) 8.0 Output Voltage (1.6V/div) Input Voltage (0.8V/div) 8.0 3.2 3.2 Output Voltage Right Scale 1.6 0 –1.6 –1.6 –3.2 –3.2 –4.8 –6.4 –6.4 –8.0 –8.0 –4.8 Input Voltage Left Scale –6.4 See Figure 2 –8.0 Time (100ns/div) Time (100ns/div) CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY INPUT AND OUTPUT RANGE vs SUPPLY VOLTAGE 100 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 –6 Input Voltage Range Output Impedance (Ω) Input and Output Voltage Range 1.6 0 Output Voltage Range 1/4 OPA4684 10 ZO 800Ω 800Ω 1 0.01 0.001 ±2 ±3 ±4 ±5 ±6 100 10k 100k 1M 10M 100M Frequency (Hz) Supply Voltage (±V) OPA4684 SBOS240B 1k www.ti.com 9 Output Voltage (1.6V/div) INVERTING OVERDRIVE RECOVERY NONINVERTING OVERDRIVE RECOVERY 4.0 ELECTRICAL CHARACTERISTICS: VS = +5V At TA = +25°C, G = +2, RF = 1kΩ, and RL = 100Ω, unless otherwise noted. NONINVERTING SMALL-SIGNAL FREQUENCY RESPONSE RF = 1kΩ 3 Normalized Gain (3dB/div) INVERTING SMALL-SIGNAL FREQUENCY RESPONSE 3 G = 50 RF = 1.0kΩ G=1 G = 100 Normalized Gain (3dB/div) 6 0 G=2 –3 –6 G = 20 –9 G = 10 –12 0 –3 –6 G = –1 G = –2 G = –5 G = –10 G = –20 –9 –15 G=5 See Figure 3 See Figure 4 –12 –18 1 10 100 1 200 10 NONINVERTING LARGE-SIGNAL FREQUENCY RESPONSE 3 VO = 0.2Vp-p 0.5Vp-p VO = 0.5Vp-p 0 6 1Vp-p Gain (dB) Gain (dB) 200 INVERTING LARGE-SIGNAL FREQUENCY RESPONSE 9 0.2Vp-p 100 Frequency (MHz) Frequency (MHz) 3 2Vp-p VO = 1Vp-p –3 VO = 2Vp-p –6 0 –9 –3 –12 10 100 200 1 10 Frequency (MHz) 200 INVERTING PULSE RESPONSE 1.6 0.4 1.6 0.3 1.2 0.3 1.2 0.2 0.8 0.1 0.4 0.2 Large-Signal Right Scale 0.1 0.8 0.4 Small-Signal Left Scale 0 0 –0.1 –0.4 –0.2 –0.8 –0.3 Output Voltage (200mV/div) 0.4 Output Voltage (400mV/div) Output Voltage (200mV/div) NONINVERTING PULSE RESPONSE –1.2 0 0 Small-Signal Left Scale –0.1 –0.4 Large-Signal Right Scale –0.2 –0.3 See Figure 3. –0.8 –1.2 See Figure 4 –0.4 –1.6 Time (10ns/div) 10 100 Frequency (MHz) –0.4 –1.6 Time (10ns/div) OPA4684 www.ti.com SBOS240B Output Voltage (400mV/div) 1 ELECTRICAL CHARACTERISTICS: VS = +5V (Cont.) At TA = +25°C, G = +2, RF = 1kΩ, and RL = 100Ω, unless otherwise noted. HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs LOAD RESISTANCE –50 –50 Harmonic Distortion (dBc) VO = 2Vp-p f = 5MHz –55 Harmonic Distortion (dBc) VO = 2Vp-p RL = 100Ω –60 3rd-Harmonic –65 –70 –75 –80 2nd-Harmonic –60 2nd-Harmonic –70 3rd-Harmonic –80 –85 See Figure 3 100 0.1 1k HARMONIC DISTORTION vs OUTPUT VOLTAGE 2-TONE, 3RD-ORDER INTERMODULATION DISTORTION 20 –50 2nd-Harmonic –60 3rd-Harmonic –70 –80 20MHz –60 10MHz –70 5MHz –80 See Figure 3 See Figure 3 –90 0.5 1 2 3 –15 –14 –13 –12 –11 –10 –9 Output Voltage (Vp-p) 5.8 –6 –5 –4 –3 0.16 G = +2 NTSC, Positive Video 0.14 Right Scale Supply Current 5.4 Left Scale Sourcing Output Current 30 5.2 Left Scale Sinking Output Current 60 Supply Current (mA) 5.6 Differential Gain (%) Differential Phase (°) 80 –7 COMPOSITE VIDEO DIFFERENTIAL GAIN/PHASE 100 90 –8 Power at Load (each tone, dBm) SUPPLY AND OUTPUT CURRENT vs TEMPERATURE Output Current (mA) 10 Frequency (MHz) 3rd-Order Spurious Level (dBc) Harmonic Distortion (dBc) 1 Load Resistance (Ω) –50 –90 See Figure 3 –90 –90 5.0 0.12 0.10 dP 0.08 0.06 0.04 dG 0.02 50 4.8 –50 –25 0 25 50 75 Ambient Temperature (°C) 100 125 1 2 3 4 Number of 150Ω Video Loads OPA4684 SBOS240B 0 www.ti.com 11 APPLICATIONS INFORMATION LOW-POWER CURRENT-FEEDBACK OPERATION The quad-channel OPA4684 gives a new level of performance in low-power current-feedback op amps. Using a new input stage buffer architecture, the OPA4684 CFBPLUS amplifier holds nearly constant AC performance over a wide gain range. This closed-loop internal buffer gives a very low and linearized impedance at the inverting node, isolating the amplifier’s AC performance from gain element variations. This allows both the bandwidth and distortion to remain nearly constant over gain, moving closer to the ideal current feedback performance of gain bandwidth independence. This low-power amplifier also delivers exceptional output power—it’s ±4V swing on ±5V supplies with > 100mA output drive gives excellent performance into standard video loads or doubly-terminated 50Ω cables. Single +5V supply operation is also supported with similar bandwidths but with reduced output power capability. For lower quiescent power in a CFBPLUS amplifier, consider the OPA683 family; while for higher output power, consider the OPA691 family. mode signal across the input stage, the slew rate for inverting operation is typically higher and the distortion performance is slightly improved. An additional input resistor, RM, is included in Figure 2 to set the input impedance equal to 50Ω. The parallel combination of RM and RG set the input impedance. As the desired gain increases for the inverting configuration, RG is adjusted to achieved the desired gain, while RM is also adjusted to hold a 50Ω input match. A point will be reached where RG will equal 50Ω, RM is removed, and the input match is set by RG only. With RG fixed to achieve an input match to 50Ω, increasing RF will increase the gain. This will, however, quickly reduce the achievable bandwidth as the feedback resistor increases from its recommended value of 800Ω. If the source does not require an input match to 50Ω, either adjust RM to get the desired load, or remove it and let the RG resistor alone provide the input load. +5V 0.1µF Figure 1 shows the DC-coupled, gain of +2, dual powersupply circuit used as the basis of the ±5V Electrical and Typical Characteristics for each channel. For test purposes, the input impedance is set to 50Ω with a resistor to ground and the output impedance is set to 50Ω with a series output resistor. Voltage swings reported in the Electrical Characteristics are taken directly at the input and output pins while load powers (dBm) are defined at a matched 50Ω load. For the circuit of Figure 1, the total effective load will be 100Ω || 1600Ω = 94Ω. Gain changes are most easily accomplished by simply resetting the RG value, holding RF constant at its recommended value of 800Ω. 0.1µF + 50Ω 1/4 OPA4684 50Ω Load RF 800Ω RG 800Ω 0.1µF + 6.8µF –5V FIGURE 1. DC-Coupled, G = +2V/V, Bipolar Supply Specifications and Test Circuit. Figure 2 shows the DC-coupled, gain of –1V/V, dual powersupply circuit used as the basis of the Inverting Typical Characteristics for each channel. Inverting operation offers several performance benefits. Since there is no common- 12 50Ω 1/4 OPA4684 50Ω Load 50Ω Source RG 800Ω RF 800Ω VI RM 53.6Ω 0.1µF + 6.8µF –5V These circuits show ±5V operation. The same circuits can be applied with bipolar supplies from ±2.5V to ±6V. Internal supply independent biasing gives nearly the same performance for the OPA4684 over this wide range of supplies. Generally, the optimum feedback resistor value (for nominally flat frequency response at G = +2) will increase in value as the total supply voltage across the OPA4684 is reduced. 6.8µF VI RM 50Ω 6.8µF FIGURE 2. DC-Coupled, G = –1V/V, Bipolar Supply Specifications and Test Circuit. +5V 50Ω Source + See Figure 3 for the AC-coupled, single +5V supply, gain of +2V/V circuit configuration used as a basis for the +5V only Electrical and Typical Characteristics for each channel. The key requirement of broadband single-supply operation is to maintain input and output signal swings within the usable voltage ranges at both the input and the output. The circuit of Figure 3 establishes an input midpoint bias using a simple resistive divider from the +5V supply (two 10kΩ resistors) to the noninverting input. The input signal is then AC-coupled into this midpoint voltage bias. The input voltage can swing to within 1.25V of either supply pin, giving a 2.5Vp-p input signal range centered between the supply pins. The input impedance of Figure 3 is set to give a 50Ω input match. If the source does not require a 50Ω match, remove this and drive OPA4684 www.ti.com SBOS240B directly into the blocking capacitor. The source will then see the 5kΩ load of the biasing network as a load. The gain resistor (RG) is AC-coupled, giving the circuit a DC gain of +1, which puts the noninverting input DC bias voltage (2.5V) on the output as well. The feedback resistor value has been adjusted from the bipolar ±5V supply condition to re-optimize for a flat frequency response in +5V only, gain of +2, operation. On a single +5V supply, the output voltage can swing to within 1.0V of either supply pin while delivering more than 70mA output current—easily giving a 3Vp-p output swing into 100Ω (8dBm maximum at the matched 50Ω load). The circuit of Figure 3 shows a blocking capacitor driving into a 50Ω output resistor, then into a 50Ω load. Alternatively, the blocking capacitor could be removed if the load is tied to a supply midpoint or to ground if the DC current required by the load is acceptable. The circuits of Figure 3 and 4 show single-supply operation at +5V. These same circuits may be used up to single supplies of +12V with minimal change in the performance of the OPA4684. +5V 0.1µF + 6.8µF 10kΩ 1/4 0.1µF 50Ω 10kΩ OPA4684 0.1µF 50Ω Load 50Ω Source DIS RF 1.3kΩ RG 0.1µF 1.3kΩ VI RM 52.3Ω +5V 0.1µF 50Ω Source + FIGURE 4. AC-Coupled, G = –1V/V, Single-Supply Specifications and Test Circuit. 6.8µF 10kΩ 0.1µF VI RM 50Ω 10kΩ 1/4 OPA4684 DIFFERENTIAL INTERFACE APPLICATIONS 0.1µF 50Ω 50Ω Load RF 1kΩ RG 1kΩ 0.1µF Dual and quad op amps are particularly suitable to differential input to differential output applications. Typically, these fall into either ADC input interface or line driver applications. Two basic approaches to differential I/O are noninverting or inverting configurations. Since the output is differential, the signal polarity is somewhat meaningless—the noninverting and inverting terminology applies here to where the input is brought into the OPA4684. Each has its advantages and disadvantages. Figure 5 shows a basic starting point for noninverting differential I/O applications. FIGURE 3. AC-Coupled, G = +2V/V, Single-Supply Specifications and Test Circuit. Figure 4 shows the AC-coupled, single +5V supply, gain of –1V/V circuit configuration used as a basis for the inverting +5V only Typical Characteristics for each channel. In this case, the midpoint DC bias on the noninverting input is also decoupled with an additional 0.1µF capacitor. This reduces the source impedance at higher frequencies for the noninverting input bias current noise. This 2.5V bias on the noninverting input pin appears on the inverting input pin and, since RG is DC-blocked by the input capacitor, will also appear at the output pin. One advantage to inverting operation is that since there is no signal swing across the input stage, higher slew rates and operation to even lower supply voltages is possible. To retain a 1Vp-p output capability, operation down to a 3V supply is allowed. At a +3V supply, the input stage is saturated, but for the inverting configuration of a current-feedback amplifier, wideband operation is retained even under this condition. +VCC 1/4 OPA4684 RF 800Ω VI RF 800Ω VO 1/4 OPA4684 –VCC FIGURE 5. Noninverting Differential I/O Amplifier. OPA4684 SBOS240B RG www.ti.com 13 This approach provides for a source termination impedance that is independent of the signal gain. For instance, simple differential filters may be included in the signal path right up to the noninverting inputs without interacting with the amplifier gain. The differential signal gain for the circuit of Figure 5 is: VO/VI = AD = 1 + 2 • RF/RG Since the OPA4684 is a CFBPLUS amplifier, its bandwidth is principally controlled with the feedback resistor value; Figure 5 shows the recommended value of 800Ω. However, the gain may be adjusted with considerable freedom using just the RG resistor. In fact, RG may be a reactive network providing a very isolated shaping to the differential frequency response. Since the inverting inputs of the OPA4684 are very low impedance closed-loop buffer outputs, the RG element does not interact with the amplifier’s bandwidth—wide ranges of resistor values and/or filter elements may be inserted here with minimal amplifier bandwidth interaction. Various combinations of single-supply or AC-coupled gains can also be delivered using the basic circuit of Figure 5. Common-mode bias voltages on the two noninverting inputs pass on to the output with a gain of 1 since an equal DC voltage at each inverting node creates no current through RG, giving that voltage a common-mode gain of 1 to the output. Figure 6 shows a differential I/O stage configured as an inverting amplifier. In this case, the gain resistors (RG) become the input resistance for the source. This provides a better noise performance than the noninverting configuration but does limit the flexibility in setting the input impedance separately from the gain. The two noninverting inputs provide an easy common-mode control input. This is particularly useful if the source is ACcoupled through either blocking caps or a transformer. In either case, the common-mode input voltages on the two noninverting inputs again have a gain of 1 to the output pins, giving an easy common-mode control for single-supply operation. The OPA4684 used in this configuration does constrain the feedback to the 800Ω region for best frequency response. With RF fixed, the input resistors may be adjusted to the desired gain but will also be changing the input impedance as well. The differential gain for this circuit is: VO/VI = –RF/RG LOW-POWER VIDEO LINE DRIVER APPLICATIONS For low-power, video line driving, the OPA4684 provides the output current and linearity to support 4 channels of either single video lines, or up to 4 video lines in parallel on each output. Figure 7 shows a typical ±5V supply video line driver application where only one channel is shown and only a single line is being driven. The improved 2nd-harmonic distortion of the CFBPLUS architecture, along with the OPA684’s high output current and voltage, gives exceptional differential gain and phase performance for a low-power solution. As the Typical Characteristics show, a single video load shows a dG/dP of 0.04%/0.02°. Multiple loads may be driven on each output, with minimal x-talk, while the dG/dP is still < 0.1%/0.1° for up to 4 parallel video loads. +5V VIDEOIN Supply decoupling not shown. 75Ω 1/4 OPA4684 +VCC Coax 75Ω Load 800Ω VCM 800Ω 1/4 OPA4684 RG 75Ω –5V FIGURE 7. Gain of +2 Video Cable Driver. RF 800Ω DC-COUPLED SINGLE-TO-DIFFERENTIAL CONVERSION VI RG RF 800Ω 1/4 OPA4684 VCM –VCC VO The previous differential output circuits were set up to receive a differential input as well as provide a differential output. Figure 8 shows one way to provide a single to differential conversion, with DC coupling, and independent output common-mode control using a quad op amp. The circuit of Figure 8 provides several useful features for isolating the input signal from the final outputs. Using the first amplifier as a simple noninverting stage gives an independent adjustment on RI (to set the source loading) while the gain can be easily adjusting in this stage using the RG resistor. Bandwidth is relatively independent of gain setting in the OPA4684. The next stage allows a separate output FIGURE 6. Inverting Differential I/O Amplifier. 14 OPA4684 www.ti.com SBOS240B common-mode level to be set up. The desired output common-mode voltage, VCM, is cut in half and applied to the noninverting input of the 2nd stage. The signal path in this stage sees a gain of –1 while this (1/2 • VCM) voltage sees a gain of +2. The output of this 2nd stage is then the original common-mode voltage plus the inverted signal from the output of the first stage. The output of this 2nd stage then appears directly at the output of the noninverting final stage. The inverting node of the inverting output stage is also biased to the common-mode voltage, equal to the commonmode voltage appearing at the output of the 2nd stage, creating no current flow and placing the desired VCM at the output of this stage as well. differential active filters. Where the filter topology is looking for a simple gain function to implement the filter, the noninverting configuration is preferred to isolate the filter elements from the gain elements in the design. Figure 9 shows an example of a very low-power, 10MHz, 4th-order Butterworth, low-pass Sallen-Key filter. Often, these filters are designed at an amplifier gain of 1 to minimize amplifier bandwidth interaction with the desired filter shape. Since the OPA4684 shows minimal bandwidth change with gain, this would not be a constraint in this design. The example of Figure 9 designs the filter for a differential gain of 4 in each differential stage. This DC-coupled design gives a signal gain of 16V/V in the passband with a f–3dB at 10MHz. The design places the higher Q stage first to allow the lower Q 2nd stage to roll off the peaked noise of the first stage. The resistor values have been adjusted slightly to account for the amplifier group delay. LOW-POWER, DIFFERENTIAL I/O, 4th-ORDER ACTIVE FILTER The OPA4684 can give a very capable gain block for lowpower active filters. The quad design lends itself very well to +5V VCM 1/4 OPA4684 1kΩ VCM 2 VI RI 1/4 OPA4684 0.1µF 1kΩ 800Ω 800Ω 1/4 OPA4684 800Ω 800Ω 800Ω 800Ω ) RG –VI (1 + 800Ω ) RG VCM 50Ω 1/4 OPA4684 RG +VI (1 + 800Ω –5V FIGURE 8. High Gain, DC-Coupled, Single to Differential Conversion. 100pF 66.5Ω 100pF 249Ω 49.9Ω +5V 392Ω VO/VI = 16V/V f–3dB = 10MHz 1/4 OPA4684 1/4 OPA4684 800Ω VI 50pF 800Ω 534Ω 249Ω 1/4 OPA4684 VO 534Ω 50pF 800Ω 66.5Ω PD = 68mW 800Ω 49.9Ω 1/4 OPA4684 392Ω 100pF 100pF GD = 4, WO = 2π 10MHz, Q = 1.31 –5V GD = 4, WO = 2π 10MHz, Q = 0.54 FIGURE 9. Low-Power, Differential I/O, 4th-Order Butterworth Active Filter. OPA4684 SBOS240B www.ti.com 15 While this circuit is bipolar, using ±5V supplies, it can easily be adapted to single-supply operation. This is typically done by providing a supply midpoint reference at the noninverting inputs then adding DC-blocking caps at each input and in series with the amplifier gain resistor, RG. This will add two real zeroes in the response, transforming this circuit into a bandpass. Figure 10 shows the frequency response for the filter of Figure 9. 27 Differential Gain (dB) 24 21 18 15 12 9 LOW-POWER DSL TRANSCEIVER INTERFACE 6 With four amplifiers available, the quad OPA4684 can meet the needs for both differential driver and receiver in a lowpower xDSL line interface design. Figure 11 shows a simplified design example. Two amplifiers are used as a noninverting differential driver while the other two implement the driver echo cancellation and receiver amplifier function. This example shows a single +12V design where the drive side is taking a 2Vp-p maximum input from the transmit filter and providing a differential gain of 7, giving a maximum 14Vp-p differential output swing. This is coupled through 50Ω matching resistors and a 1:1 transformer to give a maximum 7Vp-p on a 100Ω line. This 7Vp-p corresponds to a 10dBm line power with a 3.5 crest factor. 3 The differential receiver is configured as an inverting summing stage where the outputs of the driver are cancelled prior to appearing at the output of the receive amplifiers. This is done by summing the output voltages for the drive amplifiers 0 1 10 Frequency (MHz) 100 FIGURE 10. Low-Power, Differential 4th-Order, 100MHz Butterworth. and their attenuated and inverted levels (at the transformer input) into the inverting inputs of each receiver amplifier. The resistor values are set in Figure 11 to give perfect drive signal cancellation if the drive signal is attenuated by 1/2 going from the drive amplifier outputs to the transformer input. The signal received through the transformer has a gain of 1 through the receive amplifiers. Higher gain could easily be provided by scaling the resistors summing into the inverting inputs of the receiver amplifiers down while keeping the same ratio between them. +12V 1/4 OPA4684 10kΩ 900Ω Transmit Filter 50Ω 1:1 +6V 2Vp-p VCM 100Ω Line 14Vp-p 300Ω 10dBm 3.5 Crest Factor 50Ω 900Ω 10kΩ 1/4 OPA4684 1.6kΩ AFE 1.6kΩ 800Ω 800Ω VCM 1/4 OPA4684 800Ω Receiver Filter 800Ω 1/4 OPA4684 VCM FIGURE 11. Low-Power, XDSL Transceiver Design. 16 OPA4684 www.ti.com SBOS240B DUAL-CHANNEL, DIFFERENTIAL ADC DRIVER The circuit of Figure 12 is particularly suitable for a moderate resolution dual ADC used as I/Q samplers. The optional 500Ω resistors to ground on each amplifier output can be added to improve the 2nd- and 3rd-harmonic distortion by > 15dB if higher dynamic range is required. Figure 13 shows the harmonic distortion for the circuit of Figure 12 with and without these pull-down resistors. The 5mA added output stage current significantly improves linearity if that is required. The measured 2nd-harmonic distortion is consistently lower than the 3rd-harmonics for this balanced differential design. It is particularly helpful for this low-power design if there are no grounds in the signal path after the lowlevel signal at the transformer input. The two pull-down resistors do show a signal path ground and should be connected at the same physical point to ground to eliminate imbalanced ground return currents from degrading 2ndharmonic distortion. –50 2Vp-p Output 3rd-Harmonic Harmonic Distortion (dBc) Where a low-power, single-supply, interface to a differential input +5V ADC is required, the circuit of Figure 12 can provide a high dynamic range, medium gain interface for dual high-performance ADCs. The circuit of Figure 12 uses two amplifiers in the differential inverting configuration. The common-mode voltage is set on the noninverting inputs to the supply midscale. In this example, the input signal is coupled in through a 1:2 transformer. This provides both signal gain, single to differential conversion, and a reduction in noise figure. To show a 50Ω input impedance at the input to the transformer, two 200Ω resistors are required on the transformer secondary. These two resistors are also the amplifier gain elements. Since the same DC voltage appears on both inverting nodes in the circuit of Figure 12, no DC current will flow through the transformer, giving a DC gain of 1 to the output for this common-mode voltage, VCM. –60 2nd-Harmonic –70 No Pull-Down 3rd-Harmonic –80 2nd-Harmonic 5mA/Ch Pull-Down –90 1 10 20 Frequency (MHz) FIGURE 13. Harmonic Distortion vs Frequency. DESIGN-IN TOOLS DEMONSTRATION BOARDS Two PC boards are available to assist in the initial evaluation of circuit performance using the OPA4684 in its two package styles. Both of these are available, free, as an unpopulated PC board delivered with descriptive documentation. The summary information for these boards is shown in Table I below. PRODUCT PACKAGE BOARD PART NUMBER OPA4684ID OPA4684IDBW SO-14 TSSOP-14 DEM-OPA468xD DEM-OPA46xxDPW LITERATURE REQUEST NUMBER SBOU016 — TABLE I. Evaluation Module Ordering Information. +5V 10kΩ VCM 0.1µF 10kΩ 1/4 OPA4684 Dual ADC 500Ω 1:2 200Ω 800Ω RS 200Ω 800Ω RS 50Ω Source CL 14.7dB Noise Figure 1 of 2 Channels 1/4 OPA4684 VCM 500Ω Gain = 8V/V 18.1dB FIGURE 12. Single-Supply Differential ADC Driver (1 of 2 Channels). OPA4684 SBOS240B www.ti.com 17 MACROMODELS Computer simulation of circuit performance using SPICE is often useful in predicting the performance of analog circuits and systems. This is particularly true for Video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. Check the TI web site (www.ti.com) for SPICE macromodels within the OPA4684 product folder. These models do a good job of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do not do as well in predicting distortion or dG/dP characteristics. These models do not attempt to distinguish between the package types in their small-signal AC performance. OPERATING SUGGESTIONS SETTING RESISTOR VALUES TO OPTIMIZE BANDWIDTH approaching 1.00 and this shows up in a slightly higher CMRR than previous current-feedback op amps. RI, the buffer output impedance, is a critical portion of the bandwidth control equation. The OPA4684 reduces this element to approximately 4.0Ω using the local loop gain of the input buffer stage. This significant reduction in output impedance, on very low power, contributes significantly to extending the bandwidth at higher gains. A current-feedback op amp senses an error current in the inverting node (as opposed to a differential input error voltage for a voltage-feedback op amp) and passes this on to the output through an internal frequency-dependent transimpedance gain. The Typical Characteristics show this open-loop transimpedance response. This is analogous to the open-loop voltage gain curve for a voltage-feedback op amp. Developing the transfer function for the circuit of Figure 13 gives Equation 1: (1) Any current-feedback op amp like the OPA4684 can hold high bandwidth over signal-gain settings with the proper adjustment of the external resistor values. A low-power part like the OPA4684 typically shows a larger change in bandwidth due to the significant contribution of the inverting input impedance to loop-gain changes as the signal gain is changed. Figure 14 shows a simplified analysis circuit for any currentfeedback amplifier. VO = VI R α 1 + F RG α NG = R F + RI NG R RF + RI 1 + F 1 + Z (S ) RG 1+ Z (S ) R NG = 1 + F R G VI This is written in a loop-gain analysis format where the errors arising from a non-infinite open-loop gain are shown in the denominator. If Z(S) were infinite over all frequencies, the denominator of Equation 1 would reduce to 1 and the ideal desired signal gain shown in the numerator would be achieved. The fraction in the denominator of Equation 1 determines the frequency response. Equation 2 shows this as the loop-gain equation. α VO RI iERR Z(S) iERR RF RG Z (S ) RF + RI NG FIGURE 14. Current-Feedback Transfer Function Analysis Circuit. The key elements of this current-feedback op amp model are: α ⇒ Buffer gain from the noninverting input to the inverting input RI ⇒ Buffer output impedance iERR ⇒ Feedback error current signal Z(S) ⇒ Frequency-dependent open-loop transimpedance gain from iERR to VO The buffer gain is typically very close to 1.00 and is normally neglected from signal gain considerations. It will, however, set the CMRR for a single op amp differential amplifier configuration. For the buffer gain α < 1.0, the CMRR = –20 • log(1 – α). The closed-loop input stage buffer used in the OPA4684 gives a buffer gain more closely 18 = Loop Gain (2) If 20 • log(RF + NG • RI) were drawn on top of the open-loop transimpedance plot, the difference between the two would be the loop gain at a given frequency. Eventually, Z(S) rolls off to equal the denominator of Equation 2 at which point the loop gain has reduced to 1 (and the curves have intersected). This point of equality is where the amplifier’s closed-loop frequency response given by Equation 1 will start to roll off, and is exactly analogous to the frequency at which the noise gain equals the open-loop voltage gain for a voltage feedback op amp. The difference here is that the total impedance in the denominator of Equation 2 may be controlled somewhat separately from the desired signal gain (or NG). The OPA4684 is internally compensated to give a maximally flat frequency response for RF = 800Ω at NG = 2 on ±5V supplies. That optimum value goes to 1.0kΩ on a single +5V supply. Normally, with a current-feedback amplifier, it is OPA4684 www.ti.com SBOS240B possible to adjust the feedback resistor to hold this bandwidth up as the gain is increased. The CFBPLUS architecture has reduced the contribution of the inverting input impedance to provide exceptional bandwidth to higher gains without adjusting the feedback resistor value. The Typical Characteristics show the small-signal bandwidth over gain with a fixed feedback resistor. Putting a closed-loop buffer between the noninverting and inverting inputs does bring some added considerations. Since the voltage at the inverting output node is now the output of a locally closed-loop buffer, parasitic external capacitance on this node can cause frequency response peaking for the transfer function from the noninverting input voltage to the inverting node voltage. While it is always important to keep the inverting node capacitance low for any current-feedback op amp, it is critically important for the OPA4684. External layout capacitance in excess of 2pF will start to peak the frequency response. This peaking can be easily reduced by then increasing the feedback resistor value—but it is preferable, from a noise and dynamic range standpoint, to keep that capacitance low, allowing a close to nominal 800Ω feedback resistor for flat frequency response. Very high parasitic capacitance values on the inverting node (> 5pF) can possibly cause input stage oscillation that cannot be filtered by a feedback element adjustment. At very high gains, 2nd-order effects in the inverting output impedance cause the overall response to peak up. If desired, it is possible to retain a flat frequency response at higher gains by adjusting the feedback resistor to higher values as the gain is increased. Since the exact value of feedback that will give a flat frequency response depends strongly in inverting and output node parasitic capacitance values, it is best to experiment in the specific board with increasing values until the desired flatness (or pulse response shape) is obtained. In general, increasing RF (and adjusting RG then to the desired gain) will move towards flattening the response, while decreasing it will extend the bandwidth at the cost of some peaking. OUTPUT CURRENT AND VOLTAGE The OPA4684 provides output voltage and current capabilities that can support the needs of driving doubly-terminated 50Ω lines. For a 100Ω load at the gain of +2, (see Figure 1), the total load is the parallel combination of the 100Ω load and the 1.6kΩ total feedback network impedance. This 94Ω load will require no more than 40mA output current to support the ±3.8V minimum output voltage swing specified for 100Ω loads. This is well under the specified minimum +110mA/–90mA output current specifications over the full temperature range. The specifications described above, though familiar in the industry, consider voltage and current limits separately. In many applications, it is the voltage • current, or V-I product, which is more relevant to circuit operation. Refer to the “Output Voltage and Current Limitations” curve in the Typical Characteristics. The X- and Y-axes of this graph show the zero-voltage output current limit and the zero-current output voltage limit, respectively. The four quadrants give a more detailed view of the OPA4684’s output drive capabilities. Superimposing resistor load lines onto the plot shows the available output voltage and current for specific loads. The minimum specified output voltage and current over temperature are set by worst-case simulations at the cold temperature extreme. Only at cold startup will the output current and voltage decrease to the numbers shown in the Electrical Characteristic tables. As the output transistors deliver power, their junction temperatures will increase, decreasing their VBE’s (increasing the available output voltage swing) and increasing their current gains (increasing the available output current). In steady-state operation, the available output voltage and current will always be greater than that shown in the over temperature specifications since the output stage junction temperatures will be higher than the minimum specified operating ambient. To maintain maximum output stage linearity, no output shortcircuit protection is provided. This will not normally be a problem since most applications include a series-matching resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground. However, shorting the output pin directly to a power-supply pin will, in most cases, destroy the amplifier. If additional short-circuit protection is required, consider a small-series resistor in the power-supply leads. This will, under heavy output loads, reduce the available output voltage swing. A 5Ω series resistor in each power-supply lead will limit the internal power dissipation to less than 1W for an output short-circuit while decreasing the available output voltage swing only 0.25V for up to 50mA desired load currents. This slight drop in available swing is more if multiple channels are driving heavy loads simultaneously. Always place the 0.1µF powersupply decoupling capacitors after these supply current limiting resistors directly on the supply pins. DRIVING CAPACITIVE LOADS One of the most demanding, and yet very common load conditions, for an op amp is capacitive loading. Often, the capacitive load is the input of an ADC—including additional external capacitance which may be recommended to improve ADC linearity. A high-speed, high open-loop gain amplifier like the OPA4684 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the amplifier’s open-loop output resistance is considered, this capacitive load introduces an additional pole in the signal path that can decrease the phase margin. Several external solutions to this problem have been suggested. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the phase margin and improving stability. OPA4684 SBOS240B www.ti.com 19 The Typical Characteristics show the recommended “RS vs CLOAD” and the resulting frequency response at the load. The 1kΩ resistor shown in parallel with the load capacitor is a measurement path and may be omitted. Parasitic capacitive loads greater than 5pF can begin to degrade the performance of the OPA4684. Long PC board traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always consider this effect carefully, and add the recommended series resistor as close as possible to the OPA4684 output pin (see Board Layout Guidelines). DISTORTION PERFORMANCE The OPA4684 provides very low distortion in a low-power part. The CFBPLUS architecture also gives two significant areas of distortion improvement. First, in operating regions where the 2nd-harmonic distortion due to output stage nonlinearities is very low (frequencies < 1MHz, low output swings into light loads) the linearization at the inverting node provided by the CFBPLUS design gives 2nd-harmonic distortions that extend into the –90dBc region. Previous currentfeedback amplifiers have been limited to approximately –85dBc due to the nonlinearities at the inverting input. The second area of distortion improvement comes in a distortion performance that is largely gain independent. To the extent that the distortion at a particular output power is output stage dependent, 3rd-harmonics particularly, and to a lesser extend 2nd-harmonic distortion, is constant as the gain is increased. This is due to the constant loop gain versus signal gain provided by the CFBPLUS design. As shown in the Typical Characteristic curves, while the 3rd-harmonic is constant with gain, the 2nd-harmonic degrades at higher gains. This is largely due to board parasitic issues. Slightly imbalanced load return currents will couple into the gain resistor to cause a portion of the 2nd-harmonic distortion. At high gains, this imbalance has more gain to the output giving reduced 2nd-harmonic distortion. Differential stages using two of the channels together can reduce this 2nd-harmonic issue enormously getting back to an essentially gain independent distortion. Relative to alternative amplifiers with < 2mA/ch supply current, the OPA4684 holds much lower distortion at higher frequencies (> 5MHz) and to higher gains. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd-harmonic will dominate the distortion with a lower 3rd-harmonic component. Focusing then on the 2ndharmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network—in the noninverting configuration (see Figure 1) this is the sum of RF + RG, while in the inverting configuration it is just RF. Also, providing an additional supply decoupling capacitor (0.1µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3dB to 6dB). In most op amps, increasing the output voltage swing increases harmonic distortion directly. A low-power part like the OPA4684 includes quiescent boost circuits to provide the large-signal bandwidth in the Electrical Characteristics. These act to increase the bias in a very linear fashion only when 20 high slew rate or output power are required. This also acts to actually reduce the distortion slightly at higher output power levels. The Typical Characteristic curves show the 2ndharmonic holding constant from 500mVp-p to 5Vp-p outputs while the 3rd-harmonics actually decrease with increasing output power. The OPA4684 has an extremely low 3rd-order harmonic distortion, particularly for light loads and at lower frequencies. This also gives low 2-tone, 3rd-order intermodulation distortion as shown in the Typical Characteristic curves. Since the OPA4684 includes internal power boost circuits to retain good full-power performance at high frequencies and outputs, it does not show a classical 2-tone, 3rd-order intermodulation intercept characteristic. Instead, it holds relatively low and constant 3rd-order intermodulation spurious levels over power. The Typical Characteristic curves show this spurious level as a dBc below the carrier at fixed center frequencies swept over single-tone power at a matched 50Ω load. These spurious levels drop significantly (> 12dB) for lighter loads than the 100Ω used in the “2-Tone, 3rd-Order Intermodulation Distortion” curve. Converter inputs for instance will see < –82dBc 3rd-order spurious to 10MHz for full-scale inputs. For even lower 3rd-order intermodulation distortion to much higher frequencies, consider the OPA2691 dual or OPA691 and OPA685 single-channel current-feedback amplifiers. NOISE PERFORMANCE Wideband current-feedback op amps generally have a higher output noise than comparable voltage-feedback op amps. The OPA4684 offers an excellent balance between voltage and current noise terms to achieve low output noise in a lowpower amplifier. The inverting current noise (17pA/√Hz) is comparable to most other current-feedback op amps while the input voltage noise (3.7nV/√Hz) is lower than any unitygain stable, comparable slew rate, voltage-feedback op amp. This low input voltage noise was achieved at the price of higher noninverting input current noise (9.4pA/√Hz). As long as the AC source impedance looking out of the noninverting node is less than 200Ω, this current noise will not contribute significantly to the total output noise. The op amp input voltage noise and the two input current noise terms combine to give low output noise under a wide variety of operating conditions. Figure 15 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density ENI 1/4 OPA4684 RS EO IBN RF ERS √ 4kTRS RG 4kT RG IBI √ 4kTRF 4kT = 1.6E –20J at 290°K FIGURE 15. Op Amp Noise Analysis Model. OPA4684 www.ti.com SBOS240B terms in either nV/√Hz or pA/√Hz. The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 3 shows the general form for the output noise voltage using the terms presented in Figure 12. (3) error contribution to the output is ineffective. Evaluating the configuration of Figure 1, using worst-case +25°C input offset voltage and the two input bias currents, gives a worst-case output offset range equal to: ±(NG • VOS(MAX)) + (IBN • RS/2 • NG) ± (IBI • RF) where NG = noninverting signal gain = ±(2 • 4.0mV) ± (13µA • 25Ω • 2) ± (800Ω • 17µA) EO = ENI2 + (IBNR S ) + 4kTRS GN2 + (IBIRF ) + 4kTRF GN = ±8mV + 0.65mV ± 13.6mV Dividing this expression by the noise gain (GN = (1+RF/RG)) will give the equivalent input referred spot noise voltage at the noninverting input, as shown in Equation 4. While the last term, the inverting bias current error, is dominant in this low-gain circuit, the input offset voltage will become the dominant DC error term as the gain exceeds 5V/V. Where improved DC precision is required in a highspeed amplifier, consider the OPA656 single and OPA2822 dual voltage-feedback amplifiers. 2 2 (4) 2 I R 4kTRF 2 EN = ENI2 + (IBNR S ) + 4kTRS + BI F + GN GN = ±22.3mV THERMAL ANALYSIS Evaluating these two equations for the OPA4684 circuit and component values presented in Figure 1 will give a total output spot noise voltage of 16.3nV/√Hz and a total equivalent input spot noise voltage of 8.1nV/√Hz. This total input referred spot noise voltage is higher than the 3.7nV/√Hz specification for the op amp voltage noise alone. This reflects the noise added to the output by the inverting current noise times the feedback resistor. As the gain is increased, this fixed output noise power term contributes less to the total output noise and the total input referred voltage noise given by Equation 3 will approach just the 3.7nV/√Hz of the op amp itself. For example, going to a gain of +20 in the circuit of Figure 1, adjusting only the gain resistor to 42.1Ω, will give a total input referred noise of 3.9nV/√Hz. A more complete description of op amp noise analysis can be found in the Texas Instruments application note, AB-103, “Noise Analysis for High Speed Op Amps” (SBOA066), located at www.ti.com. DC ACCURACY AND OFFSET CONTROL A current-feedback op amp like the OPA4684 provides exceptional bandwidth in high gains, giving fast pulse settling but only moderate DC accuracy. The Electrical Characteristics show an input offset voltage comparable to high slew rate voltage-feedback amplifiers. However, the two input bias currents are somewhat higher and are unmatched. Whereas bias current cancellation techniques are very effective with most voltage-feedback op amps, they do not generally reduce the output DC offset for wideband currentfeedback op amps. Since the two input bias currents are unrelated in both magnitude and polarity, matching the source impedance looking out of each input to reduce their The OPA4684 will not require external heatsinking or airflow most applications. Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 175°C. Operating junction temperature (TJ) is given by TA + PD • θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum when the output is fixed at a voltage equal to 1/2 either supply voltage (for equal bipolar supplies). Under this condition PDL = VS2/(4 • RL) where RL includes feedback network loading. Note that it is the power in the output stage and not into the load that determines internal power dissipation. As an absolute worst-case example, compute the maximum TJ using an OPA4684IPW (TSSOP-14 package) in the circuit of Figure 1 operating at the maximum specified ambient temperature of +85°C with all channels driving a grounded 100Ω load to 2.5VDC. PD = 10V • 7.8mA + 4 • (52 /(4 • (100Ω 1.6kΩ))) = 144mW Maximum TJ = +85°C + (0.144W • 110°C/W) = 101°C. This maximum operating junction temperature is well below most system level targets. Most applications will be lower than this since an absolute worst-case output stage power was assumed in this calculation with all 4 channels running maximum output power simultaneously. OPA4684 SBOS240B www.ti.com 21 BOARD LAYOUT GUIDELINES design. Note that a 800Ω feedback resistor, rather than a direct short, is required for the unity-gain follower application. A current-feedback op amp requires a feedback resistor even in the unity-gain follower configuration to control stability. Achieving optimum performance with a high-frequency amplifier like the OPA4684 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability; on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25") from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.01µF) across the two power supplies (for bipolar operation) will improve 2nd-harmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequencies, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. c) Careful selection and placement of external components will preserve the high-frequency performance of the OPA4684. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal film and carbon composition axially-leaded resistors can also provide good highfrequency performance. Again, keep their leads and PC-board trace length as short as possible. Never use wirewound type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. The quad amplifier pinout allows each output and inverting input to be connected by the feedback element with virtually no trace length. Other network components, such as noninverting input termination resistors, should also be placed close to the package. The frequency response is primarily determined by the feedback resistor value as described previously. Increasing its value will reduce the peaking at higher gains, while decreasing it will give a more peaked frequency response at lower gains. The 800Ω feedback resistor used in the Typical Characteristics at a gain of +2 on ±5V supplies is a good starting point for 22 d) Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of recommended “R S vs C LOAD ”. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA4684 is nominally compensated to operate with a 2pF parasitic load. If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on board, and in fact a higher impedance environment will improve distortion, see the distortion versus load plots. With a characteristic board trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the output of the OPA4684 is used, as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. The high output voltage and current capability of the OPA4684 allows multiple destination devices to be handled as separate transmission lines, each with their own series and shunt terminations. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the plot of “RS vs CLOAD”. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is LOW, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. e) Socketing a high-speed part like the OPA4684 is not recommended. The additional lead length and pin-topin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA4684 onto the board. INPUT AND ESD PROTECTION OPA4684 www.ti.com SBOS240B The OPA4684 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table where an absolute maximum 13V across the supply pins is reported. All device pins have limited ESD protection using internal diodes to the power supplies, as shown in Figure 16. These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (e.g. in systems with ±15V supply parts driving into the OPA4684), current limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. +VCC External Pin –VCC FIGURE 16. Internal ESD Protection. OPA4684 SBOS240B Internal Circuitry www.ti.com 23 PACKAGE DRAWINGS MSOI002B – JANUARY 1995 – REVISED SEPTEMBER 2001 D (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE 8 PINS SHOWN 0.020 (0,51) 0.014 (0,35) 0.050 (1,27) 8 0.010 (0,25) 5 0.008 (0,20) NOM 0.244 (6,20) 0.228 (5,80) 0.157 (4,00) 0.150 (3,81) Gage Plane 1 4 0.010 (0,25) 0°– 8° A 0.044 (1,12) 0.016 (0,40) Seating Plane 0.010 (0,25) 0.004 (0,10) 0.069 (1,75) MAX PINS ** 0.004 (0,10) 8 14 16 A MAX 0.197 (5,00) 0.344 (8,75) 0.394 (10,00) A MIN 0.189 (4,80) 0.337 (8,55) 0.386 (9,80) DIM 4040047/E 09/01 NOTES: A. B. C. D. 24 All linear dimensions are in inches (millimeters). This drawing is subject to change without notice. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15). Falls within JEDEC MS-012 OPA4684 www.ti.com SBOS240B PACKAGE DRAWINGS (Cont.) MTSS001C – JANUARY 1995 – REVISED FEBRUARY 1999 PW (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE 14 PINS SHOWN 0,30 0,19 0,65 14 0,10 M 8 0,15 NOM 4,50 4,30 6,60 6,20 Gage Plane 0,25 1 7 0°– 8° A 0,75 0,50 Seating Plane 0,15 0,05 1,20 MAX PINS ** 0,10 8 14 16 20 24 28 A MAX 3,10 5,10 5,10 6,60 7,90 9,80 A MIN 2,90 4,90 4,90 6,40 7,70 9,60 DIM 4040064/F 01/97 NOTES: A. B. C. D. All linear dimensions are in millimeters. This drawing is subject to change without notice. Body dimensions do not include mold flash or protrusion not to exceed 0,15. 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