OPA2846 SBOS274A −JUNE 2003 − REVISED MARCH 2004 Dual, Wideband, Low-Noise, Voltage-Feedback Operational Amplifier FEATURES D D D D D D D D DESCRIPTION HIGH BANDWIDTH: 300MHz (G = +10) The OPA2846 provides two very low-noise, high gain bandwidth, voltage-feedback op amps in a single package. Operating from a low 12.6mA/channel quiescent current, each channel provides a 1.2nV/√Hz input voltage noise with a 1.65GHz gain bandwidth product. Minimum stable gain is specified at +7V/V while exceptional flatness is ensured at a gain of +10V/V. LOW INPUT VOLTAGE NOISE: 1.2nV/√Hz VERY LOW DISTORTION: –100dBc (5MHz) HIGH SLEW RATE: 600V/µs HIGH DC ACCURACY: VIO = 150µV LOW SUPPLY CURRENT: 12.6mA/ch The combination of low noise, high slew rate (600V/µs) and broad bandwidth allow very high SFDR differential receivers to be implemented. Additionally, decompensated, low-noise, voltage-feedback op amps are ideal for broadband transimpedance requirements. The dual channel OPA2846 provides matched channels for high-speed transimpedance requirements. With over 200MHz bandwidth at a gain of 20dB, excellent gain and phase matching are provided at IF frequencies for matched I and Q channel amplifiers. HIGH GAIN BANDWIDTH PRODUCT: 1650MHz STABLE FOR GAINS ≥ +7V/V APPLICATIONS D HIGH DYNAMIC RANGE ADC PREAMPS D LOW-NOISE, WIDEBAND, TRANSIMPEDANCE AMPLIFIERS OPA2846 RELATED PRODUCTS WIDEBAND, HIGH GAIN AMPLIFIERS LOW-NOISE DIFFERENTIAL RECEIVERS VDSL LINE RECEIVERS ULTRASOUND CHANNEL AMPLIFIERS SECURITY SENSOR FRONT ENDS Power−supply decoup ling not sho wn. 1000pF GAIN BANDWIDTH PRODUCT (MHz) OPA842 OPA843 OPA846 OPA847 2.6 2.0 1.2 0.85 200 800 1750 3900 −80 C1 100Ω 1/2 O P A 28 46 R1 L V+ 2.1pF 14−Bit 10MSPS ADS850 R2 100Ω 500Ω 100Ω 500Ω VI C 50 Ω 2.1pF R2 C1 18pF 1 /2 O P A2 84 6 100Ω V CM 0.1µF Single−to−Differential Gain of 10 1000pF INPUT NOISE VOLTAGE (nV/√Hz) +5V 18p F 1:2 SINGLES R1 Harmonic Distortion (dBc) D D D D D VO = 2VPP Differential −85 2nd−Harmonic −90 −95 −100 −105 3rd−Harmonic L V− −110 1 −5V 10 Frequency (MHz) Differential, 14-Bit, ADC Driver Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright 2003-2004, Texas Instruments Incorporated ! ! www.ti.com "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 ABSOLUTE MAXIMUM RATINGS(1) Power Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±6.5VDC Internal Power Dissipation . . . . . . . . . . . . . . See Thermal Analysis Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±1.2V Input Voltage Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VS Storage Temperature Range: D . . . . . . . . . . . . . . . . −40°C to +125°C Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . . +300°C Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C ESD Rating (Human Body Model) . . . . . . . . . . . . . . . . . . . . 2000V (Charge Device Model) . . . . . . . . . . . . . . . . . . . 1500V (Machine Model) . . . . . . . . . . . . . . . . . . . . . . . . . . 200V (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR(1) OPA2846 SO-8 D SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER TRANSPORT MEDIA, QUANTITY −40°C to +85°C OPA2846 OPA2846ID OPA2846IDR Rails, 100 Tape and Reel, 2500 (1) For the most current specification and package information, refer to our web site at www.ti.com. Top View SO OPA2846 2 Out A 1 8 V+ −In A 2 7 Out B +In A 3 6 −In B V− 4 5 +In B "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 ELECTRICAL CHARACTERISTICS: VS = ±5V Boldface limits are 100% tested at +25°C. RF = 453Ω, RL = 100Ω, and G = +10, unless otherwise noted. See Figure 1 for AC performance. OPA2846ID TYP PARAMETER AC Performance (see Figure 1) Closed-Loop Bandwidth Gain Bandwidth Product Bandwidth for 0.1dB Gain Flatness Peaking at a Gain of +7 Harmonic Distortion 2nd-Harmonic 3rd-Harmonic 2-Tone, 3rd-Order Intercept Input Voltage Noise Input Current Noise Rise-and-Fall Time Slew Rate Settling Time to 0.01% 0.1% 1% Differential Gain Differential Phase Channel-to-Channel Crosstalk DC Performance(4) Open-Loop Voltage Gain (AOL) Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift Input Common-Mode Input Range (CMIR)(5) Common-Mode Rejection Ratio (CMRR) Input Impedance Differential-Mode Common-Mode Output Output Voltage Swing Current Output, Sourcing Current Output, Sinking Closed-Loop Output Impedance MIN/MAX OVER TEMPERATURE TEST CONDITIONS +25°C +25°C(1) 0°C to +70°C(2) −40°C to +85°C(2) G = +7, RG = 50Ω, VO = 200mVPP G = +10, RG = 50Ω, VO = 200mVPP G = +20, RG = 50Ω, VO = 200mVPP G ≥ +40 G = +10, RL = 100Ω, VO = 200mVPP 425 300 100 1650 100 3 250 80 1250 40 225 76 1225 35 200 70 1200 30 UNITS MIN/ MAX TEST LEVEL (3) MHz MHz MHz MHz MHz dB typ min min min min typ C B B B B C dBc dBc dBc dBc dBm nV/√Hz pA/√Hz ns V/µs ns ns ns % deg dBc max max max max min max max max min typ max max typ typ typ B B B B B B B B B C B B C C C G = +10, f = 5MHz, VO = 2VPP RL = 100Ω RL = 500Ω RL = 100Ω RL = 500Ω G = +10, f = 10MHz f > 1MHz f > 1MHz 0.2V Step 2V Step 2V Step 2V Step 2V Step G = +10, NTSC, RL = 150Ω G = +10, NTSC, RL = 150Ω Input Referrred, f = 5MHz −76 −100 −109 −112 44 1.2 2.8 1.3 600 18 12 8 0.02 0.02 −60 −70 −89 −95 −105 41 1.3 3.5 1.6 500 −68 −87 −92 −101 40 1.4 3.6 1.7 400 −66 −85 −90 −96 38 1.5 3.6 1.9 350 14 10 16 12 18 14 VO = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V 90 ±0.15 ±0.5 −10 ±1 ±0.1 ±0.7 82 ±0.65 ±1.6 −20 ±20 ±0.4 ±3.0 81 ±0.73 ±1.6 −20.8 ±20 ±0.5 ±3.0 80 ±0.76 ±1.6 −21.2 ±35 ±0.6 ±3.5 dB mV µV/°C µΑ nA/°C µΑ nA/°C min max max max max max max A A B A B A B VCM = ±1V, Input Referred ±3.2 110 ±3.0 95 ±2.9 93 ±2.8 90 V dB min min A A VCM = 0V VCM = 0V 6.6 2.0 4.7 1.8 kΩ pF MΩ pF typ typ C C ≥ 400Ω Load 100Ω Load VO = 0V VO = 0V G = +10, f = 100kHz ±3.4 ±3.3 80 −80 0.008 V V mA mA Ω min min min min typ A A A A C ±3.3 ±3.2 65 −65 ±3.2 ±3.0 61 −61 ±3.1 ±2.9 60 −60 (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature specifications. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum CMRR specification at ± CMIR limits. 3 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 ELECTRICAL CHARACTERISTICS: VS = ±5V (continued) Boldface limits are 100% tested at +25°C. RF = 453Ω, RL = 100Ω, and G = +10, unless otherwise noted. See Figure 1 for AC performance. OPA2846ID TYP PARAMETER Power Supply Specified Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (−PSRR) TEST CONDITIONS MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to +70°C(2) −40°C to +85°C(2) UNITS MIN/ MAX TEST LEVEL (3) ±6 25.9 24.5 90 ±6 26.3 23.9 88 ±6 26.7 23.3 85 V V mA mA dB typ max max min min C A A A A −40 to +85 °C typ C 125 °C/W typ C +25°C ±5 VS = ±5V, Both Channels VS = ±5V, Both Channels −VS = −4.5 to 5.5 (Input Referred) 25.2 25.2 95 Thermal Characteristics Specified Operating Range: D Package Thermal Resistance, qJA D SO-8 Junction-to-Ambient (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature specifications. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum CMRR specification at ± CMIR limits. 4 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V TA = 25°C, G = +10, RF = 453Ω, RG = 50Ω, and RL = 100Ω, unless otherwise noted. INVERTING SMALL−SIGNAL FREQUENCY RESPONSE NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE 6 3 VO = 0.2VPP RG = 50Ω G = −12 G=7 0 0 Normalized Gain (dB) −3 G = 10 −6 G = 50 −9 G = 20 −12 VO = 0.2VPP RG = RS = 50Ω −3 −6 −12 −15 −15 See Figure 1 −18 1M See Figure 2 −18 10M 100M 1G 1M 10M NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE 29 VO = 0.2VPP 26 20 RL = 100Ω G = +10V/V 23 Gain (dB) Gain (dB) VO = 0.2VPP VO = 1VPP 14 11 20 17 14 11 VO = 2VPP VO = 5VPP 5 See Figure 1 2 10M 8 VO = 5VPP 100M See Figure 2 5 10M 1G Frequency (Hz) Large Signal ±1V Left Scale 2.0 0.4 1.6 0.3 0.8 0.2 0.4 0.1 Small Signal ±100mV Right Scale 0 0 −0.4 −0.1 −0.8 −0.2 −1.2 −0.3 −1.6 See Figure 1 Time (5ns/div) Output Voltage (400mV/div) 1.2 1G INVERTING PULSE RESPONSE 0.5 Output Voltage (100mV/div) G = +10V/V 100M Frequency (Hz) NONINVERTING PULSE RESPONSE 2.0 −2.0 VO = 1VPP RL = 100Ω RG = RS = 50Ω G = −20V/V VO = 2VPP 8 1.6 1G INVERTING LARGE−SIGNAL FREQUENCY RESPONSE 23 17 100M Frequency (Hz) Frequency (Hz) Output Voltage (400mV/div) G = −20 G = −50 −9 1.2 G = −20V/V 0.5 Large Signal ±1V Left Scale 0.8 0.4 0 0.4 0.3 0.2 Small Signal ±100mV Right Scale 0.1 0 −0.4 −0.1 −0.8 −0.2 −1.2 −0.3 −0.4 −1.6 −0.4 −0.5 −2.0 Output Voltage (100mV/div) Normalized Gain (dB) 3 −0.5 Time (5ns/div) 5 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V (continued) TA = 25°C, G = +10, RF = 453Ω, RG = 50Ω, and RL = 100Ω, unless otherwise noted. 5MHz HARMONIC DISTORTION vs LOAD RESISTANCE −60 −70 2nd−Harmonic −75 −80 −85 −90 −95 −100 −105 3rd−Harmonic −110 −115 100 200 250 300 350 400 450 −75 2nd−Harmonic −80 −85 −90 3rd−Harmonic −95 −100 −105 See Figure 1 150 G = +10V/V VO = 5VPP −70 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −65 1MHz HARMONIC DISTORTION vs LOAD RESISTANCE −65 G = +10V/V VO = 2VPP −110 100 500 150 200 Resistance (Ω ) G = +10V/V VO = 2VPP RL = 200Ω 2nd−Harmonic 3rd−Harmonic −85 −95 −105 −75 10 −95 −100 3rd−Harmonic −105 See Figure 1 0.1 100 1 HARMONIC DISTORTION vs NONINVERTING GAIN Harmonic Distortion (dBc) Harmonic Distortion (dBc) VO = 2VPP f = 5MHz RL = 200Ω −90 −95 −100 −105 −75 VO = 2VPP f = 5MHz RL = 200Ω −85 −95 −105 3rd−Harmonic −110 3rd−Harmonic See Figure 1 −115 5 10 15 20 25 30 Gain (V/V) 6 2nd−Harmonic −75 −85 HARMONIC DISTORTION vs INVERTING GAIN −65 2nd−Harmonic −80 35 10 Output Voltage (VPP) −60 −70 500 −90 Frequency (MHz) −65 450 −85 −115 1 400 2nd−Harmonic −80 −110 See Figure 1 0.1 G = +10V/V f = 5MHz RL = 200Ω −70 −75 −115 350 HARMONIC DISTORTION vs OUTPUT VOLTAGE −65 Harmonic Distortion (dBc) Normalized Gain (dB) −65 300 Resistance (Ω ) HARMONIC DISTORTION vs FREQUENCY −55 250 40 45 −115 50 10 15 20 25 30 Gain (−V/V) 35 40 45 50 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V (continued) TA = 25°C, G = +10, RF = 453Ω, RG = 50Ω, and RL = 100Ω, unless otherwise noted. INPUT VOLTAGE AND CURRENT NOISE 2−TONE, 3RD−ORDER INTERMODULATION INTERCEPT 50 10 50 Ω S o u rce G = +10V/V +5V RS 5 0Ω P IN Intercept Point (+dBm) Voltage Noise (nV/√Hz) Current Voise (pA/√Hz) 45 Current Noise 2.8pA/√Hz Voltage Noise 1.2nV/√Hz RL RF 45 3 Ω 35 RG 50Ω 30 100 1k 10k 100k 1M 10M 100M 5 10 15 20 Frequency (Hz) Normalized Gain (1dB) Deviation from 18.06dB Gain (0.1dB) 2 NG = 9.0 0 NG = 9.5 −0.2 −0.5 NG = 10.0 External Compensation See Figure 9 1M 10M 1 50 G = −4 −1 G = −2 −2 G = −1 −3 −4 External Compensation See Figure 5 −5 −6 100M 1G 1M 10M RECOMMENDED RS vs CAPACITIVE LOAD 1 Capacitive Load (pF) 1000 Normalized Gain to Capacitive Load (dB) 10 100 1G FREQUENCY RESPONSE vs CAPACITIVE LOAD G = +10V/V 10 100M Frequency (Hz) 100 RS (Ω) 45 G = −6 0 Frequency (Hz) 1 40 VO = 200mVPP RF = 400Ω NG = 8.0 NG = 8.5 −0.1 −0.4 35 LOW GAIN INVERTING BANDWIDTH 0.1 −0.3 30 3 VO = 200mVPP AV = +8 RF = 453Ω RG = 64.9Ω 0.2 25 Frequency (MHz) NONINVERTING GAIN FLATNESS TUNE 0.3 5 0Ω − 5V 40 20 10 0.4 O P A 2 8 46 25 1 0.5 PO 1 /2 50 Ω 23 RS adjusted for capacitive load. C = 10pF 20 C = 22pF 17 50 Ω So urce V IN 1/ 2 50Ω P ow er− su pp ly +5 V de c ou plin g no t sh ow n. RS VO O P A 2 846 14 CL −5V R 45 3Ω 11 C = 47pF C = 100pF RL 1kΩ (1 kΩ is op tion al.) RG 50 Ω 8 1M 10M 100M 1G Frequency (Hz) 7 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V (continued) TA = 25°C, G = +10, RF = 453Ω, RG = 50Ω, and RL = 100Ω, unless otherwise noted. COMMON−MODE REJECTION RATIO AND POWER−SUPPLY REJECTION RATIO vs FREQUENCY OPEN−LOOP GAIN AND PHASE 120 120 0 100 −30 +PSRR Open−Loop Gain (dB) CMRR and PSRR (dB) 100 90 80 70 −PSRR 60 50 −60 80 ∠AOL 60 −90 −120 40 20log (AOL) 40 30 20 −150 0 −180 −20 −210 102 20 102 103 104 105 106 107 Open−Loop Phase (_) CMRR 110 103 104 108 105 106 Frequency (Hz) 107 108 109 Frequency (Hz) OUTPUT VOLTAGE AND CURRENT LIMITATIONS CLOSED−LOOP OUTPUT IMPEDANCE vs FREQUENCY 4 10 3 RL = 100Ω Output Impedance (Ω ) 2 RL = 50Ω VO (V) 1 RL = 25Ω 0 ZO 1/2 −1 −2 O PA2846 1 453Ω 0.1 50Ω 0.01 −3 0.001 −100 −50 0 50 100 102 150 103 104 I O (mA) 10 0.8 8 0.6 6 0.4 Output 2 0 0.2 0 Output Voltage (2V/div) 4 0.3 0.1 0 −6 −0.6 −8 −0.8 −8 −1.0 −10 100 150 200 250 300 350 400 450 500 Time (50ns/div) 0.4 0 −0.4 50 G = −20V/V RL = 100Ω 0.2 −4 See Figure 1 0.5 Input 2 −0.2 0 108 4 −2 −10 8 G = +10V/V RL = 100Ω 1.0 Input Voltage (200mV/div) Output Voltage (2V/div) 6 107 INVERTING OVERDRIVE RECOVERY NONINVERTING OVERDRIVE RECOVERY Input 106 Frequency (Hz) 10 8 105 −2 Output −4 −6 −0.1 −0.2 −0.3 See Figure 2 0 50 100 150 200 250 300 350 400 450 500 Time (50ns/div) −0.4 −0.5 Input Voltage (100mV/div) −4 −150 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V (continued) TA = 25°C, G = +10, RF = 453Ω, RG = 50Ω, and RL = 100Ω, unless otherwise noted. PHOTODIODE TRANSIMPEDANCE FREQUENCY RESPONSE SETTLING TIME 0.25 0.15 0.10 0.05 0 −0.05 −0.10 −0.15 −0.20 See Figure 1 −0.25 5 10 15 20 80 CD = 100pF 77 CD = 50pF 74 71 CD = 20pF 68 65 See Figure 4 62 25 1 10 Time (ns) SUPPLY AND OUTPUT CURRENT vs TEMPERATURE TYPICAL DC DRIFT OVER TEMPERATURE 15 0.10 10 VIO 0.05 5 0 0 −0.05 −5 Ib −0.10 −10 −0.15 −15 −0.20 −20 −0.25 −50 −25 0 25 50 75 Ambient Temperature (_C) 18 Sourcing Output Current 130 16 120 110 12 100 10 90 8 6 Sinking Output Current 70 −50 −25 0 25 50 75 100 4 125 Ambient Temperature (_ C) COMMON−MODE INPUT RANGE AND OUTPUT SWING vs SUPPLY VOLTAGE COMMON−MODE AND DIFFERENTIAL INPUT IMPEDANCE 107 6 +VIN Common−Mode 4.7MΩ 4 106 +VOUT 2 0 −2 −VOUT Input Impedance (Ω ) Voltage Range (V) 14 Supply Current 80 −25 125 100 20 140 Output Current (10mA/div) 20 100 x IOS Input Bias and Offset Current (µA) Input Offset Voltage (mV) 150 25 0.15 100 Frequency (MHz) 0.25 0.20 CD = 10pF Supply Current (2mA/div) 0 RF = 10kΩ CF Adjusted 20 log(10kΩ) Transimpedance Gain (dBΩ ) Percent of Final Value (%) 83 G = +10V/V RL = 100Ω VO = 2V Step 0.20 105 6.6kΩ 104 Differential 103 −4 −VIN −6 102 2.5 3.0 3.5 4.0 4.5 Supply Voltage (±V) 5.0 5.5 6.0 102 103 104 105 106 107 108 Frequency (Hz) 9 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V (continued) TA = 25°C, G = +10, RF = 453Ω, RG = 50Ω, and RL = 100Ω, unless otherwise noted. CHANNEL−TO−CHANNEL CROSSTALK −30 Input−Referred −35 Crosstalk (5dB/div) −40 −45 −50 −55 −60 −65 −70 −75 −80 1 10 Frequency (MHz) 10 100 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 TYPICAL CHARACTERISTICS: VS = ±5V, DIFFERENTIAL CONFIGURATION TA = 25°C, G = +10, RF = 1kΩ, RG = 50Ω, and RL = 400Ω, unless otherwise noted. DIFFERENTIAL SMALL−SIGNAL FREQUENCY RESPONSE DIFFERENTIAL PERFORMANCE TEST CIRCUIT 3 +5V OPA2846 Gain = VI RG 50Ω RF 1kΩ RG 50Ω RF 1kΩ Normalized Gain (dB) 0 RF VO = = GD RG VI RL 400Ω VO OPA2846 GD = +20V/V RL = 400Ω VO = 400mVPP −3 GD = +10V/V −6 −9 GD = +30V/V −12 GD = +40V/V −15 −5V −18 1 10 100 500 Frequency (MHz) DIFFERENTIAL LARGE−SIGNAL FREQUENCY RESPONSE −60 GD = +20V/V RL = 400Ω Gain (dB) 26 23 VO = 0.4VPP 20 VO = 8VPP 17 Harmonic Distortion (dBc) 29 DIFFERENTIAL DISTORTION vs LOAD RESISTANCE −55 VO = 5VPP −70 −75 −80 −85 3rd−Harmonic −90 −95 −100 −105 −110 −115 14 1 10 100 50 300 100 150 2nd−Harmonic −85 3rd−Harmonic −95 −105 −75 G = +20V/V f = 5MHz RL = 400Ω −80 Harmonic Distortion (dBc) Harmonic Distortion (dBc) GD = +20V/V RL = 400Ω VO = 4VPP −75 250 300 350 400 450 500 DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE DIFFERENTIAL DISTORTION vs FREQUENCY −65 200 Resistance (Ω ) Frequency (MHz) −115 VO = 4VPP G = 20V/V 2nd−Harmonic −65 −85 2nd−Harmonic −90 −95 −100 −105 3rd−Harmonic −110 −115 1 10 Frequency (MHz) 100 1 10 Output Voltage Swing (VPP) 11 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 The OPA2846 provides a unique combination of features—low input voltage noise along with a very low distortion output stage—to give one of the highest dynamic range dual op amps available. Its very high Gain Bandwidth Product (GBP) can be used either to deliver high signal bandwidths at high gains, or to deliver very low distortion signals at moderate frequencies and lower gains. To achieve the full performance of the OPA2846, careful attention to PC board layout and component selection is required, as discussed in the remaining sections of this data sheet. Figure 1 shows the noninverting gain of +10 circuit used as the basis of the Electrical Characteristics and most of the Typical Characteristics. Most of the curves were characterized using signal sources with 50Ω driving impedance, and with measurement equipment presenting a 50Ω load impedance. In Figure 1, the 50Ω shunt resistor at the VI terminal matches the source impedance of the test generator, while the 50Ω series resistor at the VO terminal provides a matching resistor for the measurement equipment load. Generally, data sheet voltage swing specifications are at the output pin (VO in Figure 1), while output power (dBm) specifications are at the matched 50Ω load. The total 100Ω load at the output, combined with the 503Ω total feedback network load, presents the OPA2846 with an effective output load of 83Ω for the circuit of Figure 1. Operating the OPA2846 as an inverting amplifier has several benefits and is particularly appropriate when a matched input impedance is required. Figure 2 shows the inverting gain circuit used as the basis of the inverting mode Typical Characteristics. +5V +VS 0.1µF 0.1µF 50Ω Source 91Ω VO 1 /2 O PA 2846 RG 50Ω 6.8µF 50Ω 50Ω Load RF 1kΩ VI 0.1µF 6.8µF −VS −5V +5V +VS 0.1µF WIDEBAND, INVERTING GAIN OPERATION + WIDEBAND, NONINVERTING OPERATION set to 50Ω and RF adjusted to get the desired gain. Observing this guideline will ensure that the thermal noise contribution of the feedback network is insignificant compared to the 1.2nV/√Hz input voltage noise for the op amp itself. + APPLICATIONS INFORMATION 6.8µF + Figure 2. Inverting, G = −20 Characterization Circuit 50Ω Source 50Ω Load VI VO 1/2 O PA 2 846 50Ω 50Ω RF 453Ω RG 50Ω 6.8µF 0.1µF + −VS −5V Figure 1. Noninverting, G = +10 Specification and Test Circuit Voltage-feedback op amps, unlike current-feedback designs, can use a wide range of resistor values to set their gains. The circuit of Figure 1, and the specifications at other gains, uses the constraint that RG should always be 12 Driving this circuit from a 50Ω source, and constraining the gain resistor (RG) to equal 50Ω, will give both a signal bandwidth and noise advantage. RG acts as both the input termination resistor and the gain setting resistor for the circuit. Although the signal gain (VO/VI) for the circuit of Figure 2 is double that for Figure 1, the noise gains are in fact equal when the 50Ω source resistor is included. This has the interesting effect of doubling the equivalent GBP of the amplifier. This can be seen in comparing the G = +10 and G = −20 small-signal frequency response curves. Both show approximately 250MHz bandwidth, but the inverting configuration of Figure 2 gives 6dB higher signal gain. If the signal source is actually the low impedance output of another amplifier, RG should be increased to the minimum load resistance value allowed for that amplifier and RF should be adjusted to achieve the desired gain. For stable operation of the OPA2846, it is critical that this driving amplifier show a very low output impedance at frequencies beyond the expected closed-loop bandwidth for the OPA2846. "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 LOW-NOISE VDSL RECEIVER SINGLE-STAGE TRANSIMPEDANCE DESIGN Most xDSL transceiver channels are differential for both the driver and the receiver. The low-noise, high-gain bandwidth, and low distortion for the dual OPA2846 make it an ideal receiver channel element for the demanding requirements emerging in VDSL. One possible implementation is shown in Figure 3. This circuit presumes full duplex communication using frequency division multiplexing, with send-and-receive isolation improved through the use of a diplexer line interface. The differential receive signal is brought into the inverting channel gain resistors to get both noise and distortion improvement for a given desired gain setting. To get impedance matching, set 2RG equal to the required load looking out of the diplexer. The signal gain is then set by adjusting feedback resistors, RF. Using the OPA2846 in the inverting mode will give you a reduced noise gain as described in the Wideband, Inverting Gain Operation section of this data sheet. This will improve both the SNR and distortion performance. If the noise gain for a particular application drops below the minimum recommended stable gain (+7), consider using the Low-Gain Compensation technique described later in this data sheet. When setting up either one or both stages as a broadband photodiode amplifier, the key elements in the design are the expected diode capacitance (CD) with the reverse bias voltage (−VB) applied, the desired transimpedance gain RF, and the GBP of the OPA2846 (1650MHz). Figure 4 shows a design using a 10pF source capacitance diode and a 10kΩ transimpedance gain. With these three variables set (and including the parasitic input capacitance for the OPA2846 added to CD), the feedback capacitor value (CF) may be set to control the frequency response. +5V Power−supply decoupling not shown. 1/2 O P A28 46 VO = ID RF RF 10kΩ λ ID CD 10pF −5V CF 0.3pF −VB Figure 4. Wideband, Low-Noise, Transimpedance Amplifier 1/ 2 OPA28 46 RG RF RG RF Passive Filter Analog Front End Diplexer 1/ 2 OPA28 46 Driver Low−Noise VDSL Receiver Figure 3. Low-Noise VDSL Receiver To achieve a maximally-flat, 2nd-order Butterworth frequency response, the feedback pole should be set to: 1ń(2pR FCF) + ǸǒGBPń(4pR FC D)Ǔ (1) Adding the common-mode and differential mode input capacitance (1.8 + 2.0)pF to the 10pF diode source capacitance of Figure 4, and targeting a 10kΩ transimpedance gain using the 1650MHz GBP for the OPA2846, will require a feedback pole set to 31MHz. This will require a total feedback capacitance of 0.5pF. Typical surface-mount resistors have a parasitic capacitance of 0.2pF, leaving the required 0.3pF value shown in Figure 4 to get the required feedback pole. 13 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 This will give a −3dB bandwidth approximately equal to: f *3dB + ǸǒGBPń2pRFC DǓHz (2) The example of Figure 4 will give approximately 44MHz flat bandwidth using the 0.3pF feedback compensation. If the total output noise is bandlimited to a frequency less than the feedback pole frequency, a very simple expression for the equivalent input noise current can be derived as: I EQ + Ǹ ǒ Ǔ EN I N ) 4kT ) RF RF 2 2 (E 2pCDF) ) N 3 (3) IEQ = Equivalent input noise current if the output noise is bandlimited to F < 1/(2πRFCF) IN = Input current noise for the op amp inverting input EN = Input voltage noise for the op amp CD = Diode capacitance F = Bandlimiting frequency in Hz (usually a post filter prior to further signal processing) Evaluating this expression up to the feedback pole frequency at 31MHz for the circuit of Figure 4 gives an equivalent input noise current of 3.1pA/Hz. This is only slightly higher than the current noise of the op amp itself. TWO-STAGE TRANSIMPEDANCE DESIGN The dual OPA2846 may be used as either a dual transimpedance channel from two photodetectors, or as a very high gain stage by using one amplifier as the transimpedance stage with the second used as a post gain amplifier. See Figure 5 for an example of using one channel as a transimpedance front end from a large area detector, with the second amplifier used as a voltage gain stage to get a 100kΩ total gain (ZT) from a large 50pF detector (CD in Figure 5). 1000pF 2.67kΩ 20Ω 1 /2 O P A 28 46 2.67kΩ 732Ω λ CD 50pF 20Ω 1.9pF −VB Figure 5. High-Gain, Wideband Transimpedance Amplifier 14 ȡ Z ȣ +ȧ ȧ Ȣ2pC GBPȤ 2 RF T D (4) Where: 2 Where: 1/2 O P A 2 84 6 One key question in this design is how best to split up the first and second stage gains. If bandwidth optimization from a given photodetector capacitance (CD in Figure 5) is the primary goal, Equation 4 gives a solution for RF in the input stage that will provide an equal bandwidth in the first and second stages, giving the maximum overall channel bandwidth. ZT = Desired total transimpedance gain CD = Diode capacitance at reverse bias GBP = Amplifier Gain Bandwidth Product (MHz) This equation is used to calculate the required input stage feedback resistor in Figure 5. The remaining total signal gain is provided by the second stage; in the example of Figure 5, setting G = 37.5 gives the same bandwidth (approximately 44MHz) as the bandwidth achieved by the input stage. To set this first stage bandwidth to its maximally flat values, use Equation 5 to set the feedback capacitor value: CF + Ǹǒ f *3dB + 1 Ǹ2 CD p RF GBP (GBP) (2pC D) 1ń3 Ǔ (5) 2ń3 (Z T) 1ń3 (6) The approximate achievable bandwidth in the two stages is given by Equation 6, which gives approximately 30MHz for Figure 5. LOW-GAIN COMPENSATION FOR IMPROVED SFDR Where a low gain is desired, and inverting operation is acceptable, a new external compensation technique may be used to retain the full slew rate and noise benefits of the OPA2846 while giving increased loop gain and the associated improvement in distortion offered by the decompensated architecture. This technique shapes the loop gain for good stability while giving an easily-controlled, 2nd-order, low-pass frequency response. Considering only the noise gain (noninverting signal gain, which is also called the Noise Gain or NG) for the circuit of Figure 6, the low-frequency noise gain, (NG1) will be set by the resistor ratios while the high-frequency noise gain (NG2) will be set by the capacitor ratios. The capacitor values set both the transition frequencies and the high-frequency noise gain. If this noise gain (determined by NG2 = 1 + CS/CF) is set to a value greater than the recommended minimum stable gain for the op amp, and the noise gain pole (set by 1/RFCF) is placed correctly, a very well-controlled, 2nd-order, low-pass frequency response will result. "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 due to increased loop gain at frequencies below NG1 × Z0. The capacitor values of Figure 6 are calculated for NG1 = 3 and NG2 = 10 with no adjustment for parasitics. +5V 1/2 O P A 2846 RG 201Ω Figure 7 shows the measured frequency response for the circuit of Figure 6. This shows the expected gain of −2 (6dB) with exceptional flatness through 70MHz and a −3dB bandwidth of 170MHz. Measured distortion into a 100Ω load shows > 5dB improvement through 20MHz over the performance shown in the Typical Characteristics. Into a 500Ω load, the 5MHz, 2VPP, 2nd-harmonic improves from −85dBc to −92dBc. VO RF 402Ω VI CS 29pF CF 3.2pF −5V Figure 6. Broadband Low-Gain Inverting External Compensation 9 6 Z 0 + GBP2 NG 1 ƪǒ 1* Ǔ Ǹ1 * 2 NG ƫ NG NG 1 * NG 2 1 2 (7) Physically, this Z0 (12.4MHz for the values shown above) is set by 1/[2π × RF(CF + CS)] and is the frequency at which the rising portion of the noise gain would intersect unity gain if projected back to 0dB gain. The actual zero in the noise gain occurs at NG1 × Z0, and the pole in the noise gain occurs at NG2 × Z0. Since GBP is expressed in Hz, multiply Z0 by 2π and use this to get CF by solving: CF + 2p 1 R FZ 0NG 2 (+ 3.2pF) (8) Finally, since CS and CF set the high-frequency noise gain, determine CS by: C S + (NG 2 * 1)CF (+ 28.8pF) (9) The resulting closed-loop bandwidth will be approximately equal to: f *3dB ^ ǸZ 0 GBP (+ 143MHz) (10) For the values of Figure 6, the f−3dB will be approximately 130MHz. This is less than that predicted by simply dividing the GBP by NG1. The compensation network controls the bandwidth to a lower value while providing the full slew rate at the output and an exceptional distortion performance 3 Gain (3dB/div) To choose the values for both CS and CF, two parameters and only three equations need to be solved. The first parameter is the target high-frequency noise gain NG2, which should be greater than the minimum stable gain for the OPA2846. Here, a target NG2 of 10 will be used. The second parameter is the desired low-frequency signal gain, which also sets the low-frequency noise gain NG1. To simplify this discussion, we will target a maximally-flat, 2nd-order, low-pass Butterworth frequency response (Q = 0.707). The signal gain of −2 shown in Figure 6 will set the low-frequency noise gain to NG1 = 1 + RF/RG (NG1 = 3 in this example). Then, using only these two gains and the GBP for the OPA2846 (1650MHz), the key frequency in the compensation can be determined as: 0 −3 −6 −9 −12 −15 1M 10M 100M 1G Frequency (Hz) Figure 7. Low Gain Inverting Frequency Response DC-COUPLED, SINGLE-TO-DIFFERENTIAL ADC DRIVER Many very high performance CMOS ADCs are intended to operate with a differential input signal. Translating a single-ended source to this differential input while controlling the common-mode operating voltage can present a considerable challenge where high SFDR is required. See Figure 8 for one way to do this, where very low harmonic distortion is required, and good commonmode control and DC precision is desired. This particular example is set for a signal gain of 16 from the single-ended input to the differential output voltage. Since the common-mode control signal (from the output of the OPA820) is fed into the midpoint of the two gain resistors (93.8Ω), this DC control path requires a very low source impedance through high frequencies to maintain the desired signal path gain. A wideband, unity-gain stable, voltage-feedback op amp like the OPA820 makes an ideal choice to provide this low output impedance DC control signal. This op amp also compares the output common-mode voltage to the desired VCM, and servos the OPA2846 common-mode output voltage to that value, using an integrator loop. This holds the output commonmode voltage precisely at VCM while giving the low output impedance required of the circuit. 15 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 +5V Power−supply decoupling not shown. 12Ω VI 750Ω 95Ω 1/2 OPA2846 50ΩInput Impedance 200Ω +5V 1.3µH V+ 750Ω 3.8kΩ 30pF +5V 93.8Ω 14−Bit 10MSPS ADS850 100Ω 0.1µF OPA820 2.01kΩ 2.5V VCM +5V −5V 93.8Ω 1µF 0.1µF 750Ω 93.8Ω 3.8kΩ 750Ω 1/2 OPA2846 49.9Ω −5V 200Ω 1.3µH V− Voltage at V+ to V− = 2VPP 2nd−order Butterworth post filter f−3dB = 18MHz. Figure 8. DC-Coupled, Single-to-Differential High SFDR ADC Driver One side of the OPA2846 is operating at a gain of +9 with some attenuation of the input signal to have an equivalent +8 gain. The other side of the OPA2846 is operating at a gain of −8. To deliver a 2VPP differential input signal on a 2.5V common-mode voltage, each output must swing between 2.0V and 3.0V. Tested harmonic distortion performance for this condition from 1MHz to 10MHz is shown in Figure 9. In this case, the 2nd-harmonic distortion is still dominant due to slight signal path imbalances. The distortion levels, however, are very low. Thus, narrowband applications which are impacted by only 3rd-order terms will see very low single- and two-tone distortion levels. 16 −70 Harmonic Distortion (dBc) Operating at +2.5V output common-mode requires a DC level shifting current through the feedback resistors. Since this current is to the supply midpoint, pull-up resistors equal to the feedback resistors are connected to the positive supply to keep the output stage signal currents equal and bipolar. This significantly improves 2nd-harmonic distortion. −75 −80 2nd−Harmonic −85 3rd−Harmonic −90 −95 1 10 Frequency (MHz) Figure 9. Harmonic Distortion vs Frequency for the Circuit of Figure 8 For more information on the 2nd-order post filter, refer to RLC Filter Design for ADC Interface Applications (SBAA108), available for download at www.ti.com. "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 AC-COUPLED, SINGLE-TO-DIFFERENTIAL ADC DRIVER Where the signal path may be AC-coupled, a very balanced, high SFDR dual op amp interface circuit can easily be provided by the OPA2846. Figure 10 shows a specific example of this application, where the input single-to-differential conversion is provided by an input transformer. Once the signal source is purely differential, the circuit of Figure 10 provides low harmonic distortion with a common-mode control path that does not interact with the signal path gain. If the source is already differential, such as at the output of a balanced mixer, the input transformer could be replaced by blocking capacitors. In the example of Figure 10, the secondary of the transformer is connected into the two inverting path gain resistors (100Ω). These resistors provide both an input impedance match (assuming a 50Ω source on the primary of this 1:2 step-up transformer) and set the signal gain for each amplifier along with the 500Ω feedback resistors. Although relatively high signal gain is provided by this Power−supply decoupling not shown. VCM 1000pF circuit (10 in this case), each amplifier is operating at a relatively low noise gain (3.5V/V). This low-noise gain at low frequencies gives high loop gain for distortion suppression in the baseband. External compensation capacitors (18pF and 2.1pF) are included to hold the frequency response flat, as described in the Low-Gain Compensation For Improved SFDR section of this data sheet. The common-mode operating voltage is fed into each amplifier’s noninverting input. Since these are equal, and will appear at each inverting input as well, no DC current is produced through the transformer secondary due to this common-mode operating voltage. Since no current flows due to VCM, the output will operate at VCM as well. This is one of the few common-mode operating point control techniques that requires no current to flow. This makes the common-mode control aspect of this circuit essentially non-interactive with the signal path. To provide a 2VPP differential signal operating at a 2.5V output common-mode requires a 2.0V to 3.0V output swing on each output. Tested performance over frequency for the circuit of Figure 10 is shown in Figure 11. +5V R1 1/2 OPA2846 L V+ C 18pF 2.1pF 100Ω 500Ω R2 14−Bit 10MSPS ADS850 500Ω 1:2 VI 2.5V 50Ω VCM VCM +10VI 100Ω 500Ω 1µF 2.1pF 500Ω Single−to−Differential Gain of 10 18pF VCM 1000pF L 1/2 OPA2846 V− R1 C R2 −5V Figure 10. AC-Coupled, Single-to-Differential High SFDR ADC Driver 17 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 Harmonic Distortion (dBc) −80 VO = 2VPP Differential −85 2nd−Harmonic, 0VDC −90 PACKAGE BOARD PART NUMBER LITERATURE REQUEST NUMBER OPA2846ID SO-8 Surface-Mount DEM-OPA268xU SBOU003 Table 1. Evaluation Module Ordering Information 3rd−Harmonic, +2.5VDC −95 MACROMODELS AND APPLICATIONS SUPPORT 2nd−Harmonic, +2.5VDC −100 −105 3rd−Harmonic, 0VDC −110 1 10 Frequency (MHz) Figure 11. Harmonic Distortion for Figure 10 Figure 11 shows 2nd- and 3rd-harmonic distortion for a 2VPP differential output swing at both 0V output common-mode voltage and +2.5V common-mode voltage. Since there is no DC current required from the output to level shift to +2.5V in this circuit, no pull-up resistors to the power supply were used as in the circuit of Figure 8. The 2nd harmonic remains the dominant distortion mechanism, but shows little sensitivity to the commonmode operating voltage (improved 2nd-harmonic distortion results were achieved with this circuit using two individual OPA846’s with an extremely symmetrical layout). The 3rd harmonic is essentially unmeasureable for the ground-centered output swing, but increases as the output is shifted to a +2.5V DC output. Narrowband systems, where a bandpass filter less than an octave wide can be inserted between the amplifier and the converter, will only be concerned about 2-tone, 3rd-order intermodulation distortion. Since this bandpass filter is also AC-coupled, the outputs of Figure 10 may be operated ground-centered, giving the extremely low 3rd-order distortions of Figure 11. DESIGN-IN TOOLS DEMONSTRATION BOARDS A PC board is available to assist in the initial evaluation of circuit performance using the OPA2846. It is available free, as an unpopulated PC board delivered with descriptive documentation. The summary information for this board is shown in Table 1. Contact the Texas Instruments applications support line to request this board or visit Texas Instruments’ web site at www.ti.com. 18 PRODUCT Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This is particularly true for video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. A SPICE model for the OPA2846 is available through the Texas Instruments web page (http://www.ti.com). These models do a good job of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do not do as well in predicting the harmonic distortion characteristics. OPERATING SUGGESTIONS SETTING RESISTOR VALUES TO MINIMIZE NOISE The OPA2846 provides a very low input noise voltage while requiring a low 12.6mA/channel quiescent current. To take full advantage of this low input noise, careful attention to the other possible noise contributors is required. Figure 12 shows the op amp noise analysis model with all the noise terms included. In this model, all the noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. ENI 1/2 OPA2846 RS EO IBN ERS RF √4kTRS 4kT RG RG IBI √ 4kTRF 4kT = 1.6E − 20J at 290_K Figure 12. Op Amp Noise Analysis Model "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 The total output spot noise voltage can be computed as the square root of the squared contributing terms to the output noise voltage. This computation adds all the contributing noise powers at the output by superposition, then takes the square root to get back to a spot noise voltage. Equation 11 shows the general form for this output noise voltage using the terms shown in Figure 12. EO + Ǹǒ (11) Ǔ E2NI ) ǒIBNRSǓ ) 4kTR S NG 2 ) ǒIBIRFǓ ) 4kTRFNG 2 2 Dividing this expression by the noise gain (NG = 1 + RF/RG) will give the equivalent input-referred spot noise voltage at the noninverting input as shown in Equation 12. EN + Ǹ E2NI ) ǒIBNRSǓ 2 ǒ Ǔ I R ) 4kTR S ) BI F NG 2 4kTR F ) NG (12) Inserting high resistor values into Equation 12 can quickly dominate the total equivalent input referred noise. A 105Ω source impedance on the noninverting input will add a thermal voltage noise term equal to that of the amplifier itself. As a simplifying constraint, set RG = RS in Equation 12 and assume an RS/2 source impedance at the noninverting input (where RS is the signal’s source impedance with another matching RS to ground on the noninverting input). This results in Equation 13, where NG > 10 has been assumed to further simplify the expression. EN + Ǹ (ENI) ) 5 ǒI BR SǓ ) 4kT 4 2 2 ǒ3R2 Ǔ S (13) Evaluating this expression for RS = 50Ω will give a total equivalent input noise of 1.64nV/√Hz. Note that the NG has dropped out of this expression. This is valid only for NG > 10. FREQUENCY RESPONSE CONTROL Voltage-feedback op amps exhibit decreasing closed-loop bandwidth as the signal gain is increased. In theory, this relationship is described by the GBP shown in the specifications. Ideally, dividing GBP by the noninverting signal gain will predict the closed-loop bandwidth. In practice, this only holds true when the phase margin approaches 90°, as it does in high gain configurations. At low gains (with an increased feedback factor), most high-speed amplifiers will exhibit a more complex response with lower phase margin. The OPA2846 is compensated to give a maximally-flat, 2nd-order, Butterworth, closed-loop response at a noninverting gain of +10 (see Figure 1). This results in a typical gain of +10 bandwidth of 300MHz, far exceeding that predicted by dividing the 1650MHz GBP by 10. Increasing the gain will cause the phase margin to approach 90° and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain of +40, the OPA2846 will show the 41MHz bandwidth predicted using the simple formula and the typical GBP of 1650MHz. Inverting operation offers some interesting opportunities to increase the available GBP. When the source impedance is matched by the gain resistor (see Figure 2), the signal gain is (1 + RF/RG) while the noise gain for bandwidth purposes is (1 + RF/2RG). This cuts the noise gain almost in half, increasing the minimum stable gain for inverting operation under these condition to −12 and the equivalent GBP to 3.2GHz. DRIVING CAPACITIVE LOADS One of the most demanding and yet very common load conditions for an op amp is capacitive loading. Often, the capacitive load is the input of an A/D converter, including additional external capacitance which may be recommended to improve ADC linearity. A high-speed, high open-loop gain amplifier like the OPA2846 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the amplifier’s open-loop output resistance is considered, this capacitive load introduces an additional pole in the signal path that can decrease the phase margin. Several external solutions to this problem have been suggested. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the phase margin and improving stability. The Typical Characteristics show the recommended RS vs Capacitive Load and the resulting frequency response at the load. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA2846. Long PC board traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always consider this effect carefully, and add the recommended series resistor as close as possible to the OPA2846 output pin (see the Board Layout section). The criterion for setting this RS resistor is a maximum bandwidth, flat frequency response at the load. For the OPA2846 operating in a gain of +10, the frequency response at the output pin is very flat to begin with, allowing relatively small values of RS to be used for low capacitive loads. As the signal gain is increased, the unloaded phase margin will also increase. Driving capacitive loads at higher gains will require lower RS values than those shown for a gain of +10. 19 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 DISTORTION PERFORMANCE The OPA2846 is capable of delivering an exceptionally low distortion signal at high frequencies over a wide range of gains. The distortion plots in the Typical Characteristics show the typical distortion under a wide variety of conditions. Most of these plots are limited to 110dB dynamic range. Generally, until the fundamental signal reaches very high frequencies or powers, the 2nd-harmonic will dominate the distortion with a negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network; in the noninverting configuration, this is sum of (RF + RG), while in the inverting configuration, it is just RF (see Figure 1 and Figure 2). Increasing output voltage swing increases harmonic distortion directly. A 6dB increase in output swing will generally increase the 2nd-harmonic to 12dB and the 3rd-harmonic to 18dB. Increasing the signal gain will also increase the 2nd-harmonic distortion. Again, a 6dB increase in gain will increase the 2nd and 3rd harmonic by approximately 6dB each, even with constant output power and frequency. Finally, the distortion increases as the fundamental frequency increases, due to the rolloff in the loop gain with frequency. Conversely, the distortion will improve going to lower frequencies down to the dominant open-loop pole at approximately 100kHz. Starting from the −82dBc 2nd-harmonic for a 5MHz, 2VPP fundamental into a 200Ω load at G = +10 (from the Typical Characteristics), the 2nd-harmonic distortion for frequencies lower than 100kHz will approximately be: −82dBc − 20 log(5MHz/100kHz) = −116dBc The OPA2846 has extremely low 3rd-order harmonic distortion. This also gives a high 2-tone, 3rd-order intermodulation intercept as shown in the Typical Characteristics. This intercept curve is defined at the 50Ω load when driven through a 50Ω matching resistor to allow direct comparisons to RF MMIC devices. This matching network attenuates the voltage swing from the output pin to the load by 6dB. If the OPA2846 drives directly into the input of a high impedance device, such as an A/D converter, the 6dB attenuation is not taken. Under these conditions, the intercept will increase by a minimum 6dBm. The intercept is used to predict the intermodulation spurious for two, closely-spaced frequencies. If the two test frequencies, f1 and f2, are specified in terms of average and delta frequency, fO = (f1 + f2)/2 and f = |f2 − f1|/2, the two 3rd-order, close-in spurious tones will appear at fO ± 3 × ∆f. The difference between two equal test-tone power levels and these intermodulation spurious power levels is given by dBc = 2 × (IM3 − PO) where IM3 is the intercept taken from the Typical Characteristic and PO is the power level, in dBm, at the 50Ω load for one of the two closely-spaced test frequencies. For instance, at 10MHz, the OPA2846 at a gain of +10 has an intercept of 44dBm 20 at a matched 50Ω load. If the full envelope of the two frequencies needs to be 2VPP, this requires each tone to be 4dBm. The 3rd-order intermodulation spurious tones will then be 2 × (48 − 4) = 88dBc below the test-tone power level (−84dBm). If this same 2VPP, 2-tone envelope were delivered directly into the input of an ADC—without the matching loss or the loading of the 50Ω network—the intercept would increase to at least 50dBm. With the same signal and gain conditions, but now driving directly into a light load, the spurious tones will then be at least 2 × (54 − 4) = 100dBc below the 4dBm test-tone power levels centered on 10MHz. DC ACCURACY AND OFFSET CONTROL The OPA2846 can provide excellent DC signal accuracy due to its high open-loop gain, high common-mode rejection, high power-supply rejection, and low input offset voltage and bias current offset errors. To take full advantage of its low ±0.65mV input offset voltage, careful attention to input bias current cancellation is also required. The low noise input stage of the OPA2846 has a relatively high input bias current (10µA typical into the pins), but with a very close match between the two input currents—typically ±100nA input offset current. The total output offset voltage may be reduced considerably by matching the source impedances looking out of the two inputs. For example, one way to add bias current cancellation to the circuit of Figure 1 (page 12) would be to insert a 20Ω series resistor into the noninverting input from the 50Ω terminating resistor. When the 50Ω source resistor is DC-coupled, this will increase the source resistances for the noninverting input bias current to 45Ω. Since this is now equal to the resistance looking out of the inverting input (RF || RG), the circuit will cancel the gains for the bias currents to the output, leaving only the offset current times the feedback resistor as a residual DC error term at the output. Using the 453Ω feedback resistor, this output error will now be less than ±0.6µA × 453Ω = ±0.27mV over the full temperature range. A fine-scale output offset null, or DC operating point adjustment, is often required. Numerous techniques are available for introducing a DC offset control into an op amp circuit. Most of these techniques eventually reduce to setting up a DC current through the feedback resistor. One key consideration to selecting a technique is to insure that it has a minimal impact on the desired signal path frequency response. If the signal path is intended to be noninverting, the offset control is best applied as an inverting summing signal to avoid interaction with the signal source. If the signal path is intended to be inverting, applying the offset control to the noninverting input can be considered. For a DC-coupled inverting input signal, this DC offset signal will set up a DC current back into the source that must be considered. An offset adjustment placed on the inverting op amp input can also change the noise gain and frequency response flatness. "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 Figure 13 shows one example of an offset adjustment for a DC-coupled signal path that will have minimum impact on the signal frequency response. In this case, the input is brought into an inverting gain resistor with the DC adjustment an additional current summed into the inverting node. The resistor values setting this offset adjustment are much larger than the signal path resistors. This will insure that this adjustment has minimal impact on the loop gain and hence, the frequency response as well. Power−supply decoupling not shown. +5V 48Ω RG 50Ω 5kΩ 1/2 OPA2846 −5V VO RF 1kΩ VI 20kΩ ±200mV Output Adjustment 10kΩ 0.1µF 5kΩ PD = 10V × (26.6mA) + 2 × [52/(4 × (100Ω || 500Ω))] = 416mW Maximum TJ = +85°C + (0.416Ω × 125°C/Ω) = 137°C This absolute worst-case example will never be encountered in practice. Therefore, 137°C sets an upper limit to maximum operating junction temperature. BOARD LAYOUT +5V 0.1µF As a worst-case example, compute the maximum TJ using both channels of the OPA2846ID in the circuit of Figure 1 (page 12) operating at the maximum specified ambient temperature of +85°C and driving a grounded 100Ω load at +2.5VDC: VO R = − F = −20 VI RG −5V Figure 13. DC-Coupled, Inverting Gain of −20, with Output Offset Adjustment THERMAL ANALYSIS The OPA2846 will not require heatsinking or airflow in most applications. Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed +150°C. Operating junction temperature (TJ) is given by TA + PD × qJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum when the output is fixed at a voltage equal to 1/2 either supply voltage (for equal bipolar supplies). Under this worst-case condition, PDL = VS2/(4 × RL) where RL includes feedback network loading. Note that it is the power in the output stage and not in the load that determines internal power dissipation. Achieving optimum performance with a high-frequency amplifier like the OPA2846 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability; on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25”) from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequencies, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. c) Careful selection and placement of external components will preserve the high-frequency performance of the OPA2846. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal-film and carbon composition, axially-leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board trace length as short as possible. Never use wirewound type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other network components, such as noninverting input termination resistors, should also be placed close to the package. Where double-side component mounting is allowed, place the feedback 21 "#$% www.ti.com SBOS274A −JUNE 2003 − REVISED MARCH 2004 resistor directly under the package on the other side of the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial metal-film or surface-mount resistors have approximately 0.2pF in shunt with the resistor. For resistor values > 1.5kΩ, this parasitic capacitance can add a pole and/or a zero below 500MHz that can affect circuit operation. Keep resistor values as low as possible, consistent with load driving considerations. It has been suggested here that a good starting point for design would be to set RG to 50Ω. Doing this will automatically keep the resistor noise terms low, and minimize the effect of their parasitic capacitance. d) Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of recommended RS vs Capacitive Load. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA2846 is nominally compensated to operate with a 2pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on board, and in fact, a higher impedance environment will improve distortion, as shown in the distortion versus load plots. With a characteristic board trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the output of the OPA2846 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. If the 6dB attenuation of a doubly-terminated 22 transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the plot, RS vs Capacitive Load. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. e) Socketing a high-speed part like the OPA2846 is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network, which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA2846 onto the board. INPUT AND ESD PROTECTION The OPA2846 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 13. +VCC External Pin Internal Circuitry −VCC Figure 14. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (for example, in systems with ±15V supply parts driving into the OPA2846), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. 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