TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 D D D D D D D D NE PACKAGE (TOP VIEW) High Output Drive Capability . . . 1 A Min 3-State Outputs High Gain-Bandwidth Product 8 MHz Typ Low Total Harmonic Distortion < 0.08% Typ High Slew Rate . . . 12 V/µs Typ Class AB Output Stage Thermal Shutdown Mains-Line Driver Circuit Application Included COMP2 VCC + OUT1 VCC – VCC – OUT2 VCC + TRS2 1 16 2 15 3 14 4 13 5 12 6 11 7 10 8 9 COMP1 VCC – 1N + VCC – VCC – IN – VCC – TRS1 Terminals 4, 5, 12 and 13 are connected to the lead frame. description MAXIMUM PEAK-TO-PEAK OUTPUT VOLTAGE vs FREQUENCY VO(PP) – Maximum Peak-to-Peak Output Voltage – V The TLE2301 is a power operational amplifier that can deliver an output current of 1 A at high frequencies with very low total harmonic distortion. The device has an integral 3-state mode to drive the output stage into a high-impedance state and also to reduce the supply current to less than 3.5 mA. The combination of high output current and 3-state outputs makes the TLE2301 ideal for implementing the signalling transformer driver in mains-based telemetering modems. This combination of features also makes the device well suited for other high-current applications (e.g., motor drivers and audio circuits). Using the Texas Instruments established Excalibur process, the TLE2301 is able to achieve slew rates in excess of 12 V/µs and a gainbandwidth product of 8 MHz. The TLE2301 uses a 16-pin NE power package to provide better power handling capabilities than standard dual-inline packages. 8 VCC ± = ± 5 V TA = 25°C 7 6 RL = 4.3 Ω RL = 8.1 Ω RL = 20 Ω 5 4 3 2 1 0 100 1k 10 k 100 k f – Frequency – Hz 1M 10 M Figure 1 The TLE2301 is characterized for operation over the industrial temperature range of – 40°C to 85°C. AVAILABLE OPTION PACKAGE TA VIOmax AT 25°C THERMALLY-ENHANCED PLASTIC DIP (NE) – 40°C to 85°C 10 mV TLE2301INE Copyright 1993, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 1 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 equivalent schematic (entire device) COMP1 COMP2 VCC + OUT1 + _ TRS1 OUT2 TRS2 VCC – IN + IN – equivalent schematic (TRS1 and TRS2 inputs) VCC + TRS1 TRS2 VCC – 2 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 Terminal Functions TERMINAL DESCRIPTION NAME NO. COMP1 COMP2 16 1 COMP1 and COMP2 are compensation network terminals IN + 14 Noninverting input IN – 11 Inverting input OUT1 OUT2 3 6 Two low-distortion class-AB output stages. g Each is capable of sourcing g more than 500 mA. OUT1 and OUT2 should be connected together for all applications. TRS1 TRS2 9 8 TRS1 and TRS2 are 3-state input terminals. TRS2 should be connected to the ground of the circuit generating the 3-state command ((normally µP g ground). yµ ) The TLE2301 is brought g into 3-state mode by y raising g TRS1 2 V above TRS2. Placing g the TLE2301 in a 3-state mode reduces the supply current to below 2.2 mA (typ). Normal operation resumes by bringing TRS1 to within 0.8 V of TRS2. The 3-state function can be disabled by connecting both TRS1 and TRS2 to VCC – . VCC – 10, 15 High-impedance VCC – input terminals. Although these do not carry any of the device’s supply current, they increase the stability of the device and should be connected to the negative supply terminal (VCC –). VCC – 4, 5, 12, 13 Negative supply terminals and substrate. As with all NE packages, the substrate is directly connected to the lead frame. The result is that the junction-to-ambient thermal impedance (ZθJA) is greatly reduced by soldering the negative supply terminals to the copper area of the printed-circuit board (PCB). VCC + 2, 7 Positive supply terminals. Both terminals should be connected to the positive voltage supply. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 3 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 absolute maximum ratings over operating free-air temperature range (unless otherwise noted)† Supply voltage, VCC + (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 V Supply voltage, VCC – (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 22 V Differential input voltage, VID (see Note 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 44 V Duration of short-circuit current at (or below) 25°C (see Note 3) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . unlimited Continuous total dissipation at (or below) 25°C free-air temperature (see Notes 4 and 5) . . . . . . . 2075 mW Continuous total dissipation at 85°C case temperature (see Note 5) . . . . . . . . . . . . . . . . . . . . . . . . . 4640 mW Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 85°C Operating case or virtual junction temperature range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 150°C Storage temperature range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 65°C to 150°C † Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. NOTES: 1. All voltage values, except differential voltages, are with respect to the midpoint between VCC + and VCC – . 2. Differential voltages are at IN+ with respect to IN –. 3. The outputs when connected together may be shorted to either supply. Temperature and/or supply voltages must be limited to ensure that the maximum dissipation rating is not exceeded. 4. For operation above 25°C free-air temperature, derate linearly at the rate of 16.56 mW/°C. 5. For operation above 25°C case temperature, derate linearly at the rate of 71.4 mW/°C. To avoid exceeding the design maximum virtual junction temperature, these ratings should not be exceeded. Due to variations in individual device electrical characteristics and thermal resistance, the built-in thermal overload protection may be activated at power levels slightly above or below the rated dissipation. FREE-AIR TEMPERATURE DISSIPATION DERATING CURVE CASE TEMPERATURE DISSIPATION DERATING CURVE 10 Derating Factor = 16.56 mW/°C ZθJC = 60.4°C/ W PD – Total Continuous Power Dissipation – W PD – Total Continuous Power Dissipation – W 2.5 2 1.5 1 0.5 0 6 4 2 Derating Factor = 71.4 mW/°C ZθJC = 14°C/ W 0 25 4 8 40 55 70 TA – Free-Air Temperature – °C 85 POST OFFICE BOX 655303 0 • DALLAS, TEXAS 75265 25 50 75 TC – Case Temperature – °C 100 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 recommended operating conditions Supply voltage, VCC ± VCC± = ± 5 V VCC± = ± 15 V Common mode input voltage, voltage VIC Common-mode High-level 3-state enable voltage, VIH MIN MAX UNIT ± 4.5 ± 20 V –4 1.6 V –14 11.8 V 2 Low-level 3-state enable voltage, VIL Continuous output current Operating free-air temperature, TA – 40 V 0.8 V 1 A 85 °C electrical characteristics at specified free-air temperature, VCC ± = ±5 V, CC = 15 pF (unless otherwise noted) (see Figure 5) PARAMETER TEST CONDITIONS VIO Input offset voltage VO = 0,, RS = 50 Ω VIC = 0,, IIB Input bias current VO = 0,, RS = 50 Ω VIC = 0,, VICR Common-mode input voltage g range g RS = 50 Ω VOM + Maximum positive peak output voltage swing RL = 20 Ω Ω, See Note 6 VOM – Maximum negative peak output voltage swing RL = 20 Ω Ω, See Note 6 AVD Large signal differential voltage amplification Large-signal VO = ± 2 V,, RL = 20 Ω VIC = 0,, ri Differential input resistance ro Output resistance (see Note 7) CMRR TA† 25°C MIN TYP MAX 0.4 7 Full range 10 25°C 283 Full range 500 Full range g –4 to 1.6 25°C 3.3 Full range 3.2 25°C – 3.2 Full range – 3.1 25°C 65 Full range 60 25°C TRS1 = 0.8 V 450 mV nA V 3.5 V – 3.4 V 87 dB 1 25°C UNIT MΩ 1 Ω 100 kΩ TRS1 = 2 V, 3-state mode Common mode rejection ratio Common-mode VIC = VICRmin,, RS = 50 Ω VO = 0,, 25°C 65 88 dB kSVR Supply voltage rejection ratio (∆VCC ± /∆VIO) Supply-voltage VCC ± = ± 4.5 V to ± 20 V,, VIC = 0, No load 25°C 70 100 dB IIH Enable input current current, high VI = 2 V V, IIL current low Enable input current, 8V VI = 0 0.8 IOS Short-circuit output current (see Note 8) VO = 0, ICC 3 state mode 3-state 25°C 0.01 Full range 0.5 25°C 0.01 Full range tp ≤ 50 µs VO = 0 0, No load VO = 0, 3-state mode No load, Supply current 25°C 25°C 0.5 0.5 1 1.8 µA µA A 10 21 1.73 2.7 Full range 25°C 0.5 25 mA Full range 3.5 † Full range is – 40°C to 85°C. NOTES: 6. OUT1 and OUT2 are connected together for all tests. 7. TRS1 voltage is measured with respect to TRS2 potential. 8. Pulse testing techniques are used to maintain the junction temperature as close to the ambient temperature as possible. Thermal effects must be taken into account separately (tp = pulse duration time) . POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 5 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 electrical characteristics at specified free-air temperature, VCC ± = ±15 V, CC = 15 pF (unless otherwise noted) (see Figure 5) PARAMETER TEST CONDITIONS VIO Input offset voltage VO = 0,, RS = 50 Ω VIC = 0,, IIB Input bias current VO = 0,, RS = 50 Ω VIC = 0,, VICR Common-mode input voltage g range g RS = 50 Ω VOM + Maximum positive peak output voltage swing RL = 20 Ω Ω, See Note 6 VOM – Maximum negative peak output voltage swing RL = 20 Ω Ω, See Note 6 AVD Large signal differential voltage amplification Large-signal VO = ± 6 V,, RL = 20 Ω VIC = 0,, ri Differential input resistance ro Output resistance (see Note 7) CMRR MIN TYP MAX 0.3 10 Full range 15 25°C 260 Full range Full range g TRS1 = 0.8 V 3-state mode Common-mode rejection ratio VIC = VICRmin, RS = 50 Ω VO = 0, kSVR Supply voltage rejection ratio (∆VCC ± /∆VIO) Supply-voltage VCC ± = ± 4.5 V to ± 20 V,, VIC = 0, No load IIH Enable input current current, high VI = 2 V V, IIL Enable input current, current low VI = 0 0.8 8V IOS Short-circuit output current (see Note 8) VO = 0, tp ≤ 50 µs VO = 0 0, No load VO = 0, 3-state mode No load, 3 state mode 3-state – 14 to 11.8 25°C 13 Full range 13 25°C – 12.6 Full range – 12.5 25°C 70 Full range 65 13.5 mV nA V – 13 V 102 dB 1 25°C UNIT V MΩ 1 Ω 100 kΩ 25°C 70 97 dB 25°C 70 100 dB 25°C 0.01 Full range 25°C 25°C 25°C 0.01 0.5 0.5 1 3 11 Full range 25°C 0.5 0.5 Full range Supply current 450 500 25°C TRS1 = 2 V, ICC TA† 25°C µA A 25 30 2.2 µA 3.5 mA Full range 5 † Full range is – 40°C to 85°C. NOTES: 6. OUT1 and OUT2 are connected together for all tests. 7. TRS1 voltage is measured with respect to TRS2 potential. 8. Pulse testing techniques are used to maintain the junction temperature as close to the ambient temperature as possible. Thermal effects must be taken into account separately (tp = pulse duration time) . 6 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 operating characteristics at specified free-air temperature, VCC± = ±5 V, CC = 15 pF, TA = 25°C (unless otherwise noted) (see Figure 5) PARAMETER TEST CONDITIONS RL = 20 Ω, MIN TYP MAX 9 12 V/µs UNIT SR Slew rate at unity gain (see Figure 1) VO = ± 1.5 V, CL = 100 pF ts Settling time (see Figure 1) RL = 20 Ω,, CL = 100 pF,, 3-V step to 30 mV (1%) 07 0.7 µs Vn Equivalent input noise voltage (see Figure 2) RS = 50 Ω, f = 1 kHz 44 nV/√Hz THD Total harmonic distortion VO = 1 Vrms, RL = 20 Ω, f = 50 kHz, CL = 100 pF 0.04% B1 φm Unity-gain bandwidth (see Figure 3) RL = 20 Ω, CL = 100 pF 8 Phase margin at unity gain (see Figure 3) RL = 20 Ω, CL = 100 pF 30° MHz operating characteristics at specified free-air temperature, VCC± = ±15 V, CC = 15 pF, TA = 25°C (unless otherwise noted) (see Figure 5) PARAMETER TEST CONDITIONS TYP MAX 9 14 V/µs UNIT SR Slew rate at unity gain (see Figure 1) VO = ± 10 V, CL = 100 pF ts Settling time (see Figure 1) RL = 20 Ω,, CL = 100 pF,, 20-V step to 200 mV (1%) 18 1.8 µs Vn Equivalent input noise voltage (see Figure 2) RS = 50 Ω, f = 1 kHz 44 nV/√Hz THD Total harmonic distortion VO = 2 Vrms, RL = 20 Ω, f = 50 kHz, CL = 100 pF B1 φm Unity-gain bandwidth (see Figure 3) RL = 20 Ω, CL = 100 pF 8 Phase margin at unity gain (see Figure 3) RL = 20 Ω, CL = 100 pF 35° POST OFFICE BOX 655303 RL = 20 Ω, MIN • DALLAS, TEXAS 75265 0.08% MHz 7 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 PARAMETER MEASUREMENT INFORMATION 5 kΩ VCC + _ VI VO + VCC – CL (see Note A) VCC + _ VO + RL VCC – 50 Ω 50 Ω NOTE A: CL includes the fixture capacitance. Figure 2. Slew-Rate Test Circuit Figure 3. Noise-Voltage Test Circuit R2 10 kΩ VI VCC + _ + VCC – CL (see Note A) VI – R1 VCC + _ VO VO VI + + R3 RL COMP1 COMP1 Cc 15 pF NOTE A: CL includes the fixture capacitance. Figure 4. Gain-Bandwidth and Phase-Margin Test Circuit VCC – Figure 5. Compensation Configuration typical values Typical values presented in this data sheet represent the median (50% point) of the device parametric performance. 8 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 TYPICAL CHARACTERISTICS Table of Graphs FIGURE IIB IIO Input bias current vs Free-air temperature 6, 7 Input offset current vs Free-air temperature 6, 7 vs Frequency AVD Differential voltage amplification VO(PP) Maximum peak-to-peak output voltage VOM Maximum peak output voltage ZθJA Transient junction-to-ambient thermal impedance ICC Supply current vs Free-air temperature INPUT BIAS CURRENT AND INPUT OFFSET CURRENT vs FREE-AIR TEMPERATURE 1000 VCC ± = ± 15 V VIC = 0 100 IIB 10 1 – 50 IIO – 25 0 25 50 75 TA – Free-Air Temperature – °C 100 I IO – Input Bias and Input Offset Currents – nA IIIB IB and IIO I IO – Input Bias and Input Offset Currents – nA IIIB IB and IIO Output impedance 9 vs Frequency 10, 11 vs Output current 12, 13 vs Supply voltage 14 vs Time 15 vs Supply voltage 16 vs Free-air temperature Pulse response zo 8 17 Small signal 18, 19 Large signal 20, 21 vs Frequency 22, 23 INPUT BIAS CURRENT AND INPUT OFFSET CURRENT vs FREE-AIR TEMPERATURE 1000 VCC ± = ± 5 V VIC = 0 IIB 100 10 1 0.1 – 50 IIO – 25 0 25 50 75 TA – Free-Air Temperature – °C Figure 6 100 Figure 7 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 9 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 TYPICAL CHARACTERISTICS DIFFERENTIAL VOLTAGE AMPLIFICATION vs FREQUENCY ÁÁ ÁÁ 20° VCC ± = ± 15 V RL = 20 Ω CC = 100 pF TA = 25°C 100 110 40° 80 60° 60 80° 40 100° 20 120° 0 140° – 20 10 100 1k 10 k 100 k 1M AVD – Differential Voltage Amplification – dB AVD – Differential Voltage Amplification – dB 120 DIFFERENTIAL VOLTAGE AMPLIFICATION vs FREE-AIR TEMPERATURE ÁÁ ÁÁ 160° 10 M RL = 20 Ω 100 90 VCC± = ± 5 V 80 70 60 – 50 f – Frequency – Hz – 25 0 25 50 75 TA – Free-Air Temperature – °C Figure 8 MAXIMUM PEAK-TO-PEAK OUTPUT VOLTAGE vs FREQUENCY 25 RL = 8.1 Ω 20 15 10 5 0 100 1k 10 k 100 k f – Frequency – Hz 1M 10 M VO(PP) – Maximum Peak-to-Peak Output Voltage – V VO(PP) – Maximum Peak-to-Peak Output Voltage – V VCC ± = ± 15 V TA = 25°C RL = 20 Ω 8 VCC ± = ± 5 V TA = 25°C 7 6 RL = 4.3 Ω RL = 8.1 Ω RL = 20 Ω 5 4 3 2 1 0 100 Figure 10 10 100 Figure 9 MAXIMUM PEAK-TO-PEAK OUTPUT VOLTAGE vs FREQUENCY 30 VCC ± = ± 15 V 1k 10 k 100 k f – Frequency – Hz Figure 11 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 1M 10 M TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 TYPICAL CHARACTERISTICS MAXIMUM NEGATIVE PEAK OUTPUT VOLTAGE vs OUTPUT CURRENT VOM – – Maximum Negative Peak Output Voltage – V VOM + – Maximum Positive Peak Output Voltage – V MAXIMUM POSITIVE PEAK OUTPUT VOLTAGE vs OUTPUT CURRENT 15 TA = 25°C 12.5 VCC ± = ± 15 V 7.5 VCC ± = ± 5 V 2.5 0 0 200 800 400 600 IO – Output Current – mA TA = 25°C – 12.5 10 5 – 15 1000 VCC ± = ± 15 V – 10 – 7.5 –5 ÁÁÁ ÁÁÁ ÁÁÁ VCC ± = ± 5 V – 2.5 0 0 200 TRANSIENT JUNCTION-TO-AMBIENT THERMAL IMPEDANCE† vs ON TIME MAXIMUM PEAK OUTPUT VOLTAGE vs SUPPLY VOLTAGE Z θ JA – Transient Junction-to-Ambient Thermal Impedance – °C/mW VOM – Maximum Peak Output Voltage – V 100 RL = 20 Ω TA = 25°C 15 VOM + 10 5 0 –5 – 10 VOM – – 15 – 20 0 2 4 1000 Figure 13 Figure 12 20 400 600 800 IO – Output Current – mA 6 8 10 12 14 16 VCC ± – Supply Voltage – V 18 20 d = 50% 10 d = 20% d = 10% d = 5% d = 2% 1 Single Pulse 0.1 0.001 0.01 Figure 14 0.1 1 10 t – On Time – s 100 1000 Figure 15 † d = duty cycle POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 11 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 TYPICAL CHARACTERISTICS SUPPLY CURRENT vs SUPPLY VOLTAGE SUPPLY CURRENT vs FREE-AIR TEMPERATURE 10.8 10.7 VO = 0 No Load 10.6 10.6 I CC – Supply Current – mA I CC – Supply Current – mA 10.8 VO = 0 No Load TA = 25°C 10.5 10.4 10.3 10.2 10.1 VCC ± = ± 15 V 10.4 10.2 10 VCC ± = ± 5 V 9.8 9.6 10 9.9 0 2 4 6 8 10 12 14 16 18 9.4 – 50 20 – 25 VCC ± – Supply Voltage – V Figure 16 150 10 100 VO – Output Voltage – mV VO – Output Voltage – V VOLTAGE FOLLOWER SMALL-SIGNAL PULSE RESPONSE 15 5 0 –5 VCC ± = ± 15 V RL = 20 Ω CL = 100 pF TA = 25°C 0 2 4 6 8 t – Time – µs 50 0 – 50 VCC ± = ± 5 V RL = 20 Ω CL = 100 pF TA = 25°C – 100 – 15 –2 10 12 14 – 150 – 0.5 0 0.5 1 t – Time – µs Figure 18 12 100 Figure 17 VOLTAGE FOLLOWER SMALL-SIGNAL PULSE RESPONSE – 10 0 25 50 75 TA – Free-Air Temperature – °C Figure 19 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 1.5 2 2.5 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 TYPICAL CHARACTERISTICS VOLTAGE FOLLOWER LARGE-SIGNAL PULSE RESPONSE VOLTAGE FOLLOWER LARGE-SIGNAL PULSE RESPONSE 3 150 VCC ± = ± 15 V RL = 20 Ω CL = 100 pF TA = 25°C 2 VO – Output Voltage – V VO – Output Voltage – mV 100 50 0 – 50 – 100 1 0 –1 VCC ± = ± 5 V RL = 20 Ω CL = 100 pF TA = 25°C –2 – 150 – 0.5 0 0.5 1 1.5 t – Time – µs 2 –3 –2 2.5 0 2 4 6 8 t – Time – µs Figure 20 4 VCC ± = ± 15 V TA = 25°C 3.5 AVD = 100 3 2.5 2 AVD = 10 1.5 1 VCC ± = ± 5 V TA = 25°C 3 AVD = 100 2.5 2 1.5 AVD = 10 1 AVD = 1 AVD = 1 0.5 0.5 0 1k 14 OUTPUT IMPEDANCE vs FREQUENCY z o – Output Impedance – Ω z o – Output Impedance – Ω 3.5 12 Figure 21 OUTPUT IMPEDANCE vs FREQUENCY 4 10 10 k 0 1k f – Frequency – Hz 100 k 1M f – Frequency – Hz Figure 22 Figure 23 100 k 1M 10 M POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 10 k 10 M 13 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION circuit for mains-line driver over 40-kHz-to-90-kHz utility band The following application is a circuit for a mains-line driver over 40-kHz-to-90-kHz utility band and is based around the European standard (EN56065 –1) describing utility and consumer applications. This example shows a possible implementation for differential transmission on the mains line. This applications circuit is designed around the requirements of a domestic electricity meter operating over a utility band of 40 kHz to 90 kHz. A dual-rail power supply of ± 5 V is used for this design example to limit device power dissipation. The same design principles, however, can be applied to other applications. frequency band The frequency band for utility applications extends over an enormous range from 3 kHz to 95 kHz. In order to have a coupling network that is economical and implemented with readily available components, this circuit is designed for a subband from 40 kHz to 90 kHz. This subband is sufficiently wide to support multichannel operation; i.e., 10 channels of 5 kHz width or more if the channel widths are smaller. To avoid transmission spillover into the next band, a guard band of 5 kHz is allowed. The upper frequency of this circuit is set to 90 kHz, and the lower frequency is chosen for an economical coupling network and still has sufficient bandwidth to support multichannel operation. output drive The impedance of the mains network at these signalling frequencies is relatively low (<1 Ω to 30 Ω). This circuit has been designed to drive a 4-Ω mains line over the 40-kHz-to-90-kHz bandwidth. The signalling impedance of the mains network fluctuates as different loads are switched on during the day or over a season, and it is influenced by many factors such as: D D D Localized loading from appliances connected to the mains supply near to the connection of the communication equipment; e.g., heavy loads such as cookers and immersion heaters and reactive loads such as EMC filters and power factor correctors Distributed loading from consumers connected to the same mains cable, where their collective loading reduces the mains signalling impedance during times of peak electricity consumption; e.g., meal times Network parameters; e.g., transmission properties of cables and the impedance characteristics of distribution transformers and other system elements With such a diversity of factors, the signalling environment fluctuates enormously, irregularly, and can differ greatly from one installation to another. The signalling system should be designed for reliable communications over a wide range of mains impedances and signalling conditions. Consequently, the transmitter must be able to drive sufficient signal into the mains network under these loading conditions. The TLE2301 amplifier has 1-A output drive capability with short-circuit protection; hence, it adequately copes with the high current demands required for implementing mains signalling systems. 3-state facility When transmitting, the transmitter appears as a low-impedance signal source on the mains network. If transmitters are left in the active mode whether transmitting or not and a large number of transmitters are installed in close proximity, their combined loading would reduce the mains impedance to unacceptable levels. Not only would each transmitter need to drive into an extremely low mains impedance, but signals arriving from distant transmitters would be severely attenuated. To overcome this problem, the transmitters need to present a high impedance to the mains network when they are not transmitting. The mains network is then only loaded by a few transmitters at any one time, and the mains signalling impedance is not adversely affected. 14 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION 3-state facility (continued) The TLE2301 incorporates an output 3-state facility, removing the need for additional circuitry to achieve this function. In addition, the TLE2301 has a low standby current in the 3-state mode, making it ideal for applications where low power consumption is also essential. circuit configuration The design methodology is to minimize power dissipation in the TLE2301 by maximizing the use of the available output voltage swing of the amplifier. The amplifier’s output can swing to within 2 V of the supply rail before saturation begins. With a chosen supply of ± 5 V, the maximum peak-to-peak voltage swing is 6 V. To ensure that the amplifier’s output is not likely to clip under heavy loads, the maximum output voltage swing has been reduced by 0.5 V, giving a usable peak-to-peak output voltage swing of 5.5 V. It is assumed that the input signal to the transmitter stage has a peak-to-peak amplitude of 2.8 V (1 Vrms) as might be expected if the transmission signal is digitally synthesized by circuitry operating solely from the 5-V supply. The gain of the amplifier stage is appropriately set to: Gain output voltage swing + peak-to-peak peak-to-peak input voltage V + 5.5 2.8 V + 1.96 An inverting amplifier configuration is chosen for this example, as the input signal source is assumed to have a relatively low impedance in relation to the gain-setting resistors. CF1 15 pF CI 100 nF RF 4.7 kΩ 11 VI RS 3.3 Ω 16 – 1 RI 2.4 kΩ TRS1 (3-state control) CF2 39 pF 3 IC1 14 9 7 6 2 + 4 D1 1N4001 5 D2 1N4001 CC 470 nF Mains Supply 5V –5 V CD1 220 µF + + CD2 220 µF CD3 100 nF CD4 100 nF L1 P2820 0V Figure 24. Full-Circuit Diagram for Utility Band A noninverting amplifier configuration could be used when the input signal needs to be terminated with high impedance, but the user should take care that the amplitude of the input signal does not exceed the common-mode input range (– 4 V < VICM < 1.8 V at VCC = ± 5 V) for low-gain implementations. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 15 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION component calculations The following sections contain the calculations for input capacitors, gain resistors, coupling network, coupling capacitors, transformer-leakage inductance, series resistors, decoupling, and frequency compensation. input capacitor The incoming signal is ac coupled to remove any incoming dc offset and to provide only unity gain for the amplifier’s input offset voltage. The value of 100 nF is chosen for this input capacitor as it has very little influence on the amplifier’s signal gain over the frequency band. gain resistors The gain-setting resistors are chosen for a gain of 1.96; i.e., choosing: Gain + RRF I R F + 4.7 kΩ and RI + 2.4 kΩ kΩ + 4.7 2.4 kΩ + 1.96 The resistor values are low enough to ensure that the circuit does not suffer from stray capacitance and signal pick-up problems but not too low as to significantly load the mains impedance when the amplifier is in its high-impedance state. coupling network The function of the line interface is to provide isolation from the mains supply while coupling the communication signals onto the mains network. As the mains voltage is large in comparison with the communication signals, the mains voltage needs to be isolated from the electronic circuitry. The simple coupling network limits the current flowing from the mains supply as well as providing a convenient point at which to implement the safety isolation barrier between the mains supply and the communications circuitry. The transformer can easily achieve an isolation of 4 kV between primary and secondary windings, and the capacitor provides the low frequency roll-off to impede the mains voltage. The transformer has two other useful properties. First, the turns ratio can be selected to provide efficient power transfer between the TLE2301 amplifier and the mains network. Second, the transformer possesses leakage inductance that can be tuned with the coupling capacitor to form a band-pass filter. By altering the turns ratio, the power dissipated in the TLE2301 can be reduced while maintaining the required voltage levels on the mains line. A turns ratio of 1.67:1 was selected in this design to apply a 120-µdBV signal onto the mains line. The calculation for the turns ratio is not straightforward due to the presence of numerous complex impedances. The simplest method for deriving the turns ratio is to model the circuit with an analog simulation program such as PSpice. It is from these simulations that the 1.67:1 turns ratio has been selected. PSpice is a registered trademark of MicroSim Corporation. 16 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION coupling capacitor With such a wide frequency band, the quality factor of the coupling filter needs to be low in order to avoid unacceptably large attenuation at the band edges and to achieve a good coupling performance that is insensitive to a wide range of loads. For a band-pass filter of this configuration, the quality factor is proportional to the reciprocal of the coupling capacitance. For low Q, the value of CC needs to be large. Q + quality factor and CC + coupling capacitor Q ∝ 1 C C Counterbalancing this need for a large value of CC creates two more considerations. First, the capacitance should not be so large as to allow significant 50-Hz mains current through the transformer ( I = 2 × π × f × CC × V). Second, the coupling capacitor is required to meet certain safety standards. The coupling capacitor is actually an RFI-suppression capacitor that has been designed by the manufacturers to provide an adequate level of protection when connected across the various conductors of the mains supply (consult the UL1283 or UL1414 standards for RFI capacitors). These types of capacitors can be expensive, physically large, restricted in capacitance value, and limited in the number of manufacturers. As a reasonable compromise between all these factors, a coupling capacitor of 470 nF is chosen. This value is multisourced, moderately priced, limits the mains current through the transformer to less than 36 mA rms, and has sufficient capacitance to form the desired low-Q filter. transformer leakage inductance The transformer leakage inductance, inherent to the transformer, can be used to form an LC band-pass filter. If the capacitor alone is used to couple onto the mains network, its capacitance value needs to be quite large for it to have a reasonably low reactance at the signalling frequencies. Forming an LC filter greatly reduces the value of capacitor required. The center frequency of the filter is not the same as the midband frequency of 65 kHz. Band-pass filters show a symmetrical shape only when plotted against the logarithm of frequency, so the center frequency (fo) is given by the following formula: fo Ǹ + flower fupper + Ǹ(40 90 ) kHz + 60 kHz The leakage inductance of the transformer, as viewed from the winding connected to the coupling capacitor, is derived from 2 π fO = 1 / √ LC. The required leakage inductance of the transformer is: L + (2πf + (2π o 1 )2 C C 1 60 kHz ) 2 470 nF + 15 µH Transformer Leakage Inductance Capacitor Figure 25. Band-Pass Coupling Filter POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 17 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION series resistor The series resistor, RS, is included to limit the turn-on current, the amplifier’s offset current, and the signalling current through the filter. With dual supply rails, there is always a potential problem of large turn-on currents as the amplifier powers up. If one supply rail turns on before the other, the output of the TLE2301 amplifier could saturate near to the applied supply rail, causing a large current to flow through the transformer winding (Rwinding = 0.1 Ω for the P2820 transformer). The power supply needs to be of sufficient rating to ensure that its rails could rise to the minimum operating voltage of the amplifier, at which point the amplifier is ensured to have returned to stable operation. With a series resistor of 3.3 Ω and assuming the output saturates at the maximum peak-to-peak voltage excursion of 3 V, this turn-on current is limited to less than the device’s 1-A rating ( Itransient = 3 V / 3.3 Ω = 0.91 A). Further reduction of this turn-on current by raising the value of the series resistor deteriorates the filter’s performance into low signalling impedances on the mains network. Alternatively, this turn-on current could be blocked by means of a series capacitor, but for this frequency band the capacitor has to be large in value ( ≥ 3.3 µF ) so as not to adversely affect the filter. A nonpolarized capacitor of this value is relatively expensive, and the resistor is still required to fulfill other functions. Another way of preventing overcurrent at power up is to use the TLE2301 3-state mode. As the TRS2 control line is intended to be tied to the microprocessor’s 0-V rail, the TRS1 control line must be taken high to activate the 3-state mode, which implies that the positive rail is required to turn on first. Other schemes could be devised to take TRS2 below the 0-V rail until the power supply has stabilized if the negative rail turns on first. Instead of relying on a definite power-supply sequence or elaborate control circuitry, it is simpler to limit the current either with a series resistor or capacitor. The second function of the series resistor is to limit the dc current flow through the transformer winding due to the dc offset at the amplifier’s output, which is caused by its input offset voltage. For a worst case input offset of 20 mV, the output offset is also 20 mV as the dc gain of the circuit is unity. Offsets due to input bias currents are negligible since the values of the gain-setting resistors are low. The dc current through the transformer is therefore less than 7 mA (20 mV/3.3 Ω). This low level of dc current does not appreciatively increase the power dissipation of the amplifier or noticeably diminish the harmonic performance of the transformer. The final function of the series resistor is to limit the signalling current in the event that the mains impedance might appear as solely reactive; i.e., without a resistive component. As a rough estimate, the peak signal current from the amplifier is: I OM where: V 18 O (PP) I OM + V O (PP) R S ǒ Ǔ+ + 5.5 V 2 3.3 Ω 833 mA + Peak-to-peak output voltage swing + Peak-output-signalling current from amplifier POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION series resistor (continued) Again, the value of the series resistor is sufficient to limit the peak-signal current below the device’s maximum rating. This calculation does not take into account other resistive impedances in the signal path, which would further reduce the peak signal current from the amplifier. decoupling Power-supply decoupling for the amplifier is provided by a 220-µF electrolytic capacitor and a 100-nF ceramic capacitor per supply rail located close to the supply terminals of the TLE2301 device. The decoupling capacitors for the negative supply should be connected to a pair of VCC – terminals (4 and 5 or 12 and 13), whichever pair is most convenient from a printed-circuit-board (PCB) layout point of view. In order to minimize parasitic lead inductances, these capacitors should be located as close as possible to the device terminals to which they are connected. As the VCC+ terminals are not adjacent on the package, the decoupling capacitors should be connected to one terminal with a thick PCB track going to the other terminal. The 220-µF electrolytic capacitor is chosen to provide good decoupling performance (less than 25-mV ripple under the worst-case loading for the utility circuit). This value could be reduced to 100 µF for higher-frequency consumer bands. The level of ripple depends on the source impedance of the power supply and the equivalent series resistance of the chosen decoupling capacitors. The 100-nF ceramic capacitor provides high-frequency decoupling for the amplifier. CF1 15 pF CF2 39 pF 16 11 – 1 3 IC1 14 7 2 + 6 15 10 13 12 VCC + 5 4 220 µF + 100 nF VCC – 220 µF + 100 nF 0V Figure 26. Amplifier Decoupling and Compensation POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 19 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION frequency compensation The TLE2301 amplifier requires one compensation capacitor. However, when driving heavy loads, stability can be increased by connecting VCC – terminals 10 and 15 to VCC – terminals 12 and 13 and using another capacitor between COMP2 and the outputs. The circuit included in this application has been designed with two compensation capacitors. The component values chosen are: + 15 pF C + 33 pF F2 C F1 These component values could be adjusted if the amplifier is used for higher-frequency applications. power dissipation The impedance of the mains network fluctuates greatly for many reasons, but its impedance at the supplydistribution transformer is typically very low, less than 1 Ω, whereas the mains impedance in a house commonly has a higher value, from 4 Ω to 40 Ω. For utility-metering applications, a master transmitter may be sited at the supply-distribution transformer and would need to deliver more power into the mains network than the household transmitter when generating comparable signal amplitudes. NE thermally-enhanced dual in-line package 14 mm d 5 mm d TLE2301 Z θJA – Junction-to-Ambient Thermal Impedance – °C/W The TLE2301 utilizes the four center terminals of the dual-in-line package (NE) to transfer heat to a copper area on the PCB. A copper area of 1290 mm2 provides a junction-to-ambient thermal impedance, ZθJA, of 34°C / W, allowing the device to dissipate up to 1.9 W at 85°C for a junction temperature of 150°C or up to 1.5 W at 85°C for a junction temperature of 135°C. JUNCTION-TO-AMBIENT THERMAL IMPEDANCE vs DIMENSIONS 50 45 40 35 30 25 20 0 10 20 30 d – Dimensions – mm NOTE: When d = 25 mm, ZθJA = 34°C/ W Figure 27. PCB Heatsink 20 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 40 50 TLE2301 EXCALIBUR 3-STATE-OUTPUT WIDE-BANDWIDTH POWER OPERATIONAL AMPLIFIER SLOS131 – DECEMBER 1993 APPLICATION INFORMATION power dissipation in amplifier Ǔ)ǒ Ǔ For sinusoidal waveforms, the dissipation in the amplifier, PAMP , is: P where: + AMP ǒ 2 2 V CC I V CC CC π I OM – P O + AmplifierȀs quiescent current I + Peak-output-signalling current from amplifier OM P + Output power consumed by coupling network and load O I CC The power dissipated in the amplifier is minimized if the amplifier’s peak output current, IOM, is minimized. Since the output power consumed by the coupling and load is a function of current and voltage ( PO ≈ IO × VO ), the amplifier’s peak output current can be minimized by maximizing the amplifier’s output voltage swing. circuit parts list The associated parts list is: REFERENCE FIGURE COMPONENT DESCRIPTION IC1 Figure 24, Figure 26 TLE2301 operational amplifier Texas Instruments TLE230INE L1 Figure 24 1.67:1, 15-µH leakage transformer Electronics Techniques P2820 (European manufacturer) CC Figure 24 470-nF capacitor Metalized paper, safety standards UL1414 CI Figure 24 100-nF capacitor Ceramic, general purpose CF1 Figure 24, Figure 26 15-pF capacitor Ceramic, general purpose CF2 Figure 24, Figure 26 39-pF capacitor Ceramic, general purpose CD1, CD2 Figure 24 220-µF, 10-V min capacitors Aluminum electrolytic, general purpose CD3, CD4 Figure 24 100-nF capacitors Ceramic, general purpose RF Figure 24 4.7-kΩ, 0.125-W min resistor Metal film, general purpose RI Figure 24 2.4-kΩ, 0.125-W min resistor Metal film, general purpose RS Figure 24 3.3-kΩ, 1-W min, resistor D1, D2 Figure 24 1N4001 series, 1-A min diodes POST OFFICE BOX 655303 General purpose • DALLAS, TEXAS 75265 21 IMPORTANT NOTICE Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue any product or service without notice, and advise customers to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those pertaining to warranty, patent infringement, and limitation of liability. TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. Testing and other quality control techniques are utilized to the extent TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed, except those mandated by government requirements. CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE (“CRITICAL APPLICATIONS”). TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS. INCLUSION OF TI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO BE FULLY AT THE CUSTOMER’S RISK. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other intellectual property right of TI covering or relating to any combination, machine, or process in which such semiconductor products or services might be or are used. TI’s publication of information regarding any third party’s products or services does not constitute TI’s approval, warranty or endorsement thereof. Copyright 1998, Texas Instruments Incorporated