TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 650-kHz/1.2-MHz 18.5-V STEP-UP DC-DC CONVERTER Check for Samples: TPS61085T FEATURES APPLICATIONS • • • • • • • • • 1 • • • • • 2.3-V to 6-V Input Voltage Range 18.5-V Boost Converter With 2.0-A Switch Current 650-kHz/1.2-MHz Selectable Switching Frequency Adjustable Soft-Start Thermal Shutdown Undervoltage Lockout 8-Pin MSOP Package Handheld Devices GPS Receiver Digital Still Camera Portable Applications DSL Modem PCMCIA Card TFT LCD Bias Supply DESCRIPTION The TPS61085 is a high-frequency high-efficiency dc-to-dc converter with an integrated 2.0-A 0.13-Ω power switch capable of providing an output voltage up to 18.5 V. The selectable frequency of 650 kHz and 1.2 MHz allows the use of small external inductors and capacitors and provides fast transient response. The external compensation allows optimizing the application for specific conditions. A capacitor connected to the soft-start pin minimizes inrush current at startup. L 3.3 mH VIN 2.3 V to 6 V 6 CIN Cby 1 mF 16 V 5 IN 3 10 mF 16 V D PMEG2010AEH VS 12 V/300 mA SW EN 2 R1 156 kW Cout 1 R2 18 kW 2* 10 mF 25 V FB 7 COMP FREQ 4 Rcomp 51 kW 8 GND SS TPS61085 Css 100 nF Ccomp 1.6 nF 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2009, Texas Instruments Incorporated TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION (1) PACKAGE (2) TA –40°C to 105°C (1) (2) ORDERABLE PART NUMBER TOP-SIDE MARKING MSOP-8 – DGK Reel of 2500 TPS61085TDGKR PTQI TSSOP-8 – PW Reel of 2000 TPS61085TPWR 61085T For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. Package drawings, thermal data, and symbolization are available at www.ti.com/packaging. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) VALUE UNIT Input voltage range IN (2) –0.3 to 7 V Voltage range on pins EN, FB, SS, FREQ, COMP –0.3 to 7 V Voltage on pin SW 20 V ESD rating HBM 2 kV ESD rating MM 200 V ESD rating CDM 500 V Continuous power dissipation See Dissipation Rating Table Operating junction temperature range –40 to 150 °C Storage temperature range –65 to 150 °C 260 °C Lead temperature (soldering, 10 sec) (1) (2) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability All voltage values are with respect to network ground terminal. DISSIPATION RATINGS (1) (1) (2) (2) PACKAGE RθJA TA ≤ 25°C POWER RATING TA = 70°C POWER RATING TA = 105°C POWER RATING MSOP 181°C/W 552 mW 303 mW 110 mW TSSOP 160°C/W 625 mW 343 mW 125 mW PD = (TJ – TA) / RθJA RθJA given for High-K PCB board RECOMMENDED OPERATING CONDITIONS MIN VIN Input voltage range VS Boost output voltage range TA TJ 2 TYP MAX UNIT 2.3 6 V VIN + 0.5 18.5 V Operating free-air temperature –40 105 °C Operating junction temperature –40 125 °C Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 ELECTRICAL CHARACTERISTICS VIN = 3.3 V, EN = IN, VS = 12 V, TA = –40°C to 105°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 6 V 70 100 μA SUPPLY VIN Input voltage range IQ Operating quiescent current into IN Device not switching, VFB = 1.3 V ISDVIN Shutdown current into IN EN = GND 1 μA UVLO Undervoltage lockout threshold VIN falling 2.2 V VIN rising 2.3 V TSD Thermal shutdown TSD(HYS) Thermal shutdown hysteresis 2.3 Temperature rising 150 °C 14 °C LOGIC SIGNALS EN, FREQ VIH High level input voltage VIN = 2.3 V to 6 V VIL Low level input voltage VIN = 2.3 V to 6 V 2 0.5 V V Ilkg Input leakage current EN = FREQ = GND 0.1 μA 18.5 V 1.246 V BOOST CONVERTER VS Boost output voltage VIN + 0.5 VFB Feedback regulation voltage 1.230 gm Transconductance error amplifier IFB Feedback input bias current VFB = 1.238 V RDS(on) N-channel MOSFET on-resistance VIN = VGS = 5 V, ISW = current limit VIN = VGS = 3.3V, ISW = current limit 1.238 μA/V 107 0.1 μA 0.13 0.20 Ω 0.15 0.24 2.0 2.6 3.2 A 7 10 13 μA MHz Ilkg SW leakage current ILIM N-Channel MOSFET current limit EN = GND, VSW = 6V ISS Soft-start current VSS = 1.238 V fosc Oscillator frequency FREQ = high 0.9 1.2 1.5 FREQ = low 480 650 820 Line regulation VIN = 2.3 V to 6 V, IOUT = 10 mA Load regulation VIN = 3.3 V, IOUT = 1 mA to 400 mA 10 Product Folder Link(s): TPS61085T kHz 0.0002 %/V 0.11 %/A Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated µA 3 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com PIN ASSIGNMENT DGK, PW PACKAGES (TOP VIEW) COMP 1 8 SS FB 2 7 FREQ EN 3 6 IN PGND 4 5 SW 8-PIN 4.9mm x 3mm x 1.1mm MSOP (DGK) 8-PIN 6.4mm x 3mm x 1.2mm TSSOP (PW) TERMINAL FUNCTIONS TERMINAL NAME NO. I/O DESCRIPTION COMP 1 I/O FB 2 I Feedback pin EN 3 I Shutdown control input. Connect this pin to logic high level to enable the device PGND 4 Power ground SW 5 Switch pin IN 6 Input supply pin FREQ 7 SS 8 4 I Compensation pin Frequency select pin. The power switch operates at 650 kHz if FREQ is connected to GND and at 1.2 MHz if FREQ is connected to IN Soft-start control pin. Connect a capacitor to this pin if soft-start needed. Open = no soft-start Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE η Efficiency vs Load current, VS = 12 V, VIN = 3.3V Figure 1 η Efficiency vs Load current, VS = 9 V, VIN = 3.3 V Figure 2 PWM switching - discontinuous conduction Figure 3 PWM switching - continuous conduction Figure 4 Load transient response at High frequency Figure 5 Load transient response at Low frequency Figure 6 Soft-start Figure 7 Supply current vs Supply voltage Figure 8 Frequency vs Load current Figure 9 Frequency vs Supply voltage Figure 10 EFFICIENCY vs LOAD CURRENT 100 EFFICIENCY vs LOAD CURRENT 100 f = 650 kHz L = 6.8 mH 90 80 80 f = 1.2 Mhz L = 3.3 mH f = 1.2 Mhz L = 3.3 mH 70 Efficiency = % 70 Efficiency = % f = 650 kHz L = 6.8 mH 90 60 50 40 60 50 40 30 30 20 20 VIN = 3.3 V VS = 12 V 10 0 VIN = 3.3 V VS = 9 V 10 0 0 0.05 0.10 0.15 0.20 0.25 0.30 0.35 0.40 0.45 0.50 0 0.10 IO - Load current - A 0.20 0.30 0.40 0.50 0.60 0.70 0.80 IO - Load current - A Figure 1. Figure 2. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 5 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com PWM SWITCHING DISCONTINUOUS CONDUCTION MODE PWM SWITCHING CONTINUOUS CONDUCTION MODE VSW 5 V/div VSW 5 V/div VS_AC 50 mV/div VS_AC 50 mV/div VIN = 3.3 V VS = 12 V/1 mA IL 1 A/div VIN = 3.3 V VS = 12 V/300 mA IL 200 mA/div 200 ns/div 200 ns/div Figure 3. Figure 4. LOAD TRANSIENT RESPONSE HIGH FREQUENCY (1.2 MHz) LOAD TRANSIENT RESPONSE LOW FREQUENCY (650 kHz) Cout = 20 mF VIN = 3.3 V VS = 12 V VS_AC 200 mV/div Cout = 20 mF VIN = 3.3 V VS = 12 V L = 3.3 mH Rcomp = 51 kW Ccomp = 1.6 nF L = 6.8 mH Rcomp = 24 kW Ccomp = 3.3 nF VS_AC 200 mV/div IOUT = 50 mA - 200 mA IOUT = 50 mA - 200 mA IOUT 100 mA/div IOUT 100 mA/div 200µs/div 200 ms/div 200 ms/div Figure 5. 6 Figure 6. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 SUPPLY CURRENT vs SUPPLY VOLTAGE SOFT-START 2 1.8 EN 5 V/div Switching f = 1.2 Mhz L = 3.3 mH ICC - Supply Current - mA 1.6 VIN = 3.3 V VS = 12 V/300 mA VS 5 V/div CSS = 100 nF IL 1 A/div 1.4 1.2 1 0.8 Switching f = 650 kHz L = 6.8 mH 0.6 0.4 Not Switching 0.2 2 ms/div 0 2 2.5 3 3.5 4 4.5 5 VCC - Supply Current - V Figure 7. Figure 8. FREQUENCY vs LOAD CURRENT FREQUENCY vs SUPPLY VOLTAGE 1600 FREQ = VIN VIN = 3.3 V VS = 12 V 1200 FREQ = VIN L = 3.3 mH L = 3.3 mH 1000 f - Frequency - kHz f - Frequency - kHz 1200 1000 FREQ = GND L = 6.8 mH 600 800 FREQ = GND L = 6.8 mH 600 400 400 200 200 0 0 6 1400 1400 800 5.5 VS = 12 V / 200 mA 0 0.1 0.3 0.2 0.4 IO - Load current - mA 0.5 0.6 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 VCC - Supply Voltage - V Figure 9. Figure 10. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 7 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com DETAILED DESCRIPTION VIN VS EN SS IN SW FREQ Current limit and Soft Start Toff Generator Bias Vref = 1.24 V UVLO Thermal Shutdown Ton PWM Generator Gate Driver of Power Transistor COMP GM Amplifier FB Vref PGND Figure 11. Block Diagram The boost converter is designed for output voltages up to 18.5 V with a switch peak current limit of 2.0 A minimum. The device, which operates in a current mode scheme with quasi-constant frequency, is externally compensated for maximum flexibility and stability. The switching frequency is selectable between 650 kHz and 1.2 MHz and the minimum input voltage is 2.3 V. To control the inrush current at start-up a soft-start pin is available. During the on-time, the voltage across the inductor causes the current in it to rise. When the current reaches a threshold value set by the internal GM amplifier, the power transistor is turned off, the energy stored into the inductor is then released and the current flows through the Schottky diode towards the output of the boost converter. The off-time is fixed for a certain VIN and VS, and therefore maintains the same frequency when varying these parameters. However, for different output loads, the frequency may slightly change due to the voltage drop across the Rdson of the power transistor which will have an effect on the voltage across the inductor and thus on tON (tOFF remains fixed). Some slight frequency changes might also appear with a fixed output load due to the fact that the output voltage VS is not sensed directly but via the SW Pin, which affects accuracy. Because of the quasi-constant frequency behavior of the device, the TPS61085 eliminates the need for an internal oscillator and slope compensation, which provides better stability for the system over a wide of input and output voltages range, and more stable and accurate current limiting operation compared to boost converters operating with a conventional PWM scheme. The TPS61085 topology has also the benefits of providing very good load and line regulations, and excellent load transient response. 8 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 Design Procedure The first step in the design procedure is to verify that the maximum possible output current of the boost converter supports the specific application requirements. A simple approach is to estimate the converter efficiency, by taking the efficiency numbers from the provided efficiency curves or to use a worst case assumption for the expected efficiency, e.g. 90%. D = 1- 1. Duty Cycle: VIN ´h VS DI ö æ Iout = ç I swpeak - L ÷ ´ (1 - D ) 2 ø è 2. Maximum output current: 3. Peak switch current: DI L = with I swpeak = I DI L + out 2 1- D VIN ´ D fs ´ L and Iswpeak = converter switch current (minimum switch current limit = 2.0 A) fs = Converter switching frequency (typically 1.2 MHz) L = Selected inductor value η = Estimated converter efficiency (please use the number from the efficiency plots or 90% as an estimation) ΔIL = Inductor peak-to-peak ripple current The peak switch current is the steady state peak switch current that the integrated switch, inductor and external Schottky diode has to be able to handle. The calculation must be done for the minimum input voltage where the peak switch current is the highest. Soft-start The boost converter has an adjustable soft-start to prevent high inrush current during start-up. To minimize the inrush current during start-up an external capacitor connected to the soft-start pin SS is used to slowly ramp up the internal current limit of the boost converter when charged with a constant current. When the EN pin is pulled high, the soft-start capacitor CSS) is immediately charged to 0.3 V. The capacitor is then charged at a constant current of 10 μA typically until the output of the boost converter VS has reached its Power Good threshold (90% of VS nominal value). During this time, the SS voltage directly controls the peak inductor current, starting with 0 A at VSS = 0.3 V up to the full current limit at VSS ≈ 800 mV. The maximum load current is available after the soft-start is completed. The larger the capacitor the slower the ramp of the current limit and the longer the soft-start time. A 100 nF capacitor is usually sufficient for most of the applications. When the EN pin is pulled low, the soft-start capacitor is discharged to ground. Inductor Selection The TPS61085 is designed to work with a wide range of inductors. The main parameter for the inductor selection is the saturation current of the inductor which should be higher than the peak switch current as calculated in the Design Procedure section with additional margin to cover for heavy load transients. An alternative, more conservative, is to choose an inductor with a saturation current at least as high as the maximum switch current limit of 3.2 A. The other important parameter is the inductor dc resistance. Usually, the lower the dc resistance the higher the efficiency. It is important to note that the inductor dc resistance is not the only parameter determining the efficiency. Especially for a boost converter where the inductor is the energy storage element, the type and core material of the inductor influences the efficiency as well. At high switching frequencies of 1.2 MHz inductor core losses, proximity effects and skin effects become more important. Usually, an inductor with a larger form factor gives higher efficiency. The efficiency difference between different inductors can vary between 2% to 10%. For the TPS61085, inductor values between 3 μH and 6 μH are a good choice with a switching frequency of 1.2 MHz, typically 3.3 μH. At 650 kHz we recommend inductors between 6 μH and 13 μH, typically 6.8 μH. Possible inductors are shown in Table 1. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 9 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com Typically, it is recommended that the inductor current ripple is below 20% of the average inductor current. The following equation can therefore be used to calculate the inductor value: 2 æ VS-VIN ö æ h ö æ VIN ö L= ç ×ç ÷×ç ÷ ÷ è VS ø è Iout_max×f ø è 0.35 ø (1) Table 1. Inductor Selection L (μH) SUPPLIER COMPONENT CODE SIZE (L×W×H mm) DCR TYP (mΩ) Isat (A) 3.3 Sumida CDH38D09 4.7 4x4x1 240 1.25 Sumida CDPH36D13 5 × 5 × 1.5 155 3.3 1.36 Sumida CDPH4D19F 5.2 x 5.2 x 2 33 1.5 1.2 MHz 3.3 Sumida CDRH6D12 6.7 x 6.7 x 1.5 62 2.2 4.7 Würth Elektronik 7447785004 5.9 × 6.2 × 3.3 60 2.5 5 Coilcraft MSS7341 7.3 × 7.3 × 4.1 24 2.9 6.8 Sumida CDP14D19 5.2 x 5.2 x 2 50 1 10 Coilcraft LPS4414 4.3 × 4.3 × 1.4 380 1.2 6.8 Sumida CDRH6D12/LD 6.7 x 6.7 x 1.5 95 1.25 10 Sumida CDR6D23 5 × 5 × 2.4 133 1.75 10 Würth Elektronik 744778910 7.3 × 7.3 × 3.2 51 2.2 6.8 Sumida CDRH6D26HP 7 x 7 x 2.8 52 2.9 650 kHz Rectifier Diode Selection To achieve high efficiency, a Schottky type should be used for the rectifier diode. The reverse voltage rating should be higher than the maximum output voltage of the converter. The averaged rectified forward current Iavg, the Schottky diode needs to be rated for, is equal to the output current Iout: I avg = I out (2) Usually a Schottky diode with 2 A maximum average rectified forward current rating is sufficient for most applications. The Schottky rectifier can be selected with lower forward current capability depending on the output current Iout but has to be able to dissipate the power. The dissipated power is the average rectified forward current times the diode forward voltage. PD = Iavg × Vforward Typically the diode should be able to dissipate around 500mW depending on the load current and forward voltage. Table 2. Rectifier Diode Selection 10 CURRENT RATING Iavg Vr Vforward / Iavg SUPPLIER COMPONENT CODE PACKAGE TYPE 750 mA 20 V 0.425 V / 750 mA Fairchild Semiconductor FYV0704S SOT 23 1A 20 V 0.39 V / 1 A NXP PMEG2010AEH SOD 123 1A 20 V 0.52 V / 1 A Vishay Semiconductor B120 SMA 1A 20 V 0.5 V / 1 A Vishay Semiconductor SS12 SMA 1A 20 V 0.44 V / 1 A Vishay Semiconductor MSS1P2L μ-SMP (Low Profile) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 Setting the Output Voltage The output voltage is set by an external resistor divider. Typically, a minimum current of 50 μA flowing through the feedback divider gives good accuracy and noise covering. A standard low side resistor of 18 kΩ is typically selected. The resistors are then calculated as: R2 = Vref » 18k W 70 m A æ VS ö R1 = R 2 ´ ç - 1÷ è Vref ø (3) Compensation (COMP) The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The COMP pin is the output of the internal transconductance error amplifier. Standard values of RCOMP = 13 kΩ and CCOMP = 3.3 nF will work for the majority of the applications. Please refer to Table 3 for dedicated compensation networks giving an improved load transient response. The following equations can be used to calculate RCOMP and CCOMP: 125 × V IN × V S × Cout L × Iout_max R COMP = C COMP = V S × Cout 5 × Iout_max × R COMP (4) Table 3. Recommended Compensation Network Values at High/Low Frequency FREQUENCY L VS 15 V High (1.2 MHz) 3.3 µH 12 V 9V 15 V Low (650 kHz) 6.8 µH 12 V 9V VIN ± 20% RCOMP CCOMP 5V 82 kΩ 1.1 nF 3.3 V 75 kΩ 1.6 nF 5V 51 kΩ 1.1 nF 3.3 V 47 kΩ 1.6 nF 5V 30 kΩ 1.1 nF 3.3 V 27 kΩ 1.6 nF 5V 43 kΩ 2.2 nF 3.3 V 39 kΩ 3.3 nF 5V 27 kΩ 2.2 nF 3.3 V 24 kΩ 3.3 nF 5V 15 kΩ 2.2 nF 3.3 V 13 kΩ 3.3 nF Table 3 gives conservatives Rcomp and Comp values for certain inductors, input and output voltages providing a very stable system. For a faster response time, a higher Rcomp value can be used to enlarge the bandwidth, as well as a slightly lower value of Ccomp to keep enough phase margin. These adjustments should be performed in parallel with the load transient response monitoring of TPS61085. Input Capacitor Selection For good input voltage filtering low ESR ceramic capacitors are recommended. TPS61085 has an analog input IN. Therefore, a 1 μF bypass is highly recommended as close as possible to the IC from IN to GND. One 10 μF ceramic input capacitors are sufficient for most of the applications. For better input voltage filtering this value can be increased. Refer to Table 4 and typical applications for input capacitor recommendations. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 11 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com Output Capacitor Selection For best output voltage filtering a low ESR output capacitor like ceramic capcaitor is recommended. Two 10 μF ceramic output capacitors (or one 22 μF) work for most of the applications. Higher capacitor values can be used to improve the load transient response. Refer to Table 4 for the selection of the output capacitor. Table 4. Rectifier Input and Output Capacitor Selection CAPACITOR VOLTAGE RATING SUPPLIER COMPONENT CODE CIN 10 μF/1206 16 V Taiyo Yuden EMK212 BJ 106KG IN bypass 1 μF/0603 16 V Taiyo Yuden EMK107 BJ 105KA COUT 10 μF/1206 25 V Taiyo Yuden TMK316 BJ 106KL Frequency Select Pin (FREQ) The frequency select pin FREQ allows to set the switching frequency of the device to 650 kHz (FREQ = low) or 1.2 MHz (FREQ = high). Higher switching frequency improves load transient response but reduces slightly the efficiency. The other benefits of higher switching frequency are a lower output ripple voltage. Usually, it is recommended to use 1.2 MHz switching frequency unless light load efficiency is a major concern. Undervoltage Lockout (UVLO) To avoid mis-operation of the device at low input voltages an undervoltage lockout is included that disables the device, if the input voltage falls below 2.2 V. Thermal Shutdown A thermal shutdown is implemented to prevent damages due to excessive heat and power dissipation. Typically the thermal shutdown threshold is 150°C. When the thermal shutdown is triggered the device stops switching until the temperature falls below typically 136°C. Then the device starts switching again. 12 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 APPLICATION INFORMATION L 3.3 mH VIN 3.3 V ±20% 6 Cin 10 mF 16 V Cby 1 mF 16 V 5 IN 3 VS 12 V/600 mA max SW EN 2 R1 156 kW Cout 1 R2 18 kW 2* 10 mF 25 V FB 7 FREQ 4 D PMEG2010AEH COMP Rcomp 51kW 8 GND SS Css TPS61085 Ccomp 1.6 nF 100 nF Figure 12. Typical Application, 3.3 V to 12 V (fsw = 1.2 MHz) L 6.8 mH VIN 3.3 V ±20% 6 Cin 10 mF 16 V Cby 1 mF 16 V SW IN 3 EN 5 D PMEG2010AEH VS 12 V/600 mA max 2 R1 156 kW Cout 1 R2 18 kW 2* 10 mF 25 V FB 7 FREQ COMP 4 Rcomp 24 kW 8 GND SS TPS61085 Css Ccomp 3.3 nF 100 nF Figure 13. Typical Application, 3.3 V to 12 V (fsw = 650 kHz) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 13 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com L 3.3 mH VIN 3.3 V ±20% 6 Cin 10 mF 16 V Cby 1 mF 16 V SW IN 3 EN 5 D PMEG2010AEH VS 9 V/800 mA max 2 R1 113 kW Cout 1 R2 18 kW 2* 10 mF 25 V FB 7 FREQ COMP 4 Rcomp 28 kW 8 GND SS Css TPS61085 Ccomp 1.6 nF 100 nF Figure 14. Typical Application, 3.3 V to 9 V (fsw = 1.2 MHz) L 6.8 mH VIN 3.3 V ±20% 6 Cin 10 mF 16 V Cby 1 mF 16 V SW IN 3 EN 5 D PMEG2010AEH VS 9 V/800 mA max 2 R1 113 kW Cout 1 R2 18 kW 2* 10 mF 25 V FB 7 FREQ COMP 4 Rcomp 14 kW 8 GND SS TPS61085 Css Ccomp 3.3 nF 100 nF Figure 15. Typical Application, 3.3 V to 9 V (fsw = 650 kHz) 14 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 Riso 10 kW L 6.8 mH VIN 3.3 V ±20% Cby 1 mF/16 V 6 3 Cin 10 mF 16 V 7 4 IN SW EN FB FREQ COMP GND SS D PMEG2010AEH 5 Ciso 1 mF/ 25 V 2 VS 12 V/300 mA BC857C Rff1 156 kW 1 8 Css 100nF TPS61085 Rff2 18 kW Rcomp 24 kW Cout 2*10 mF 25 V Ccomp 3.3 nF Figure 16. Typical Application with External Load Disconnect Switch TFT LCD APPLICATION Vgl -7 V/ 20 mA T1 BC857B -Vs C4 100nF/ 50V D2 BAT54S C3 100 nF 50 V C2 R8 7 kW 470 nF 25 V C1 1µF/ 35V D3 BAT54S D1 BZX84C7V5 D4 BAT54S C6 470 nF 50 V D5 BAT54S C5 100 nF 50 V D6 BAT54S R10 13 kW 2.Vs C7 470 nF 50 V Vgh 20 V/20 mA T2 BC850B 3.Vs C8 1 µF 35 V D8 BZX84C 20V D7 BAT54S L 3.3µH VIN 3.3 V± 20% 6 Cin 10 µF 16 V Cby 1 µF 16 V 5 VIN SW EN FB 3 D PMEG2010AEH R1 113 kW 2 7 R2 18 kW 1 FREQ COMP SS GND TPS 61085 Cout 2*10 µF 25 V Rcomp 28 kW 8 4 VS 9 V/500 mA Css Ccomp 1.6 nF 100 nF Figure 17. Typical Application 3.3 V to 9 V (fsw = 1.2 MHz) for TFT LCD with External Charge Pumps (VGH, VGL) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 15 TPS61085T SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 www.ti.com WHITE LED APPLICATIONS L 6.8 mH VIN 3.3 V ± 20% Cin 10 mF/ 16 V Cby 1 mF/ 16 V D PMEG2010AEH 6 3 7 4 IN SW EN FB FREQ COMP PGND SS 5 Dz BZX84C 18 V VS 300 mA 3S3P wLED LW E67C 2 Cout 2* 10 mF/ 25 V 1 Rcomp 24 kW 8 Css 100 nF TPS61085 Rsense 14 W Ccomp 3.3 nF Figure 18. Simple Application (3.3V input - fsw = 650 kHz) for wLED Supply (3S3P) (with optional clamping Zener diode) L 6.8 mH VIN 3.3 V ± 20% Cin 10 mF/ 16 V Cby 1 mF/ 16 V D PMEG2010AEH 6 3 7 4 IN SW EN FB FREQ COMP PGND SS PWM 100 Hz to 500 Hz TPS61085 5 Dz BZX84C 18 V 2 VS 300 mA 3S3P wLED LW E67C Cout 2* 10 mF/ 25 V 1 Rcomp 24 kW 8 Css 100 nF Ccomp 3.3 nF Rsense 14 W Figure 19. Simple Application (3.3V input - fsw = 650 kHz) for wLED Supply (3S3P) with Adjustable Brightness Control using a PWM Signal on the Enable Pin (with optional clamping Zener diode) 16 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T TPS61085T www.ti.com SLVSA41A – NOVEMBER 2009 – REVISED DECEMBER 2009 L 6.8 mH VIN 3.3 V ± 20% Cin 10 mF/ 16 V Cby 1 mF/ 16 V D PMEG2010AEH 6 3 7 4 IN SW EN FB FREQ COMP PGND SS TPS61085 5 Dz BZX84C 18 V 3S3P wLED LW E67C 2 Cout 2* 10 mF/ 25 V R1 1 Rcomp 24 kW 8 Css 100 nF Ccomp 3.3 nF VS 300 mA 180 kW R2 127 kW Rsense 14 W Analog Brightness Control 3.3 V ~ wLED off 0 V ~ lled = 30 mA (each string) PWM Signal Can be used Swinging from 0 V to 3.3 V Figure 20. Simple Application (3.3V input - fsw = 650 kHz) for wLED Supply (3S3P) with Adjustable Brightness Control using an Analog Signal on the Feedback Pin (with optional clamping Zener diode) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61085T 17 PACKAGE OPTION ADDENDUM www.ti.com 16-Aug-2012 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/ Ball Finish MSL Peak Temp TPS61085TDGKR ACTIVE VSSOP DGK 8 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS61085TPWR ACTIVE TSSOP PW 8 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM (3) Samples (Requires Login) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 16-Aug-2012 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant TPS61085TDGKR VSSOP DGK 8 2000 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 TPS61085TPWR TSSOP PW 8 2000 330.0 12.4 7.0 3.6 1.6 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 16-Aug-2012 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) TPS61085TDGKR VSSOP DGK 8 2000 367.0 367.0 35.0 TPS61085TPWR TSSOP PW 8 2000 367.0 367.0 35.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46C and to discontinue any product or service per JESD48B. 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