LM6211 Low Noise, RRO Operational Amplifier with CMOS Input and 24V Operation General Description Features The LM6211 is a wide bandwidth, low noise op amp with a wide supply voltage range and a low input bias current. The LM6211 operates with a single supply voltage of 5V to 24V, is unity gain stable, has a ground-sensing CMOS input stage, and offers rail-to-rail output swing. The LM6211 is designed to provide optimal performance in high voltage, low noise systems. The LM6211 has a unity gain bandwidth of 20 MHz and an input referred voltage at 10 kHz. The LM6211 noise density of 5.5 nV/ achieves these specifications with a low supply current of only 1 mA. The LM6211 has a low input bias current of 2.3 pA, an output short circuit current of 25 mA and a slew rate of 5.6 V/us. The LM6211 also features a low commonmode input capacitance of 5.5 pF which makes it ideal for use in wide bandwidth and high gain circuits. The LM6211 is well suited for low noise applications that require an op amp with very low input bias currents and a large output voltage swing, like active loop-filters for wide-band PLLs. A low total harmonic distortion, 0.01% at 1 kHz with loads as high as 600Ω, also makes the LM6211 ideal for high fidelity audio and microphone amplifiers. The LM6211 is available in the small SOT package, allowing the user to implement ultra-small and cost effective board layouts. (Typical 24V supply unless otherwise noted) n Supply voltage range 5V to 24V n Input referred voltage noise 5.5 nV/ n Unity gain bandwidth 20 MHz n 1/f corner frequency 400 Hz n Slew rate 5.6 V/µs n Supply current 1.05 mA n Low input capacitance 5.5 pF n Temperature range -40˚C to 125˚C n Total harmonic distortion 0.01% @ 1 kHz, 600Ω n Output short circuit current 25 mA Applications n n n n PLL loop filters Low noise active filters Strain gauge amplifiers Low noise microphone amplifiers Typical Application 20120303 20120304 © 2006 National Semiconductor Corporation DS201203 www.national.com LM6211 Low Noise, RRO Op-Amp with CMOS Input and 24V Operation May 2006 LM6211 Absolute Maximum Ratings (Note 1) Junction Temperature (Note 3) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Soldering Information +150˚C Infrared or Convection (20 sec) 235˚C Wave Soldering Lead Temp. (10 sec) 260˚C ESD Tolerance (Note 2) Human Body Model 2000V Machine Model Operating Ratings (Note 1) 200V Temperature Range ± 0.3V VIN Differential + − Supply Voltage (VS = V – V ) 25V Voltage at Input/Output pins V+ +0.3V, V− −0.3V Storage Temperature Range −65˚C to +150˚C 5V Electrical Characteristics −40˚C to +125˚C Supply Voltage (VS = V+ – V−) 5V to 24V Package Thermal Resistance (θJA (Note 3)) 5-Pin SOT23 178˚C/W (Note 4) Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 5V, V− = 0V, VCM = VO = V+/2. Boldface limits apply at the temperature extremes. Symbol Parameter Conditions Min (Note 6) Typ (Note 5) Max (Note 6) 0.1 ± 2.5 ± 2.8 VOS Input Offset Voltage VCM = 0.5V TC VOS Input Offset Average Drift VCM = 0.5V (Note 7) IB Input Bias Current VCM = 0.5V (Notes 8, 9) 0.5 IOS Input Offset Current VCM = 0.5V 0.1 CMRR Common Mode Rejection Ratio 0 V ≤ VCM ≤ 3V 0.4 V ≤ VCM ≤ 2.3 V 83 70 98 PSRR Power Supply Rejection Ratio V+ = 5V to 24V, VCM = 0.5V 85 78 98 V+ = 4.5V to 25V, VCM = 0.5V 80 95 2 CMVR Input Common-Mode Voltage Range CMRR ≥ 65 dB CMRR ≥ 60 dB 0 0 AVOL Large Signal Voltage Gain VO = 0.35V to 4.65, RL = 2 kΩ to V+/2 82 80 110 VO = 0.25V to 4.75, RL = 10 kΩ to V+/2 85 82 110 VO Output Swing High Output Swing Low IOUT Output Short Circuit Current IS Supply Current SR Slew Rate GBW Gain Bandwidth Product en Input-Referred Voltage Noise in Input-Referred Current Noise www.national.com Units mV µV/C 5 10 pA nA pA dB dB 3.3 2.4 dB RL = 2 kΩ to V+/2 50 150 165 RL = 10 kΩ to V+/2 20 85 90 RL = 2 kΩ to V+/2 39 150 170 RL = 10 kΩ to V+/2 13 85 90 Sourcing to V+/2 VID = 100 mV (Note 10) 13 10 16 Sinking to V+/2 VID = −100 mV (Note 10) 20 10 30 0.96 V mV from rail mA 1.10 1.25 mA AV = +1, 10% to 90% (Note 11) 5.5 V/µs 17 MHz f = 10 kHz 5.5 f = 1 kHz 6.0 f = 1 kHz 0.01 2 nV/ pA/ (Note 4) (Continued) Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 5V, V− = 0V, VCM = VO = V+/2. Boldface limits apply at the temperature extremes. Symbol THD Parameter Total Harmonic Distortion Conditions Min (Note 6) AV = 2, RL = 600Ω to V+/2 24V Electrical Characteristics Typ (Note 5) Max (Note 6) Units 0.01 % (Note 4) Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 24V, V− = 0V, VCM = VO = V+/2. Boldface limits apply at the temperature extremes. Symbol Parameter Conditions VOS Input Offset Voltage VCM = 0.5V TC VOS Input Offset Average Drift VCM = 0.5V (Note 7) IB Input Bias Current VCM = 0.5V (Notes 8, 9) IOS Input Offset Current VCM = 0.5V CMRR Common Mode Rejection Ratio 0 ≤ VCM ≤ 21V 0.4 ≤ VCM ≤ 20V PSRR Power Supply Rejection Ratio Min (Note 6) Typ (Note 5) Max (Note 6) 0.25 ± 2.7 ± 3.0 ±2 2 + V = 5V to 24V, VCM = 0.5V + 105 85 78 98 98 V = 4.5V to 25V, VCM = 0.5V 80 Input Common-Mode Voltage Range CMRR ≥ 65 dB CMRR ≥ 60 dB 0 0 AVOL Large Signal Voltage Gain VO = 1.5V to 22.5V, RL = 2 kΩ to V+/2 82 77 120 VO = 1V to 23V, RL = 10 kΩ to V+/2 85 82 120 Output Swing High Output Swing Low IOUT Output Short Circuit Current IS Supply Current SR Slew Rate GBW Gain Bandwidth Product en Input-Referred Voltage Noise pA nA pA dB dB CMVR VO mV µV/C 25 10 0.1 85 70 Units 21.5 20.5 dB RL = 2 kΩ to V+/2 212 400 520 RL = 10 kΩ to V+/2 48 150 165 RL = 2 kΩ to V+/2 150 350 420 RL = 10 kΩ to V+/2 38 150 170 Sourcing to V+/2 VID = 100 mV (Note 10) 20 15 25 Sinking to V+/2 VID = −100 mV (Note 10) 30 20 38 1.05 V mV from rail mA 1.25 1.40 mA AV = +1, VO = 18 VPP 10% to 90% (Note 11) 5.6 V/µs 20 MHz f = 10 kHz 5.5 f = 1 kHz 6.0 in Input-Referred Current Noise f = 1 kHz 0.01 THD Total Harmonic Distortion AV = 2, RL = 2 kΩ to V+/2 0.01 3 nV/ pA/ % www.national.com LM6211 5V Electrical Characteristics LM6211 Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics Tables. Note 2: Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 200 pF. Note 3: The maximum power dissipation is a function of TJ(MAX), θJA, and TA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/θJA . All numbers apply for packages soldered directly onto a PC board. Note 4: Electrical table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device. Note 5: Typical values represent the most likely parametric norm at the time of characterization. Note 6: Limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlations using the Statistical Quality Control (SQC) method. Note 7: Offset voltage average drift is determined by dividing the change in VOS at the temperature extremes into the total temperature change. Note 8: Positive current corresponds to current flowing into the device. Note 9: Input bias current is guaranteed by design. Note 10: The device is short circuit protected and can source or sink its limit currents continuously. However, care should be taken such that when the output is driving short circuit currents, the inputs do not see more than ± 0.3V differential voltage. Note 11: Slew rate is the average of the rising and falling slew rates. Connection Diagram 5-Pin SOT23 20120301 Top View Ordering Information Package 5-Pin SOT-23 www.national.com Part Number LM6211MF LM6211MFX Package Marking Transport Media 1k Units Tape and Reel AT1A 3k Units Tape and Reel 4 NSC Drawing MF05A Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS, − V = 0 V, VCM = VS/2. VOS vs. Supply Voltage Supply Current vs. Supply Voltage 20120319 20120318 VOS vs. VCM VOS vs. VCM 20120321 20120320 Input Bias Current vs. VCM Input Bias Current vs. VCM 20120351 20120350 5 www.national.com LM6211 Typical Performance Characteristics LM6211 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS, V− = 0 V, VCM = VS/2. (Continued) Input Bias Current vs. VCM Input Bias Current vs. VCM 20120352 20120353 Sourcing Current vs. Supply Voltage Sinking Current vs. Supply Voltage 20120334 20120333 Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage 20120330 www.national.com 20120332 6 V− = 0 V, VCM = VS/2. (Continued) Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage 20120331 20120329 Sourcing Current vs. Output Voltage Sinking Current vs. Output Voltage 20120328 20120327 Sourcing Current vs. Output Voltage Sinking Current vs. Output Voltage 20120325 20120326 7 www.national.com LM6211 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS, LM6211 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS, V− = 0 V, VCM = VS/2. (Continued) Open Loop Gain and Phase with Capacitive Load Open Loop Gain and Phase with Resistive Load 20120309 20120308 Input Referred Voltage Noise vs. Frequency THD+N vs. Frequency 20120304 20120317 THD+N vs. Output Amplitude THD+N vs. Output Amplitude 20120315 20120316 www.national.com 8 V− = 0 V, VCM = VS/2. (Continued) Slew Rate vs. Supply Voltage Overshoot and Undershoot vs. Capacitive Load 20120313 20120314 Small Signal Transient Response Large Signal Transient Response 20120322 20120324 Phase Margin vs. Capacitive Load (Stability) Phase Margin vs. Capacitive Load (Stability) 20120310 20120311 9 www.national.com LM6211 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS, LM6211 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS, V− = 0 V, VCM = VS/2. (Continued) Closed Loop Output Impedance vs. Frequency PSRR vs. Frequency 20120305 20120312 CMRR vs. Frequency 20120306 www.national.com 10 ADVANTAGES OF THE LM6211 Small Size The small footprint of the LM6211 package saves space on printed circuit boards, and enables the design of smaller and more compact electronic products. Long traces between the signal source and the op amp make the signal path susceptible to noise. By using a physically smaller package, the LM6211 can be placed closer to the signal source, reducing noise pickup and enhancing signal integrity High Supply Voltage, Low Power Operation The LM6211 has performance guaranteed at supply voltages of 5V and 24V. The LM6211 is guaranteed to be operational at all supply voltages between 5V and 24V. In this large range of operation, the LM6211 draws a fairly constant supply current of 1 mA, while providing a wide bandwidth of 20 MHz. The wide operating range makes the LM6211 a versatile choice for a variety of applications ranging from portable instrumentation to industrial control systems. STABILITY OF OP AMP CIRCUITS Stability and Capacitive Loading The LM6211 is designed to be unity gain stable for moderate capacitive loads, around 100 pF. That is, if connected in a unity gain buffer configuration, the LM6211 will resist oscillation unless the capacitive load is higher than about 100 pF. For higher capacitive loads, the phase margin of the op amp reduces significantly and it tends to oscillate. This is because an op amp cannot be designed to be stable for high capacitive loads without either sacrificing bandwidth or supplying higher current. Hence, for driving higher capacitive loads, the LM6211 needs to be externally compensated. Low Input Referred Noise The LM6211 has very low flatband input referred voltage . The 1/f corner frequency, also very low, noise, 5.5 nV/ is about 400 Hz. The CMOS input stage allows for an extremely low input current (2 pA) and a very low input ). This allows the referred current noise (0.01 pA/ LM6211 to maintain signal fidelity and makes it ideal for audio, wireless or sensor based applications. Low Input Bias Current and High Input Impedance The LM6211 has a CMOS input stage, which allows it to have very high input impedance, very small input bias currents (2 pA) and extremely low input referred current noise ). This level of performance is essential for op (0.01 pA/ amps used in sensor applications, which deal with extremely low currents of the order of a few nanoamperes. In this case, the op amp is being driven by a sensor, which typically has a source impedance of tens of MΩ. This makes it essential for the op amp to have a much higher impedance. Low Input Capacitance The LM6211 has a comparatively small input capacitance for a high voltage CMOS design. Low input capacitance is very beneficial in terms of driving large feedback resistors, required for higher closed loop gain. Usually, high voltage CMOS input stages have a large input capacitance, which when used in a typical gain configuration, interacts with the feedback resistance to create an extra pole. The extra pole causes gain-peaking and can compromise the stability of the op amp. The LM6211 can, however, be used with larger resistors due to its smaller input capacitance, and hence provide more gain without compromising stability. This also makes the LM6211 ideal for wideband transimpedance amplifiers, which require a wide bandwidth, low input referred noise and low input capacitance. 20120337 FIGURE 1. Gain vs. Frequency for an Op Amp An op amp, ideally, has a dominant pole close to DC, which causes its gain to decay at the rate of 20 dB/decade with respect to frequency. If this rate of decay, also known as the rate of closure (ROC), remains at 20 dB/decade at the unity gain bandwidth of the op amp, the op amp is stable. If, however, a large capacitance is added to the output of the op amp, it combines with the output impedance of the op amp to create another pole in its frequency response before its unity gain frequency (Figure 1). This increases the ROC to 40 dB/decade and causes instability. In such a case a number of techniques can be used to restore stability to the circuit. The idea behind all these schemes is to modify the frequency response such that it can be restored to a ROC of 20 dB/decade, which ensures stability. RRO, Ground Sensing and Current Limiting The LM6211 has a rail-to-rail output stage, which provides the maximum possible output dynamic range. This is especially important for applications requiring a large output swing, like wideband PLL synthesizers which need an active loop filter to drive a wide frequency range VCO. The input common mode range includes the negative supply rail which allows direct sensing at ground in a single supply operation. The LM6211 also has a short circuit protection circuit which limits the output current to about 25 mA sourcing and 38 mA sinking, and allows the LM6211 to drive short circuit loads indefinitely. However, while driving short circuit loads care 11 www.national.com LM6211 should be taken to prevent the inputs from seeing more than ± 0.3V differential voltage, which is the absolute maximum differential input voltage. Application Notes LM6211 Application Notes Compensation by External Resistor (Continued) In some applications it is essential to drive a capacitive load without sacrificing bandwidth. In such a case, in the loop compensation is not viable. A simpler scheme for compensation is shown in Figure 3. A resistor, RISO, is placed in series between the load capacitance and the output. This introduces a zero in the circuit transfer function, which counteracts the effect of the pole formed by the load capacitance, and ensures stability. In the Loop Compensation Figure 2 illustrates a compensation technique, known as ‘in the loop’ compensation, that employs an RC feedback circuit within the feedback loop to stabilize a non-inverting amplifier configuration. A small series resistance, RS, is used to isolate the amplifier output from the load capacitance, CL, and a small capacitance, CF, is inserted across the feedback resistor to bypass CL at higher frequencies. 20120356 FIGURE 3. Compensation By Isolation Resistor The value of RISO to be used should be decided depending on the size of CL and the level of performance desired. Values ranging from 5Ω to 50Ω are usually sufficient to ensure stability. A larger value of RISO will result in a system with lesser ringing and overshoot, but will also limit the output swing and the short circuit current of the circuit. 20120338 FIGURE 2. In the Loop Compensation The values for RS and CF are decided by ensuring that the zero attributed to CF lies at the same frequency as the pole attributed to CL. This ensures that the effect of the second pole on the transfer function is compensated for by the presence of the zero, and that the ROC is maintained at 20 dB/decade. For the circuit shown in Figure 2 the values of RS and CF are given by Equation (1). Table 1 shows different values of RS and CF that need to be used for maintaining stability with different values of CL, as well as the phase margins to be expected. RF and RIN are assumed to be 10 kΩ, RL is taken as 2 kΩ, while ROUT is taken to be 60Ω. Stability and Input Capacitance In certain applications, for example I-V conversion, transimpedance photodiode amplification and buffering the output of current-output DAC, capacitive loading at the input of the op amp can endanger stability. The capacitance of the source driving the op amp, the op amp input capacitance and the parasitic/wiring capacitance contribute to the loading of the input. This capacitance, CIN, interacts with the feedback network to introduce a peaking in the closed loop gain of the circuit, and hence causes instability. (1) TABLE 1. CL (pF) RS (Ω) CF (pF) Phase Margin (˚) 250 60 4.5 39.8 300 60 5.4 49.5 500 60 9 53.1 20120349 FIGURE 4. Compensating for Input Capacitance This peaking can be eliminated by adding a feedback capacitance, CF, as shown in Figure 4. This introduces a zero in the feedback network, and hence a pole in the closed loop response, and thus maintains stability. An optimal value of CF is given by Equation (2). A simpler approach is to select CF = (R1/R2)CIN for a 90˚ phase margin. This approach, however, limits the bandwidth excessively. Although this methodology provides circuit stability for any load capacitance, it does so at the price of bandwidth. The closed loop bandwidth of the circuit is now limited by RS and C F. www.national.com 12 ACTIVE LOOP FILTER FOR PLLs A typical phase locked loop, or PLL, functions by creating a negative feedback loop in terms of the phase of a signal. A simple PLL consists of three main components: a phase detector, a loop filter and a voltage controlled oscillator (VCO). The phase detector compares the phase of the output of the PLL with that of a reference signal, and feeds the error signal into the loop filter, thus performing negative feedback. The loop filter performs the important function of averaging (or low-pass filtering) the error and providing the VCO with a DC voltage, which allows the VCO to modify its frequency such that the error is minimized. The performance of the loop filter affects a number of specifications of the PLL, like its frequency range, locking time and phase noise. Since a loop filter is a very noise sensitive application, it is usually suggested that only passive components be used in its design. Any active devices, like discrete transistors or op amps, would add significantly to the noise of the circuit and would hence worsen the in-band phase noise of the PLL. But newer and faster PLLs, like National’s LMX2430, have a power supply voltage of less than 3V, which limits the phasedetector output of the PLL. If a passive loop filter is used with such circuits, then the DC voltage that can be provided to the VCO is limited to couple of volts. This limits the range of frequencies for which the VCO, and hence the PLL, is functional. In certain applications requiring a wider operating range of frequencies for the PLL, like set-top boxes or base stations, this level of performance is not adequate and requires active amplification, hence the need for active loop filters. Figure 7 shows the LM6211 used with the LMX2430 to create an RF frequency synthesizer. The LMX2430 detects the PLL output, compares it with its internal reference clock and outputs the phase error in terms of current spikes. The LM6211 is used to create a loop filter which averages the error and provides a DC voltage to the VCO. The VCO generates a sine wave at a frequency determined by the DC voltage at its input. This circuit can provide output signal frequencies as high as 2 GHz, much higher than a comparative passive loop filter. Compared to a similar passive loop filter, the LM6211 doesn’t add significantly to the phase noise of the PLL, except at the edge of the loop bandwidth, as shown in Figure 6. A peaking of loop gain is expected, since the loop filter is deliberately designed to have a wide bandwidth and a low phase margin so as to minimize locking time. An active loop filter typically consists of an op amp, which provides the gain, accompanied by a three or four pole RC filter. The non-inverting input of the op amp is biased to a fixed value, usually the mid-supply of the PLL, while a feedback network provides the gain as well as one, or two, poles for low pass filtering. Figure 5 illustrates a typical active loop filter. 20120303 20120355 FIGURE 5. A Typical Active Loop Filter FIGURE 6. Effect of LM6211 on Phase Noise of PLL 13 www.national.com LM6211 Certain performance characteristics are essential for an op amp if it is to be used in a PLL loop filter. Low input referred voltage and current noise are essential, as they directly affect the noise of the filter and hence the phase noise of the PLL. Low input bias current is also important, as bias current affects the level of ‘reference spurs’, artifacts in the frequency spectrum of the PLL caused by mismatch or leakage at the output of the phase detector. A large input and output swing is beneficial in terms of increasing the flexibility in biasing the op amp. The op amp can then be biased such that the output range of the PLL is mapped efficiently onto the input range of the VCO. With a CMOS input, ultra low input bias currents (2 pA) and ), the LM6211 low input referred voltage noise (5.5 nV/ is an ideal op amp for using in a PLL active loop filter. The LM6211 has a ground sensing input stage, a rail-to-rail output stage, and an operating supply range of 5V - 24V, which makes it a versatile choice for the design of a wide variety of active loop filters. Typical Applications LM6211 Typical Applications (Continued) 20120336 FIGURE 7. LM6211 in the Active Loop Filter for LMX2430 ADC INPUT DRIVER A typical application for a high performance op amp is as an ADC driver, which delivers the analog signal obtained from sensors and actuators to ADCs for conversion to the digital domain and further processing. Important requirements in this application are a slew rate high enough to drive the ADC input and low input referred voltage and current noise. If an op amp is used with an ADC, it is critical that the op amp noise does not affect the dynamic range of the ADC. The LM6211, with low input referred voltage and current noise, provides a great solution for this application. For example, the LM6211 can be used to drive an ADS121021, a 12-bit ADC from National. If it provides a gain of 10 to a maximum input signal amplitude of 100 mV, for a bandwidth as wide as 100 kHz, the average noise seen at the input of the ADC is only 44.6 µVrms. Hence the dynamic range of the ADC, www.national.com measured in Effective Number of Bits or ENOB, is only reduced by 0.3 bits, despite amplifying the input signal by a gain of 10. Low input bias currents and high input impedance also help as they prevent the loading of the sensor and allow the measurement system to function over a large range. Figure 8 shows a circuit for monitoring fluid pressure in a hydraulic system, in which the LM6211 is used to sense the error voltage from the pressure sensor. Two LM6211 amplifiers are used to make a difference amplifier which senses the error signal, amplifies it by a gain of 100, and delivers it to the ADC input. The ADC converts the error voltage into a pressure reading to be displayed and drives the DAC, which changes the voltage driving the resistance bridge sensor. This is used to control the gain of the pressure measurement circuit, such that the range of the sensor can be modified to obtain the best resolution possible. 14 LM6211 Typical Applications (Continued) 20120335 FIGURE 8. Hydraulic Pressure Monitoring System DAC OUTPUT AMPLIFIER Op amps are often used to improve a DAC’s output driving capability. High performance op amps are required as I-V converters at the outputs of high resolution current output DACs. Since most DACs operate with a single supply of 5V, a rail-to-rail output swing is essential for this application. A low offset voltage is also necessary to prevent offset errors in the waveform generated. Also, the output impedance of DACs is quite high, more than a few kΩ in some cases, so it is also advisable for the op amp to have a low input bias current. An op amp with a high input impedance also prevents the loading of the DAC, and hence, avoids gain errors. The op amp should also have a slew rate which is fast enough to not affect the settling time of the DAC output. The LM6211, with a CMOS input stage, ultra low input bias current, a wide bandwidth (20 MHz) and a rail-to-rail output swing for a supply voltage of 24V is an ideal op amp for such an application. Figure 9 shows a typical circuit for this application. The op amp is usually expected to add another time constant to the system, which worsens the settling time, but the wide bandwidth of the LM6211 (20 MHz) allows the system performance to improve without any significant degradation of the settling time. 20120340 FIGURE 9. DAC Driver Circuit 15 www.national.com LM6211 Typical Applications (Continued) AUDIO PREAMPLIFIER With low input referred voltage noise, low supply voltage and low supply current, and low harmonic distortion, the LM6211 is ideal for audio applications. Its wide unity gain bandwidth allows it to provide large gain over a wide frequency range and it can be used to design a preamplifier to drive a load of as low as 600Ω with less than 0.001% distortion. Two amplifier circuits are shown in Figure 10 and Figure 11. Figure 10 is an inverting amplifier, with a 10 kΩ feedback resistor, R2, and a 1 kΩ input resistor, R1, and hence provides a gain of −10. Figure 11 is a non-inverting amplifier, using the same values for R1 and R2, and provides a gain of 11. In either of these circuits, the coupling capacitor CC1 decides the lower frequency at which the circuit starts providing gain, while the feedback capacitor CF decides the frequency at which the gain starts dropping off. Figure 12 shows the frequency response of the circuit in Figure 10 with different values of CF. 20120343 FIGURE 12. Frequency Response of the Non-Inverting Preamplifier TRANSIMPEDANCE AMPLIFIER A transimpedance amplifier converts a small input current into a voltage. This current is usually generated by a photodiode. The transimpedance gain, measured as the ratio of the output voltage to the input current, is expected to be large and wide-band. Since the circuit deals with currents in the range of a few nA, low noise performance is essential. The LM6211, being a CMOS input op amp, provides a wide bandwidth and low noise performance while drawing very low input bias current, and is hence ideal for transimpedance applications. A transimpedance amplifier is designed on the basis of the current source driving the input. A photodiode is a very common capacitive current source, which requires transimpedance gain for transforming its miniscule current into easily detectable voltages. The photodiode and amplifier’s gain are selected with respect to the speed and accuracy required of the circuit. A faster circuit would require a photodiode with lesser capacitance and a faster amplifier. A more sensitive circuit would require a sensitive photodiode and a high gain. A typical transimpedance amplifier is shown in Figure 13. The output voltage of the amplifier is given by the equation VOUT = −IINRF. Since the output swing of the amplifier is limited, RF should be selected such that all possible values of IIN can be detected. The LM6211 has a large gain-bandwidth product (20 MHz), which enables high gains at wide bandwidths. A rail-to-rail output swing at 24V supply allows detection and amplification of a wide range of input currents. A CMOS input stage with negligible input current noise and low input voltage noise allows the LM6211 to provide high fidelity amplification for wide bandwidths. These properties make the LM6211 ideal for systems requiring wide-band transimpedance amplification. 20120341 FIGURE 10. Inverting Audio Amplifier 20120342 FIGURE 11. Non-Inverting Audio Preamplifier www.national.com 16 An essential component for obtaining a maximally flat response is the feedback capacitor, CF. The capacitance seen at the input of the amplifier, CIN, combined with the feedback resistor, RF, generates a phase lag which causes gainpeaking and can destabilize the circuit. CIN is usually just the sum of CD and CCM. The feedback capacitor CF creates a pole, fP in the noise gain of the circuit, which neutralizes the zero in the noise gain, fZ, created by the combination of RF and CIN. If properly positioned, the noise gain pole created by CF can ensure that the slope of the gain remains at 20 dB/decade till the unity gain frequency of the amplifier is reached, thus ensuring stability. As shown in Figure 16, fP is positioned such that it coincides with the point where the noise gain intersects the op amp’s open loop gain. In this case, fP is also the overall 3 dB frequency of the transimpedance amplifier. The value of CF needed to make it so is given by Equation (2). A larger value of CF causes excessive reduction of bandwidth, while a smaller value fails to prevent gain peaking and maintain stability. (Continued) 20120344 FIGURE 13. Photodiode Transimpedance Amplifier (2) Calculating CF from Equation (2) can sometimes return unreasonably small values ( < 1 pF), especially for high speed applications. In these cases, it is often more practical to use the circuit shown in Figure 15 in order to allow more reasonable values. In this circuit, the capacitance CF’ is (1+ RB/RA) times the effective feedback capacitance, CF. A larger capacitor can now be used in this circuit to obtain a smaller effective capacitance. The following parameters are used to design a transimpedance amplifier: the amplifier gain-bandwidth product, A0; the amplifier input capacitance, CCM; the photodiode capacitance, CD; the transimpedance gain required, RF; and the amplifier output swing. Once a feasible RF is selected using the amplifier output swing, these numbers can be used to design an amplifier with the desired transimpedance gain and a maximally flat frequency response. The input common-mode capacitance with respect to VCM for the LM6211 is give in Figure 14. 20120347 FIGURE 15. Modifying CF 20120354 For example, if a CF of 0.5 pF is needed, while only a 5 pF capacitor is available, RB and RA can be selected such that RB/RA = 9. This would convert a CF’ of 5 pF into a CF of 0.5 pF. This relationship holds as long as RA << RF FIGURE 14. Input Common-Mode Capacitance vs. VCM 17 www.national.com LM6211 Typical Applications LM6211 Typical Applications (Continued) 20120346 FIGURE 16. Method for CF selection SENSOR INTERFACES The low input bias current and low input referred noise of the LM6211 make it ideal for sensor interfaces. These circuits are required to sense voltages of the order of a few µV, and currents amounting to less than a nA, and hence the op amp needs to have low voltage noise and low input bias current. Typical applications include infra-red (IR) thermometry, thermocouple amplifiers and pH electrode buffers. Figure 17 is an example of a typical circuit used for measuring IR radiation intensity, often used for estimating the temperature of an object from a distance. The IR sensor generates a voltage proportional to I, which is the intensity of the IR radiation falling on it. As shown in Figure 17, K is the constant of proportionality relating the voltage across the IR sensor (VIN) to the radiation intensity, I. The resistances RA and RB are selected to provide a high gain to amplify this voltage, while CF is added to filter out the high frequency noise. www.national.com 20120348 FIGURE 17. IR Radiation Sensor 18 inches (millimeters) unless otherwise noted 5-Pin SOT23 NS Package Number MF05A National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. 2. 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National Semiconductor Americas Customer Support Center Email: [email protected] Tel: 1-800-272-9959 www.national.com National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530 85 86 Email: [email protected] Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +44 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: [email protected] National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: [email protected] Tel: 81-3-5639-7560 LM6211 Low Noise, RRO Op-Amp with CMOS Input and 24V Operation Physical Dimensions