TPS61086 Actual Size 3 mm x 3 mm www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 18.5 V PFM/PWM STEP-UP DC-DC CONVERTER WITH 2.0 A SWITCH FEATURES 1 • • • • • • • • APPLICATIONS 2.3 V to 6.0 V Input Voltage Range 18.5 V Boost Converter With 2.0 A Switch Current 1.2 MHz Switching Frequency Power Save Mode for improved Efficiency at Low Output Power or Forced PWM Adjustable Soft-Start Thermal Shutdown Undervoltage Lockout 10-Pin QFN Package • • • • • • • Handheld Devices GPS Receiver Digital Still Camera Portable Applications DSL Modem PCMCIA Card TFT LCD Bias Supply DESCRIPTION The TPS61086 is a high frequency, high efficiency DC to DC converter with an integrated 2.0 A, 0.13 Ω power switch capable of providing an output voltage up to 18.5 V. The implemented boost converter is based on a fixed frequency of 1.2MHz, pulse-width-modulation (PWM) controller that allows the use of small external inductors and capacitors and provides fast transient response. At light load, the device can operate in Power Save Mode with pulse-frequency-modulation (PFM) to improve the efficiency while keeping a low output voltage ripple. For very noise sensitive applications, the device can be forced to PWM Mode operation over the entire load range by pulling the MODE pin high. The external compensation allows optimizing the application for specific conditions. A capacitor connected to the soft-start pin minimizes inrush current at startup. L 3.3 mH VIN 2.5 V to 6 V Cin 10 mF 16 V 8 Cby 1 mF 16 V 3 9 4 5 IN SW EN SW MODE FB AGND COMP PGND SS TPS61086 6 D PMEG2010AEH VS 12 V R1 156 kW 7 Cout 2* 10 mF 25 V 2 R2 18 kW 1 Rcomp 68 kW 10 Css 100 nF Ccomp 820 pF 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2009, Texas Instruments Incorporated TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. ORDERING INFORMATION (1) (2) (1) (2) TA ORDERING PACKAGE PACKAGE MARKING –40 to 85°C TPS61086DRC QFN-10 (DRC) PSRI The DRC package is available taped and reeled. For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) Input voltage range IN (2) VALUE UNIT –0.3 to 7.0 V Voltage range on pins EN, FB, SS, FREQ, COMP –0.3 to 7.0 V Voltage on pin SW –0.3 to 20 V ESD rating HBM 2 kV ESD rating MM 200 V ESD rating CDM 500 V Continuous power dissipation See Dissipation Rating Table Operating junction temperature range –40 to 150 °C Storage temperature range –65 to 150 °C (1) (2) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability All voltage values are with respect to network ground terminal. DISSIPATION RATINGS (1) (2) (1) (2) PACKAGE RθJA TA ≤ 25°C POWER RATING TA = 70°C POWER RATING TA = 85°C POWER RATING QFN 40°C/W 3.3 W 1.8 W 1.3 W PD = (TJ – TA)/RθJA. The exposed thermal die is soldered to the PCB using thermal vias. For more information, please refer to the Texas Instruments Application report SLMA002 regarding thermal characteristics of the PowerPAD package. RECOMMENDED OPERATING CONDITIONS MIN VIN Input voltage range VS Boost output voltage range TA Operating free-air temperature TJ Operating junction temperature 2 Submit Documentation Feedback TYP MAX UNIT 2.3 6.0 V VIN + 0.5 18.5 V –40 85 °C –40 125 °C Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 ELECTRICAL CHARACTERISTICS VIN = 3.3 V, EN = IN, VS = 12 V, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SUPPLY VIN Input voltage range 2.3 IQ Operating quiescent current into IN Device not switching, VFB = 1.3 V ISDVIN Shutdown current into IN EN = GND VUVLO Under-voltage lockout threshold VIN falling TSD Thermal shutdown TSDHYS Thermal shutdown hysteresis 75 VIN rising 6.0 V 100 µA 1 µA 2.2 V 2.3 Temperature rising V 150 °C 14 °C LOGIC SIGNALS EN, FREQ VIH High level input voltage VIN = 2.3 V to 6.0 V VIL Low level input voltage VIN = 2.3 V to 6.0 V 2 0.5 V V IINLEAK Input leakage current EN = GND 0.1 µA 18.5 V BOOST CONVERTER VS Boost output voltage VIN + 0.5 VFB Feedback regulation voltage 1.230 gm Transconductance error amplifier IFB Feedback input bias current VFB = 1.238 V 0.1 µA rDS(on) N-channel MOSFET on-resistance VIN = VGS = 5 V, ISW = current limit 0.13 0.20 Ω VIN = VGS = 3.3 V, ISW = current limit 0.16 0.23 ISWLEAK SW leakage current 10 µA ILIM N-Channel MOSFET current limit 2.6 3.2 A ISS Soft-start current fS Oscillator frequency 1.238 1.246 EN = GND, VSW = 6.0V 2.0 VSS = 1.238 V Line regulation VIN = 2.3 V to 6.0 V, IOUT = 10 mA Load regulation VIN = 3.3 V, IOUT = 1 mA to 400 mA 7 10 13 µA 0.9 1.2 1.5 MHz 0.0002 %/V 0.11 %/A Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 V µA/V 107 3 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com PIN ASSIGNMENT DRC PACKAGE (TOP VIEW) COMP SS FB EN MODE Thermal Pad IN AGND SW PGND SW 10-PIN 3mm x 3mm x 1mm QFN TERMINAL FUNCTIONS TERMINAL NAME NO. I/O DESCRIPTION COMP 1 I/O FB 2 I Feedback pin EN 3 I Shutdown control input. Connect this pin to logic high level to enable the device AGND PGND SW Compensation pin 4, Thermal Pad Analog ground 5 Power ground 6, 7 IN 8 MODE 9 SS 10 Switch pin Input supply pin I Operating mode selection pin. MODE = 'high' for forced PWM operation. MODE = 'low' for PFM operation Soft-start control pin. Connect a capacitor to this pin if soft-start needed. Open = no soft-start TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE η Efficiency vs Load current- PFM VIN = 3.3 V, VS = 9 V, 12 V, 15 V Figure 1 η Efficiencyvs Load current - Forced PWM VIN = 3.3 V, VS = 9 V, 12 V, 15 V Figure 2 PFM switching 1 - discontinuous conduction VIN = 3.3 V, VS = 12 V, Iout = 50 mA Figure 3 PFM switching 1 - discontinuous conduction VIN = 3.3 V, VS = 12 V, Iout = 50 mA Figure 4 PFM switching - discontinuous conduction VIN = 3.3 V, VS = 12 V, Iout = 4 mA Figure 5 Forced PWM switching - discontinuous conduction VIN = 3.3 V, VS = 12 V, Iout = 4 mA Figure 6 Iout(max) PFM / PWM switching - continuous conduction VIN = 3.3 V, VS = 12 V, Iout = 300 mA Figure 7 Maximum output current Figure 8 Load transient response - PFM VIN = 3.3 V, VS = 12 V, Iout = 50 mA...150 mA Figure 9 Load transient response - Forced PWM VIN = 3.3 V, VS = 1 2V, Iout = 50 mA...150 mA Figure 10 Line transient response - PFM VIN = 2.3 V...6.0 V, VS = 12 V, Iout = 0 mA Figure 11 Line transient response - Forced PWM VIN = 2.3 V...6.0 V, VS = 12 V, Iout = 150 mA Figure 12 fS Switching frequency - Forced PWM vs Load current, VIN = 3.3 V, VS = 12 V Figure 13 fS Switching frequency - Forced PWM vs Supply voltage, VS = 12 V, Iout = 200 mA Figure 14 Soft-start Supply current 4 Figure 15 vs Supply voltage,VIN = 3.3 V, VS = 12 V Submit Documentation Feedback Figure 16 Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 The typical characteristics are measured with the inductor CDRH6D12 3.3 µH from Sumida and the rectifier diode SL22. PFM MODE EFFICIENCY vs OUTPUT CURRENT 100 FORCE PWM MODE EFFICIENCY vs OUTPUT CURRENT 100 VS = 9 V 90 80 80 VS = 15 V 70 VS = 12 V Efficiency - % Efficiency - % 70 60 50 40 30 VS = 15 V VS = 12 V 60 50 40 30 FREQ = GND VIN = 3.3 V L = 3.3 µH 20 10 0 VS = 9 V 90 0.1 1 10 100 IO - Output Current - mA FREQ = VIN VIN = 3.3 V L = 3.3 µH 20 10 1000 0 1 10 100 IO - Output Current - mA Figure 1. Figure 2. PFM MODE SWITCHING PULSE PFM MODE SWITCHING PULSES VSW 5 V/div VSW 5 V/div VS_AC 50 mV/div VS_AC 50 mV/div Il 0.5 A/div Il 0.5 A/div 1000 VIN = 3.3 V, VS = 12 V/50 mA VIN = 3.3 V, VS = 12 V/50 mA 10 µs/div 10 µs/div Figure 3. Figure 4. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 5 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com PFM MODE - LIGHT LOAD FORCED PWM MODE - LIGHT LOAD VSW 10 V/div VSW 10 V/div VS_AC 50 mV/div VS_AC 50 mV/div Il 0.5 A/div Il 0.5 A/div VIN = 3.3 V, VS = 12 V/4 mA VIN = 3.3 V, VS = 12 V/4 mA 100 µs/div 100 µs/div Figure 5. Figure 6. FORCED PWM / PFM MODE - HEAVY LOAD OUTPUT CURRENT vs SUPPLY VOLTAGE 1.6 VSW 10 V/div VS = 9 V IO - Output current - A 1.4 VS_AC 50 mV/div Il 1 A/div VIN = 3.3 V VS = 12 V/300 mA 1.2 VS = 12 V 1.0 0.8 0.6 VS = 15 V 0.4 VS = 18.5 V 0.2 L = 3.3 µH 0.0 2.5 400 ns/div Figure 7. 6 3.0 3.5 4.0 4.5 5.0 5.5 VIN - Supply voltage - V 6.0 Figure 8. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 LOAD TRANSIENT RESPONSE PFM MODE VS_AC 50 mV/div LOAD TRANSIENT RESPONSE FORCE PWM MODE VS_AC 100 mV/div Rcomp = 16 kΩ Ccomp = 2.7 nF COUT = 40 µF L = 3.3 µH IOUT 50 mA/div Rcomp = 16 kΩ Ccomp = 2.7 nF COUT = 40 µF L = 3.3 µH IOUT 50 mA/div VIN = 3.3 V VS = 12 V/50 mA - 150 mA VIN = 3.3 V VS = 12 V/50 mA - 150 mA 400 µs/div 400 µs/div Figure 9. Figure 10. LINE TRANSIENT RESPONSE LIGHT LOAD LINE TRANSIENT RESPONSE HEAVY LOAD VIN 2 V/div VIN 2 V/div COUT = 40 µF L = 3.3 µH Rcomp = 16 kΩ Ccomp = 2.7 nF VS_AC 200 mV/div COUT = 40 µF L = 3.3 µH Rcomp = 16 kΩ Ccomp = 2.7 nF VS_AC 200 mV/div VIN = 2.3 V - 6.0V VS = 12 V/0 mA VIN = 2.3 V - 6.0V VS = 12 V/150 mA 400 µs/div 400 µs/div Figure 11. Figure 12. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 7 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com FREQUENCY vs LOAD CURRENT FREQUENCY vs SUPPLY VOLTAGE 1600 1400 MODE = VIN Forced PWM L = 3.3 µH 1400 1200 MODE = VIN Forced PWM L = 3.3 µH 1000 f - Frequency - kHz f - Frequency - kHz 1200 1000 800 600 400 VIN = 3.3 V VS = 12 V 200 800 600 400 200 VS = 12 V / 200 mA 0 0 0.1 0.2 0.3 0.4 0.5 0.6 2.5 IO - Load current - mA Figure 13. Figure 14. SOFT-START SUPPLY CURRENT vs SUPPLY VOLTAGE 5.5 6.0 2.5 VIN = 3.3 V VS = 12 V / 300 mA ICC - Supply Current - mA EN 5 V/div VS 5 V/div 2.0 MODE = VIN Forced PWM 1.5 MODE = GND (PFM) 1.0 0.5 IL 1 A/div VIN = 3.3 V VS = 12 V/50 mA CSS = 100 nF 0 2.0 2 ms/div Figure 15. 8 3.0 3.5 4.0 4.5 5.0 VCC - Supply Voltage - V 2.5 3.0 3.5 4.0 4.5 5.0 5.5 VCC - Supply Voltage - V 6.0 Figure 16. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 DETAILED DESCRIPTION VIN VS EN SS IN SW MODE SW Current limit and Soft Start Toff Generator AGND Bias Vref = 1.24 V UVLO Thermal Shutdown Ton PWM Generator Gate Driver of Power Transistor COMP GM Amplifier FB Vref PGND Figure 17. Block Diagram The boost converter is designed for output voltages up to 18.5 V with a switch peak current limit of 2.0 A minimum. The device, which operates in a current mode scheme with quasi-constant frequency, is externally compensated for maximum flexibility and stability. The switching frequency is fixed to 1.2 MHz and the minimum input voltage is 2.3 V. To limit the inrush current at start-up a soft-start pin is available. TPS61086 boost converter’s novel topology using adaptive off-time provides superior load and line transient responses and operates also over a wider range of applications than conventional converters. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 9 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com Design Procedure The first step in the design procedure is to verify that the maximum possible output current of the boost converter supports the specific application requirements. A simple approach is to estimate the converter efficiency, by taking the efficiency numbers from the provided efficiency curves or to use a worst case assumption for the expected efficiency, e.g. 90%. 1. Duty cycle, D: D = 1- VIN ×h VS (1) 2. Maximum output current, Iout(max): DI ö æ I out (max) = ç I LIM (min) - L ÷ × (1 - D ) 2 ø è (2) 3. Peak switch current in application, Iswpeak: I swpeak = I DI L + out 2 1- D (3) with the inductor peak-to-peak ripple current, ΔIL DI L = VIN × D fS × L (4) and VIN Minimum input voltage VS Output voltage ILIM(min) Converter switch current limit (minimum switch current limit = 2.0 A) fS Converter switching frequency (typically 1.2 MHz) L Selected inductor value η Estimated converter efficiency (please use the number from the efficiency plots or 90% as an estimation) The peak switch current is the steady state peak switch current that the integrated switch, inductor and external Schottky diode has to be able to handle. The calculation must be done for the minimum input voltage where the peak switch current is the highest. Soft-start The boost converter has an adjustable soft-start to prevent high inrush current during start-up. To minimize the inrush current during start-up an external capacitor, connected to the soft-start pin SS and charged with a constant current, is used to slowly ramp up the internal current limit of the boost converter. When the EN pin is pulled high, the soft-start capacitor CSS is immediately charged to 0.3 V. The capacitor is then charged at a constant current of 10 µA typically until the output of the boost converter VS has reached its Power Good threshold (90% of VS nominal value). During this time, the SS voltage directly controls the peak inductor current, starting with 0 A at VSS = 0.3 V up to the full current limit at VSS ≈ 800 mV. The maximum load current is available after the soft-start is completed. The larger the capacitor the slower the ramp of the current limit and the longer the soft-start time. A 100 nF capacitor is usually sufficient for most of the applications. When the EN pin is pulled low, the soft-start capacitor is discharged to ground. Inductor Selection The TPS61086 is designed to work with a wide range of inductors. The main parameter for the inductor selection is the saturation current of the inductor which should be higher than the peak switch current as calculated in the Design Procedure section with additional margin to cover for heavy load transients. An alternative, more conservative, is to choose an inductor with a saturation current at least as high as the maximum switch current limit of 3.2 A. The other important parameter is the inductor DC resistance. Usually the lower the DC resistance the higher the efficiency. It is important to note that the inductor DC resistance is not the only parameter 10 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 determining the efficiency. Especially for a boost converter where the inductor is the energy storage element, the type and core material of the inductor influences the efficiency as well. Usually an inductor with a larger form factor gives higher efficiency. The efficiency difference between different inductors can vary between 2% to 10%. For the TPS61086, inductor values between 3 µH and 6 µH are a good choice. Possible inductors are shown in Table 1. Typically, it is recommended that the inductor current ripple is below 35% of the average inductor current. The following equation can therefore be used to calculate the inductor value, L: 2 æ V ö æ V -V ö æ h ö L = ç IN ÷ × ç S IN ÷ × ç ÷ è VS ø è I out × f S ø è 0.35 ø (5) with VIN Minimum input voltage VS Output voltage Iout Maximum output current in the application fS Converter switching frequency (typically 1.2 MHz) η Estimated converter efficiency (please use the number from the efficiency plots or 90% as an estimation) Table 1. Inductor Selection L (µH) SUPPLIER COMPONENT CODE SIZE (L×W×H mm) DCR TYP (mΩ) Isat (A) 3.3 Sumida CDH38D09 4x4x1 240 1.25 4.7 Sumida CDPH36D13 5 x 5 x 1.5 155 1.36 3.3 Sumida CDPH4D19F 5.2 x 5.2 x 2 33 1.5 3.3 Sumida CDRH6D12 6.7 x 6.7 x 1.5 62 2.2 4.7 Würth Elektronik 7447785004 5.9 x 6.2 x 3.3 60 2.5 5 Coilcraft MSS7341 7.3 x 7.3 x 4.1 24 2.9 Rectifier Diode Selection To achieve high efficiency a Schottky type should be used for the rectifier diode. The reverse voltage rating should be higher than the maximum output voltage of the converter. The averaged rectified forward current Iavg, the Schottky diode needs to be rated for, is equal to the output current Iout: I avg = I out (6) Usually a Schottky diode with 1A maximum average rectified forward current rating is sufficient for most applications. The Schottky rectifier can be selected with lower forward current capability depending on the output current Iout but has to be able to dissipate the power. The dissipated power, PD, is the average rectified forward current times the diode forward voltage, Vforward. PD = I avg × V forward (7) Typically the diode should be able to dissipate around 500mW depending on the load current and forward voltage. Table 2. Rectifier Diode Selection CURRENT RATING Iavg Vr Vforward/Iavg SUPPLIER COMPONENT CODE PACKAGE TYPE 750 mA 20 V 0.425 V / 1 A Fairchild Semiconductor FYV0704S SOT 23 1A 20 V 0.39 V / 1 A NXP PMEG2010AEH SOD 123 1A 20 V 0.5 V / 1 A Vishay Semiconductor SS12 SMA 1A 20 V 0.44 V / 1 A Vishay Semiconductor MSS1P2L µ -SMP 2A 20 V 0.44 V / 2 A Vishay Semiconductor SL22 SMB Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 11 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com Setting the Output Voltage The output voltage is set by an external resistor divider. Typically, a minimum current of 50 µA flowing through the feedback divider gives good accuracy and noise covering. A standard low side resistor of 18 kΩ is typically selected. The resistors are then calculated as: VS R2 = VFB » 18k W 70 m A æ V ö R1 = R 2 × ç S - 1÷ è VFB ø R1 VFB VFB = 1.238V R2 (8) Compensation (COMP) The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The COMP pin is the output of the internal transconductance error amplifier. Standard values of RCOMP = 16 kΩ and CCOMP = 2.7 nF will work for the majority of the applications. Please refer to Table 3 for dedicated compensation networks giving an improved load transient response. The following equations can be used to calculate RCOMP and CCOMP: RCOMP = 110 × VIN × VS × Cout L × I out CCOMP = Vs × Cout 7.5 × I out × RCOMP (9) with VIN Minimum input voltage VS Output voltage Cout Output capacitance L Inductor value, e.g. 3.3 µH or 4.7 µH Iout Maximum output current in the application Make sure that RCOMP < 120 kΩ and CCOMP> 820 pF, independent of the results of the above formulas. Table 3. Recommended Compensation Network Values at High/Low Frequency L VS 15 V 3.3 µH 12 V 9V VIN ± 20% RCOMP CCOMP 820 pF 5V 100 kΩ 3.3 V 91 kΩ 1.2 nF 5V 68 kΩ 820 pF 3.3 V 68 kΩ 1.2 nF 5V 39 kΩ 820 pF 3.3 V 39 kΩ 1.2 nF Table 3 gives conservative RCOMP and CCOMP values for certain inductors, input and output voltages providing a very stable system. For a faster response time, a higher RCOMP value can be used to enlarge the bandwidth, as well as a slightly lower value of CCOMP to keep enough phase margin. These adjustments should be performed in parallel with the load transient response monitoring of TPS61086. Input Capacitor Selection For good input voltage filtering low ESR ceramic capacitors are recommended. TPS61086 has an analog input IN. Therefore, a 1 µF bypass is highly recommended as close as possible to the IC from IN to GND. One 10 µF ceramic input capacitors are sufficient for most of the applications. For better input voltage filtering this value can be increased. Refer to Table 4 and typical applications for input capacitor recommendation 12 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 Output Capacitor Selection For best output voltage filtering a low ESR output capacitor like ceramic capcaitor is recommended. Two to four 10 µF ceramic output capacitors (or two 22 µF) work for most of the applications. Higher capacitor values can be used to improve the load transient response. Refer to Table 4 for the selection of the output capacitor. Table 4. Rectifier Input and Output Capacitor Selection CAPACITOR/SI ZE VOLTAGE RATING SUPPLIER COMPONENT CODE CIN 22 µF/1206 16 V Taiyo Yuden EMK316 BJ 226ML IN bypass 1 µF/0603 16 V Taiyo Yuden EMK107 BJ 105KA COUT 10 µF/1206 25 V Taiyo Yuden TMK316 BJ 106KL To calculate the output voltage ripple, the following equation can be used: DVC = VS - VIN I out × VS × f S Cout DVC _ ESR = I L ( peak ) × RC _ ESR (10) with ΔVC Output voltage ripple dependent on output capacitance,output current and switching frequency VS Output voltage VIN Minimum input voltage of boost converter fS Converter switching frequency (typically 1.2 MHz) Iout Output capacitance ΔVC_ESR Output voltage ripple due to output capacitors ESR (equivalent series resistance) ISWPEAK Inductor peak switch current in the application RC_ESR Output capacitors equivalent series resistance (ESR) ΔVC_ESR can be neglected in many cases since ceramic capacitors provide very low ESR. Operating Mode (MODE) Power Save Mode Connecting the MODE pin to GND (or any low logic level) enables the Power Save Mode operation. The converter operates in quasi fixed frequency PWM (Pulse Width Modulation) mode at moderate to heavy load and in the PFM (Pulse Frequency Modulation) mode during light loads, which maintains high efficiency over a wide load current range. In PFM mode the converter is skipping switch pulses. However, within a PFM pulse, the switching frequency is still fixed to 1.2 MHz typically and the duty cycle determined by the input and output voltage. Therefore, the inductor peak current will remain constant for a defined application. With an increasing output load current, the PFM pulses become closer and closer (the PFM mode frequency gets higher) until no pulse is skipped anymore: the device operates then in CCM (Continuous Conduction Mode) with normal PWM mode. The PFM mode frequency (between each PFM pulse) depends on the load current, the external components like the inductor or the output capacitor values as well as the output voltage. The device enters Power Save Mode as the inductor peak current falls below a 0.6A typically and switches until VS is 1% higher than its nominal value. The converter stops switching when VS = VS + 0.5%. The output voltage will thenrefore oscillate between 0.5% and 1% more than its nominal value which will provide excellent transient response to sudden load change, since the output voltage drop will be reduced due to this slight positive offset (see Figure 9). Forced PWM Mode Pulling the MODE pin high forces the converter to operate in a continuous PWM mode evan at light load currents. The advantage is that the converter operates with a quai constant frequency that allows simple filtering of the swithcing frequency for noise-sensitive applications. In this mode and at light load, the efficiency is lower compared to the Power Save Mode. Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 13 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com For additional flexibility, it is possible to switch from Power Save Mode to Forced PWM Mode during operation. This allows efficient power management by adjusting the operation of the converter to the specific system requirements. Undervoltage Lockout (UVLO) To avoid mis-operation of the device at low input voltages an undervoltage lockout is included that disables the device, if the input voltage falls below 2.2 V. Thermal Shutdown A thermal shutdown is implemented to prevent damages due to excessive heat and power dissipation. Typically the thermal shutdown happens at a junction temperature of 150°C. When the thermal shutdown is triggered the device stops switching until the junction temperature falls below typically 136°C. Then the device starts switching again. Overvoltage Prevention If overvoltage is detected on the FB pin (typically 3 % above the nominal value of 1.238 V) the part stops switching immediately until the voltage on this pin drops to its nominal value. This prevents overvoltage on the output and secures the circuits connected to the output from excessive overvoltage. 14 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 APPLICATION INFORMATION L 3.3 µH VIN 3.3 V ± 20% 8 Cin 10 µF 16 V Cby 1 µF 16 V 3 9 4 5 IN SW EN SW MODE FB AGND COMP PGND SS VS 12 V/50 mA D PMEG2010AEH 6 R1 156 kΩ 7 Cout 2*10 µF 25 V 2 R2 18 kΩ 1 Rcomp 68kΩ 10 Css 100 nF TPS61086 Ccomp 1.2 nF Figure 18. Typical Application, 3.3 V to 12 V (PFM MODE) L 3.3 µH VIN 3.3 V ± 20% 8 Cin 10 µF 16 V Cby 1 µF 16 V 3 9 4 5 IN SW EN SW MODE FB AGND COMP PGND SS TPS61086 VS 12 V/500 mA max. D PMEG2010AEH 6 R1 156 kΩ 7 Cout 10 µF 25 V 2 R2 18 kΩ 1 Rcomp 68kΩ 10 Css 100 nF Ccomp 1.2 nF Figure 19. Typical Application, 3.3V to 12 V (FORCE PWM MODE) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 15 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com Riso 10 kW L 3.3 µH VIN 5 V ± 20% Cby 1 µF/16 V 8 Cin 2* 10 µF/ 16 V 3 9 4 Enable SW IN SW EN MODE FB AGND COMP PGND SS 5 VS 15 V/50 mA BC857C D PMEG2010AEH 6 Ciso 1 µF/ 25 V 7 R1 200 kΩ 2 Cout 4*10 µF/ 25 V R2 18 kΩ 1 Rcomp 100 kΩ 10 TPS61086 Css 100 nF Ccomp 820 pF Figure 20. Typical Application with External Load Disconnect Switch L 3.3 µH Overvoltage D PMEG2010AEH Protection VIN 3.3 V ± 20% 8 Cin 10 µF 16 V Cby 1 µF 16 V 3 9 4 5 IN SW EN SW MODE FB COMP AGND PGND SS TPS61086 VS 15 V/30 mA 6 Dz BZX84C 18V 7 R1 200 kΩ Cout 10 µF 25 V 2 Rlimit 110 Ω 1 R2 18 kΩ Rcomp 91 kΩ 10 Css 100 nF Ccomp 1.2 nF Figure 21. Typical Application, 3.3 V to 15 V (PFM MODE) with Overvoltage Protection 16 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 TFT LCD APPLICATION T2 BC850B 3·Vs VGL -7 V/20 mA T1 BC857B -Vs R8 6.8 kΩ C13 1 µF/ 35 V C16 470 nF/ 50 V C14 470 nF/ 25 V D4 BAV99 C15 470 nF/ 50 V D3 BAV99 C18 470 nF/ 50 V R10 13 kΩ 2·Vs C17 470 nF/ 50 V D2 BAV99 D8 BZX84C7V5 Vgh 26.5 V/20 mA C20 1 µF/ 35 V C19 470 nF/ 50 V D9 BZX84C27V L 3.3 µH VIN 5 V ± 20% Cin 2*10 µF/ 16 V Cby 1 µF/ 16 V D SL22 8 IN SW EN SW 3 7 9 R1 200 kΩ Cout 4*10µF/ 25V 2 MODE FB 4 5 VS 15 V/500 mA 6 R2 18 kΩ 1 AGND COMP PGND SS TPS61086 Rcomp 100 kΩ 10 Css 100 nF Ccomp 820 pF Figure 22. Typical Application 5 V to 15 V (FORCE PWM MODE) for TFT LCD with External Charge Pumps (VGH, VGL) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 17 TPS61086 SLVSA05 – AUGUST 2009 ................................................................................................................................................................................................ www.ti.com WHITE LED APPLICATIONS L 3.3 µH optional Cby 1 µF/ 16 V VIN 3.3 V ± 20% 6 8 Cin 10 µF/ 16 V 3 9 4 5 IN SW EN SW D PMEG2010AEH Dz BZX84C 18 V VS 300 mA 3S3P wLED LW E67C 7 Cout 2* 10 µF/ 25 V 2 MODE FB AGND COMP PGND SS Rlimit 110 Ω 1 Rcomp 68 kΩ 10 TPS61086 Css 100 nF Rsense 15 Ω Ccomp 1.2 nF Figure 23. Simple Application (3.3 V input voltage - FORCED PWM MODE) for wLED Supply (3S3P) (with optional clamping Zener diode) L 3.3 µH optional VIN 3.3 V ± 20% Cin 10 µF/ 16 V Cby 1 µF/ 16 V 3 9 4 PWM 100 Hz to 500 Hz 6 8 5 IN SW EN SW D PMEG2010AEH Dz BZX84C 18 V VS 300 mA 3S3P wLED LW E67C 7 Cout 2* 10 µF/ 25 V 2 MODE FB AGND COMP PGND SS TPS61086 Rlimit 110 Ω 1 Rcomp 68 kΩ 10 Css 100 nF Rsense 15 Ω Ccomp 1.2 nF Figure 24. Simple Application (3.3 V input voltage - FORCED PWM MODE) for wLED Supply (3S3P) with Adjustable Brightness Control using a PWM Signal on the Enable Pin (with optional clamping Zener diode) 18 Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 TPS61086 www.ti.com ................................................................................................................................................................................................ SLVSA05 – AUGUST 2009 L 3.3 µH optional VIN 3.3 V ± 20% Cby 1 µF/ 16 V 6 8 D PMEG2010AEH VS 300 mA 3S3P wLED LW E67C Dz BZX84C SW IN 18 V Cin 10 µF/ 16 V 3 9 4 5 7 EN Cout 2* 10 µF/ 25 V SW 2 MODE FB AGND COMP PGND SS TPS61086 R1 180 kΩ Rlimit 110 Ω 1 10 Css 100 nF Rcomp 68 kΩ Ccomp 1.2 nF R2 127 kΩ Rsense 15 Ω Analog Brightness Control 3.3 V ~ wLED off 0 V ~ lLED = 30 mA (each string) PWM Signal Can be used swinging from 0 V to 3.3 V Figure 25. Simple Application (3.3 V input voltage - FORCED PWM MODE) for wLED Supply (3S3P) with Adjustable Brightness Control using an Analog Signal on the Feedback Pin (with optional clamping Zener diode) Submit Documentation Feedback Copyright © 2009, Texas Instruments Incorporated Product Folder Link(s): TPS61086 19 PACKAGE OPTION ADDENDUM www.ti.com 7-Sep-2009 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty TPS61086DRCR ACTIVE SON DRC 10 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS61086DRCT ACTIVE SON DRC 10 250 CU NIPDAU Level-2-260C-1 YEAR Green (RoHS & no Sb/Br) Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 12-Sep-2009 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant TPS61086DRCR SON DRC 10 3000 330.0 12.4 3.3 3.3 1.1 8.0 12.0 Q2 TPS61086DRCT SON DRC 10 250 180.0 12.4 3.3 3.3 1.1 8.0 12.0 Q2 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 12-Sep-2009 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) TPS61086DRCR SON DRC 10 3000 346.0 346.0 29.0 TPS61086DRCT SON DRC 10 250 190.5 212.7 31.8 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. 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