SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 FEATURES D Operating Input Voltage 2.25 V to 5.5 V D Output Voltage as Low as 0.7 V D 1% Internal 0.7 V Reference D Predictive Gate Drivet N-Channel MOSFET APPLICATIONS D Networking Equipment D Telecom Equipment D Base Stations D Servers D DSP Power D Power Modules Drivers for Higher Efficiency D Externally Adjustable Soft-Start and D D D D D Overcurrent Limit Fixed-Frequency Voltage-Mode Control − TPS40007, 300 kHz − TPS40009, 600 kHz Source/Sink with VOUT Prebias 10-Lead MSOP PowerPadt Package for Higher Performance Thermal Shutdown Internal Boostrap Diode DESCRIPTION The TPS4000x are controllers for low-voltage, non-isolated synchronous buck regulators. These controllers drive an N-channel MOSFET for the primary buck switch, and an N-channel MOSFET for the synchronous rectifier switch, thereby achieving very high-efficiency power conversion. In addition, the device controls the delays from main switch off to rectifier turn-on and from rectifier turn-off to main switch turn-on in such a way as to minimize diode losses (both conduction and recovery) in the synchronous rectifier with TI’s proprietary Predictive Gate Drivet technology. The reduction in these losses is significant and increases efficiency. For a given converter power level, smaller FETs can be used, or heat sinking can be reduced or even eliminated. SIMPLIFIED APPLICATION DIAGRAM VIN TPS40007 TPS40009 1 ILIM BOOT 10 2 FB HDRV 9 3 COMP SW 8 VDD 7 LDRV 6 4 5 VOUT SS/SD GND UDG−03161 PowerPADt and Predictive Gate Drivet are trademarks of Texas Instruments Incorporated. !" # $%&" !# '%()$!" *!"&+ *%$"# $ " #'&$$!"# '& ",& "&# &-!# #"%&"# #"!*!* .!!"/+ *%$" '$&##0 *&# " &$&##!)/ $)%*& "&#"0 !)) '!!&"&#+ Copyright 2003, 2004 Texas Instruments Incorporated www.ti.com 1 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 DESCRIPTION (continued) The current-limit threshold is adjustable with a single resistor connected to the device. The TPS4000x controllers implement a closed-loop soft start function. Startup ramp time is set by a single external capacitor connected to the SS/SD pin. The SS/SD pin is also used for shutdown. ORDERING INFORMATION TA FREQUENCY PACKAGED DEVICES MSOP(1) (DGQ) 300 kHz TPS40007DGQ 600 kHz TPS40009DGQ −40°C to 85°C (1) The DGQ package is available taped and reeled. Add R suffix to device type (e.g. TPS40007DGQR) to order quantities of 2,500 devices per reel and 80 units per tube. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range unless otherwise noted(2) TPS4000x BOOT Input voltage range, VIN COMP, FB, ILIM, SS/SD VSW + 6.5 −0.3 to 6 SW −3 to 10.5 SWT (SW transient < 50 ns) VDD UNIT V −5 6 Operating junction temperature range, TJ −40 to 150 Storage temperature, Tstg −55 to 150 °C C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260 (2) Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DGQ PACKAGE(3)(4) (TOP VIEW) ILIM FB COMP SS/SD GND 1 10 2 9 3 8 4 7 5 6 ACTUAL SIZE 3,05mm x 4,98mm (3) (4) 2 See technical brief SLMA002 for PCB guidelines for PowerPAD packages. PowerPADt heat slug should be connected to GND (pin 5). www.ti.com BOOT HDRV SW VDD LDRV SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 ELECTRICAL CHARACTERISTICS temperature range, TA = −40_C to 85_C, VDD = 5.0 V, TA = TJ; all parameters measured at zero power dissipation (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT INPUT SUPPLY VDD VHGATE IDD UVLO Input voltage range 2.25 High-side gate voltage 5.5 6 V Shutdown current VBOOT − VSW SS/SD = 0 V, 0.25 0.45 Quiescent current FB = 0.8 V 1.4 2.0 Switching current No load at HDRV/LDRV 1.5 4.0 1.95 2.05 2.15 V 80 150 220 mV 250 300 350 500 600 700 0.80 0.93 1.07 0.24 0.31 0.44 87.0% 94.0% 83.0% 93.0% Outputs off Minimum on-voltage Hysteresis mA OSCILLATOR TPS40007 fOSC Oscillator frequency VRAMP Ramp voltage TPS40009 2.25 V ≤ VDD ≤ 5.00 V VPEAK − VVALLEY Ramp valley voltage kHz V PWM Maximum duty cycle(2) TPS40007 TPS40009 FB = 0 V, VDD = 3.3 V Minimum duty cycle 0% Minimum controllable pulse width(1)(3) 100 150 ns ERROR AMPLIFIER Line, Temperature 0.690 0.700 0.711 0.693 0.700 0.707 30 130 VFB FB input voltage IFB VOH FB input bias current High-level output voltage FB = 0 V, VOL IOH Low-level output voltage FB =VDD, IOH = 1.0 mA IOL = 0.5 mA Output source current COMP = 0.7 V, FB = GND 2 6 IOL GBW Output sink current Gain bandwidth(1) COMP = 0.7 V, FB = VDD 3 8 5 10 MHz 55 85 dB TA = 25°C AOL Open loop gain SHORT CIRCUIT CURRENT PROTECTION ISINK ISINK ILIM sink current VOS VILIM tON Minimum HDRV pulse time in overcurrent 2.0 V nA 2.5 0.08 0.15 V mA µA 11 15 19 ILIM sink current VDD = 5 V VDD = 2.25 V 9.5 13.0 16.5 µA Offset voltage SW vs ILIM(1) 2.25 V ≤ VDD ≤ 5.00 −20 0 20 mV Input voltage range 2 VDD = 3.3 V SW leading edge blanking pulse in overcurrent detection(1) 220 100 tSS Soft-start capacitor cycles as fault timer(1) (1) Ensured by design. Not production tested. (2) Derate the maximum duty cycle by 3% for VDD < 3 V (3) Operating at PWM on-times of less than 100 ns could lead to overlap between HDRV and LDRV pulses. www.ti.com VDD V 330 ns ns 6 3 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 ELECTRICAL CHARACTERISTICS temperature range, TA = −40_C to 85_C, VDD = 5.0 V, TA = TJ; all parameters measured at zero power dissipation (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT OUTPUT DRIVER RHDHI HDRV pull-up resistance VBOOT−VSW = 3.3 V, ISOURCE = −100 mA 3 5.5 RHDLO HDRV pull-down resistance VBOOT − VSW = 3.3 V, ISINK = 100 mA 1.5 3 RLDHI LDRV pull-up resistance RLDLO LDRV pull-down resistance tRLD tFLD tRHD tFHD HDRV rise time VDD = 3.3 V, VDD = 3.3 V, ISOURCE = −100 mA ISINK = 100 mA 3 5.5 1.0 2.0 LDRV rise time 15 35 LDRV fall time 10 25 15 35 10 25 CLOAD = 1 nF HDRV fall time Ω ns PREDICTIVE DELAY VSWP TLDHD THDLD Sense threshold to modulate delay time −350 mV Maximum delay modulation range time LDRV OFF − to − HDRV ON 45 70 95 Predictive counter delay time per bit LDRV OFF − to − HDRV ON 2.8 4.3 6.2 Maximum delay modulation range HDRV OFF − to − LDRV ON 50 80 110 Predictive counter delay time per bit HDRV OFF − to − LDRV ON 3.0 4.8 6.6 0.21 0.26 0.31 0.25 0.29 0.35 ns SHUTDOWN VSD VEN Shutdown threshold voltage Outputs OFF Device active threshold voltage V SOFTSTART ISS VSS Soft-start source current Outputs OFF Soft-start voltage to begin VOUT start 2.0 3.7 5.4 µA 0.35 0.65 0.95 V 50 100 35 70 BOOTSTRAP RBOOT Bootstrap switch resistance VOUT PRE-BIAS Recommended VOUT pre-bias level as % of final regulation(1)(4) VDD = 3.3 V VDD = 5 V FB percent of 700 mV Ω 90% SW NODE ISW Leakage current in shutdown THERMAL SHUTDOWN tSD Shutdown temperature(1) 2 165 Restart from thermal shutdown(1) −15 (1) Ensured by design. Not production tested. (2) Derate the maximum duty cycle by 3% for VDD < 3 V. (3) Operating at PWM on-times of less than 100 ns could lead to overlap between HDRV and LDRV pulses. (4) Prebiased output greater than 90% of final regulation may lead to sinking current from the prebias output. 4 www.ti.com µA °C SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 Terminal Functions TERMINAL I/O DESCRIPTION NAME NO. BOOT 10 O Provides a bootstrapped supply for the topside MOSFET driver, enabling the gate of the topside MOSFET to be driven above the input supply rail COMP 3 O Output of the error amplifier FB 2 I Inverting input of the error amplifier. In normal operation the voltage at this pin is the internal reference level of 700 mV. GND 5 − Power supply return for the device. The power stage ground return on the board requires a separate path from other sensitive signal ground returns. HDRV 9 O This is the gate drive output for the topside N-channel MOSFET. HDRV is bootstrapped to near 2 × VDD for good enhancement of the topside MOSFET. ILIM 1 I A resistor is connected between this pin and VDD to set up the over current threshold voltage. A 15-µA current sink at the pin establishes a voltage drop across the external resistor that represents the drain-to-source voltage across the top side N-channel MOSFET during an over current condition. The ILIM over current comparator is blanked for the first 100 ns to allow full enhancement of the top MOSFET. Set the ILIM voltage level such that it is within 800 mV of VDD; that is, (VDD − 0.8) ≤ IILIM ≤ VDD. LDRV 6 O Gate drive output for the low-side synchronous rectifier N-channel MOSFET I Soft-start and overcurrent fault shutdown times are set by charging and discharging a capacitor connected to this pin. A closed loop soft-start occurs when the internal 3-µA current source charges the external capacitor. There is a 0.65-V offset between external SS pin and internal soft-start voltage at the error amplifier input. This allows the device to be enabled before starting VOUT, thus ensuring that VOUT soft starts smoothly. When the SS/SD voltage is less than 0.25 V, the device is shutdown and the HDRV and LDRV are driven low. In normal operation, the capacitor is charged to VDD. When a fault condition is asserted, the soft-start capacitor goes through six charge/discharge cycles, restarting the converter on the seventh cycle. SS/SD 4 SW 8 O Connect to the switched node on the converter. This pin is used for overcurrent sensing in the topside N-channel MOSFET, and level sensing for predictive delay circuit. Overcurrent is determined, when the topside N-channel MOSFET is on, by comparing the voltage on SW with respect to VDD and the voltage on the ILIM with respect to VDD. This pin is also used for the return of the topside N-channel MOSFET driver. VDD 7 I Power input for the chip, 5.5-V maximum. Decouple close to the pin with a low-ESR capacitor, 1-µF or larger. FUNCTIONAL BLOCK DIAGRAM VDD VDD 7 VDD + + CLK PWM 0.65 V UVLO 3.7 µA SS ACTIVE SOFT START 9 HDRV PWM LOGIC 8 SW 6 LDRV 1 ILIM PREDICTIVE GATE DRIVE (VDD−1.2 V) FAULT FAULT COUNTER OC VDD DISCHARGE 0.26 V BOOT UVLO 3 4 10 HI OSC REF SS/SD PWM COMP ERROR AMPLIFIER 2 0.7 V COMP LDRV UVLO 2V FB THERMAL SHUTDOWN 100 ns DELAY SHUT DOWN LO EN GND 5 CURRENT LIMIT COMP 15 µA UDG−03162 www.ti.com 5 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION The TPS4000x series of synchronous buck controller devices is optimized for high-efficiency dc-to-dc conversion in non-isolated distributed power systems. A typical application circuit is shown in Figure 1. The TPS40007 and TPS40009 are the controllers of choice for general-purpose synchronous buck designs. They are designed to startup into applications where the output voltage is pre-biased, and without having the synchronous rectifier interfere with the pre-bias condition. PWM pulses are enabled when the soft-start voltage crosses the feedback level dictated by the pre-bias output. Moreover, the pre-biased output ramps up smoothly from its pre-bias value and into regulation. VDD 3.0 V to 5.5 V 10 µF 100 µF 20 kW TPS40007 3.6 nF 1 ILIM BOOT 10 2 FB HDRV 9 3 COMP SW 8 7.68 kΩ Si4866DY IHLP5050CE−01 100 nF 470 µF 100 pF VOUT 1.8 V 10 A 4 SS/SD VDD 7 LDRV 6 10 µF 243 Ω Si4866DY 4.7 nF 15.7 kΩ 5 GND 3.3 nF 10 kΩ UDG−03159 Figure 1. Typical Application Circuit 6 www.ti.com SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION ERROR AMPLIFIER The error amplifier has a bandwidth of greater than 5 MHz, with open loop gain of at least 55 dB. The COMP output voltage is clamped to a level above the oscillator ramp in order to improve large-scale transient response. OSCILLATOR The oscillator uses an internal resistor and capacitor to set the oscillation frequency. The ramp waveform is a sawtooth at the PWM frequency with a peak voltage of 1.25 V, and a valley of 0.31 V. The PWM duty cycle is limited to a maximum of 94%, allowing the bootstrap capacitor to charge during every cycle. BOOTSTRAP/CHARGE PUMP There is an internal switch between VDD and BOOT. This switch charges the external bootstrap capacitor for the floating supply. If the resistance of this switch is too high for the application, an external schottky diode between VDD and BOOT can be used. The peak voltage on the bootstrap capacitor is approximately equal to VDD. DRIVER The HDRV and LDRV MOSFET drivers are capable of driving gate-to-source voltages up to 5.5 V. At VIN, = 5 V and using appropriate MOSFETs, a 20-A converter can be achieved. The LDRV driver switches between VDD and ground, while the HDRV driver is referenced to SW and switches between BOOT and SW. SYNCHRONOUS RECTIFICATION AND PREDICTIVE DELAY In a normal buck converter, when the main switch turns off, current is flowing to the load in the inductor. This current cannot be stopped immediately without using infinite voltage. In order to provide a path for current to flow and maintain voltage levels at a safe level, a rectifier or catch device is used. This device can be either a conventional diode, or it can be a controlled active device if a control signal is available to drive it. The TPS4000x provides a signal to drive an N-channel MOSFET as a rectifier. This control signal is carefully coordinated with the drive signal for the main switch so that there is minimum delay from the time that the rectifier MOSFET turns off and the main switch turns on, and minimum delay from when the main switch turns off and the rectifier MOSFET turns on. This scheme, Predictive Gate Drivet delay, uses information from the current switching cycle to adjust the delays that are to be used in the next cycle. Figure 2 shows the switch-node voltage waveform for a synchronously rectified buck converter. Illustrated are the relative effects of a fixed-delay drive scheme (constant, pre-set delays for the turn-off to turn-on intervals), an adaptive delay drive scheme (variable delays based upon voltages sensed on the current switching cycle) and the predictive delay drive scheme. Note that the longer the time spent in diode conduction during the rectifier conduction period, the lower the efficiency. Also, not described in Figure 2 is the fact that the predictive delay circuit can prevent the body diode from becoming forward biased at all. This results in a significant power savings when the main MOSFET turns on, and minimizes reverse recovery loss in the body diode of the rectifier MOSFET. www.ti.com 7 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION GND Channel Conduction Body Diode Conduction Fixed Delay Adaptive Delay Predictive Delay UDG−03166 Figure 2. Switch Node Waveforms for Synchronous Buck Converter SHORT CIRCUIT PROTECTION Overcurrent conditions in the TPS4000x are sensed by detecting the voltage across the main MOSFET while it is on. Basic Description If the voltage exceeds a pre-set threshold, the current pulse is terminated, and a counter inside the device is incremented. If this counter fills up, a fault condition is declared and the device disables switching for a period of time and then attempts to restart the converter with a full soft-start cycle. 8 www.ti.com SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION Detailed Description During each switching cycle, a comparator looks at the voltage across the top side MOSFET while it is on. This comparator is enabled after the SW node reaches a voltage greater than (VDD−1.2 V) followed by a 100-ns blanking time. If the voltage across that MOSFET exceeds the programmed voltage, the current-switching pulse is terminated and a 3-bit counter is incremented by one count. If, during the switching cycle, the topside MOSFET voltage does not exceed a preset threshold, then this counter is decremented by one count. (The counter does not wrap around from 7 to 0 or from 0 to 7). If the counter reaches a full count of 7, the device declares that a fault condition exists at the output of the converter. In this fault state, HDRV and LDRV are turned off, and the soft-start capacitor is discharged. LDRV is maintained OFF during fault timeout to effectively support pre-bias applications. The counter is decremented by one by the soft start capacitor (CSS) discharge. When the soft-start capacitor is fully discharged, the discharging circuit is turned off and the capacitor is allowed to charge up at the nominal charging rate. When the soft-start capacitor reaches approximately 1.3 V, it is discharged again and the overcurrent counter is decremented by one count. The capacitor is charged and discharged, and the counter decremented until the count reaches zero (a total of six times). When this happens, the outputs are again enabled as the soft-start capacitor generates a reference ramp for the converter to follow while attempting to restart. During this soft-start interval (whether or not the controller is attempting to do a fault recovery or starting for the first time), pulse-by-pulse current limiting is in effect, but overcurrent pulses are not counted to declare a fault until the soft-start cycle has been completed. It is possible to have a supply attempt to bring up a short circuit for the duration of the soft start period plus seven switching cycles. Power stage designs should take this into account if it makes a difference thermally. Figure 3 shows the details of the overcurrent operation. (+) VTS (−) Short Circuit Protection Threshold Voltage Internal PWM VTS 0V External Main Drive Normal Cycle Overcurrent Cycle UDG−03165 Figure 3. Short Circuit Operation www.ti.com 9 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION Figure 4 shows the behavior of key signals during initial startup, during a fault and a successfully fault recovery. At time t0, power is applied to the converter. The voltage on the soft-start capacitor (VCSS) begins to ramp up. At t1, the soft-start period is completed and the converter is regulating its output at the desired voltage level. From t0 to t1, pulse-by-pulse current limiting is in effect, and from t1 onward, overcurrent pulses are counted for purposes of determining a possible fault condition. At t2, a heavy overload is applied to the converter. This overload is in excess of the overcurrent threshold. The converter starts limiting current and the output voltage falls to some level depending on the overload applied. During the period from t2 to t3, the counter is counting overcurrent pulses, and at time t3 reaches a full count of 7. The soft-start capacitor is then discharged, the counter is decremented, and a fault condition is declared. VDD VCSS ~ 1.3 V 0.6 V ~ 0.6 V FAULT ILOAD VOUT t t0 COUNTER t1 t4 t2 t3 0 6 t5 5 t6 4 t7 3 t8 2 t9 1 t10 0 1 2 3 4 5 6 7 UDG−03160 Figure 4. Overcurrent/Fault Waveforms When the soft start capacitor is fully discharged, it begins charging again at the same rate that it does on startup, with a nominal 3.7-µA current source. When the capacitor voltage crosses 1.3 V, it is discharged again and the counter is decremented by one count. These transitions occur at t3 through t9. Not shown in Figure 4 is that between t3 and t9, LDRV is maintained OFF. At t9, the counter has been decremented to 0. The fault logic is then cleared, the outputs are enabled, and the converter attempts to restart with a full soft-start cycle. The converter comes into regulation at t10. 10 www.ti.com SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION SETTING THE CURRENT LIMIT Connecting a resistor from VDD to ILIM sets the current limit. A 15-µA current sink internal to the device causes a voltage drop at ILIM that becomes the short circuit threshold. Ensure that (VDD−0.8 V) ≤ VILIM ≤ VDD. The tolerance of the current sink is too loose to do an accurate current limit. The main purpose is for hard fault protection of the power switches. Given the tolerance of the ILIM sink current, and the RDS(on) range for a MOSFET, it is generally possible to apply a load that thermally damages the converter. This device is intended for embedded converters where load characteristics are defined and can be controlled. A local capacitor (with a value 50 pF to 150 pF) placed across the resistor between VDD and ILIM may improve coupling a common mode noise between VDD and ILIM. SOFT-START AND SHUTDOWN These two functions are combined on the SS/SD pin. There is a VBE offset (0.65-V) between the external SS/SD pin and internal soft-start voltage at the error amplifier input, allowing the device to be enabled before starting VOUT as shown in Figure 5. This reduces the transient current required to charge the output capacitor at startup, and allows for a smooth startup with no overshoot of the output voltage. SS/SD (200 mV/ div) FB (200 mV/ div) t − Time − 1 ms/div Figure 5. Offset Between SS/SD and FB at Startup www.ti.com 11 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION A shutdown feature can be implemented as shown in Figure 6. The device shuts down when the voltage at the SS/SD pin falls below 260 mV. Because of this limitation, it is recommended that a MOSFET be used as the controlling device, as in Figure 6. During shutdown, the total leakage current on the SW pin (ISW) is less than 2 µA. When VSS/SD is greater than 290 mV, the device is enabled with normal SW active bias currents. TPS40007/9 3.7 µA SS/SD ERROR AMPLIFIER 4 C SS R 0.7 V FB + + COMP SHUTDOWN SDN 0.26 V + UDG−01163 Figure 6. Shutdown Implementation Long soft start times may experience extended regions where the PWM pulse width is less than 100 ns. This could lead to momentary overlap between HDRV and LDRV. As a result, there is a momentary increase in ground or supply noise. It is important to ensure that the ground return of the synchronous rectifier be connected directly to the ground return of the input bank of bypass capacitors, in order to minimize ground noise from interfering with the controller during soft start. Also, if an external shutdown transistor is used in the application, it is important to place a local bypass capacitor between its gate and source on the board in order to minimize noise from interfering with the controller during soft-start. OUTPUT PRE-BIAS The TPS4000x supports pre-biased VOUT voltage applications. In cases, where the VOUT voltage is held up by a pre-biasing supply while the controller is off, full synchronous rectification is disabled during the initial phase of soft starting the VOUT voltage. When the first PWM pulses are detected during soft-start, the controller slowly activates synchronous rectification by starting the first LDRV pulses with a narrow on-time. It then increments that on-time on a cycle-by-cycle basis until it coincides with the time dictated by (1−D), where D is the duty cycle of the converter. This scheme prevents the initial sinking the pre-bias output, and ensures that the VOUTvoltage starts and ramps up smoothly into regulation. Note, if the VOUT voltage is pre-biased, PWM pulses start when the error amplifier soft-start input voltage rises above the commanded FB voltage. Figure 7 depicts the waveforms of the HDRV and LDRV output signals at the beginning PWM pulses. When HDRV turns off, diode rectification is enabled. Before the next PWM cycle starts, LDRV is turned on for a short pulse. With every cycle, the leading edge of LDRV is modulated, and the on-time of the synchronous rectifier is increased. Eventually, the leading edge of LDRV coincides with the falling edge of HDRV to achieve full synchronous rectification. At most, synchronous rectifier modulation takes place for the first 128 cycles after PWM pulses start. Note that during the synchronous rectifier modulation region, the controller monitors pulse skipping. If the main HDRV skips a pulse, the controller also skips a LDRV pulse. Pulse skipping could be experienced if the loop response is much faster than the commanding soft-start ramp, especially when soft start times are long. The output voltage ratchets up as the soft-start ramp catches up to it. Appropriate setting of loop response curbs this effect. During normal regulation of the VOUT voltage, the controller operates in full two-quadrant source/sink mode. 12 www.ti.com SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION Figure 8 shows startup waveforms of a 1.2-V VOUT voltage under different pre-bias scenarios. The first trace is when the output voltage starts with zero pre−bias. The second and third traces, respectively, the pre-bias levels are 0.5 V and 1.0 V. VIN = 5 V VOUT = 1.2 V (200 mV/div) PREBIAS = 1 V VHDRV PREBIAS = 0.5 V PREBIAS = 0 V VLDRV t − Time − 2 µs/div t − Time − 500 µs/div Figure 7. MOSFET Drivers at Beginning of Soft-Start Figure 8. Startup Waveforms The recommended VOUT voltage pre-bias range is less than or equal to 90% of final regulation. That is, a pre-bias level between 90% and 100% of final regulation could lead to sinking the pre-bias supply. If the VOUT voltage is initially set to higher than 100% of final regulation, the controller forces sinking current at the end of soft-start in order to bring the output quickly into regulation. The following pages include design ideas for a few applications. For more ideas, detailed design information, and helpful hints, visit the TPS40000 resources at http://power.ti.com. www.ti.com 13 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION VDD 3.3 V 22 µF TPS40009 15 kΩ 1 nF 1 ILIM BOOT 10 2 FB HDRV FDS6894A 1.0 µH 9 VOUT 1.2 V 5A 1 µF 8.66 kΩ 68 pF 22 µF 3 COMP SW 8 4 SS/SD VDD 7 5 GND LDRV 6 2.2 Ω 22 µF 22 µF FDS6894A 4.7 nF PWP 1 µF 0.0033 µF 12.1 kΩ 470 pF 1 kΩ 16.9 kΩ UDG−03164 Figure 9. Small-Form Factor Converter for 3.3 V to 1.2 V at 5 A. 14 www.ti.com + 3.3 V VDD 330 µF 4.7 nF 6.19 kΩ 82 pF 2.2 nF 330 µF + www.ti.com LDRV VDD SW HDRV 6 7 8 9 BOOT 10 PWP GND SS/SD 4 5 COMP FB 2 3 ILIM TPS40007 1 11 kΩ 22 µF 1 µF 1 µF 22 µF Si4866DY Si4866DY 10 nF 2.2 Ω 1.5 µH 22 µF 22 µF 14 kW 22 µF 392 Ω 10 kΩ 22 µF 22 µF 1200 pF VOUT 1.2 V 10 A SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION UDG−04014 Figure 10. High-Current Converter for 3.3 V to 1.2 V at 10 A. 15 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION VDD 2.5 V 100 pF 22 µF FDS6894A TPS40009 1500 pF 22 µF BAT54 15 kΩ 1 µF 1 ILIM BOOT 10 2 FB HDRV 9 3 COMP SW 8 4 SS/SD VDD 7 1.0 µH L1 VOUT 1.2 V 5A 5.62 kΩ 2.2 Ω 22 µF 4.7 nF 22 µF FDS6894A 5 GND LDRV PWP 6 1 µF 3.3 nF 6.19 kΩ 536 Ω 1000 pF 8.66 kΩ UDG−04028 Figure 11. Ultra-Low-Input Voltage Converter for 2.5 V to 1.2 V at 5 A 16 www.ti.com SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION VIN = 3.3 V 1 2 + J1 3 + C2 330 µF C1 330 µF 4 R2 16.2 kΩ TPS40007DGQ 1 ILIM C7 1.5 nF 2 FB R4 5.9 kΩ 3 COMP 4 SS/SD C11 180 pF C13 4.7 nF C3 22 µF C5 22 µF C4 22 µF C6 1 µF Q1 Si4866DY BOOT 10 L1 1.0 µH HDRV 9 VOUT = 2.5 V 10 A SW 8 1 Q2 Si4866DY VDD 7 5 GND LDRV 6 PWP R3 2.2 Ω + 2 + 3 J2 C12 10n F C8 470 µF C9 470 µF 4 C14 1 µF R6 10 kΩ R7 698 Ω C15 6.8 nF R8 3.92 kΩ UDG−03169 Figure 12. TPS40007EVM−001 Ultra-High-Efficiency Converter for 3.3 V to 2.5 V at 10 A www.ti.com 17 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 APPLICATION INFORMATION Layout Considerations Successful operation of the TPS4000x controllers is dependent upon proper converter layout and grounding techniques. High current returns for the SR MOSFET’s source, and ground connection of the input and output capacitors, should be kept on a single ground plane. Bypassing capacitors at the device should return closely to the GND (pin 5) of the device. The GND (pin 5) and PowerPAD should connect together at the device and return to the main ground plane. Proper operation of the Predictive Gate Drive circuits is dependent upon detecting low-voltage thresholds on the SW node. To ensure that the signal at the SW pin accurately represents the voltage at the main switching node, the connection from SW (pin 8) to the main switching node of the converter should be kept as short and as wide as possible. If the SW trace should traverse multiple board layers between the device and the MOSFETs, multiple vias should be used. Gate drive outputs, LDRV and HDRV, should be kept as short as possible to minimize inductances of the traces. While the controller does not require the usage of external resistors between the driver pins and the gates of the MOSFETs, adding small resistors in series with very high gate charge MOSFETs could minimize the effects of high frequency ringing. The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD derives its name and low thermal impedance from the large bonding pad on the bottom of the device. The circuit board must have an area of solder-tinned-copper underneath the package. The dimensions of this area depend on the size of the PowerPAD package (See Thermal Pad Mechanical Data on page 21) 18 www.ti.com SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 TYPICAL CHARACTERISTICS OSCILLATOR FREQUENCY PERCENT CHANGE vs INPUT VOLTAGE OSCILLATOR FREQUENCY PERCENT CHANGE vs TEMPERATURE ∆fOSC − Change in Oscillator Frequency − % ∆fOSC − Change in Oscillator Frequency − % 6 5 4 3 2 1 0 2.0 2.5 3.0 3.5 4.0 4.5 5.0 1 0 −1 −2 −3 −4 −5 −6 5.5 −50 −25 0 25 50 75 100 125 Temperature − °C VIN − Input Voltage − V Figure 14 Figure 13 FEEDBACK VOLTAGE vs INPUT VOLTAGE FEEDBACK VOLTAGE vs TEMPERATURE 0.707 0.7010 0.7005 VFB − Feedback Voltage − V VFB − Feedback Voltage − V 0.705 0.7000 0.6995 0.703 0.701 0.699 0.697 0.695 0.6990 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 0.693 VIN − Input Voltage − V Figure 15 −50 −25 0 25 50 Temperature − °C 75 100 125 Figure 16 www.ti.com 19 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 TYPICAL CHARACTERISTICS CURRENT LIMIT SINK CURRENT vs INPUT VOLTAGE CURRENT LIMIT SINK CURRENT vs TEMPERATURE 16.0 15.0 ILIMIT − Sink Current Limit − µA ILIMIT − Sink Current Limit − µA 15.5 14.5 14.0 13.5 15.5 15.0 14.5 13.0 12.5 14.0 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 −50 0 25 50 Temperature − °C VIN − Input Voltage − V Figure 17 Figure 18 SHORT CIRCUIT PROTECTION SS/SD Node SW Node t − Time − 1 ms/div Figure 19 20 −25 www.ti.com 75 100 125 SLUS589A− NOVEMBER 2003 − REVISED MAY 2004 THERMAL PAD MECHANICAL DATA PowerPADt PLASTIC SMALL-OUTLINE DGQ (S−PDSO−G10) www.ti.com 21 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. 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