OPA4872-EP OP A4 87 2 www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 4:1 HIGH-SPEED MULTIPLEXER FEATURES APPLICATIONS • • • • • • • • • • • • • • 1 2 500-MHz Small-Signal Bandwidth 500-MHz, 2-VPP Bandwidth 0.1-dB Gain Flatness to 120 MHz 10-ns Channel-Switching Time Low Switching Glitch: 40 mVPP 2300-V/µs Slew Rate 0.035%/0.005° Differential Gain, Phase Quiescent Current = 10.6 mA 1.1-mA Quiescent Current in Shutdown Mode 88-dB Off Isolation in Disable or Shutdown (10 MHz) Video Router LCD and Plasma Display High Speed PGA Drop-In Upgrade to AD8174 SUPPORTS DEFENSE, AEROSPACE, AND MEDICAL APPLICATIONS • • • • • • • (1) Controlled Baseline One Assembly/Test Site One Fabrication Site Available in Military (–55°C/125°C), Temperature Range (1) Extended Product Life Cycle Extended Product-Change Notification Product Traceability Additional temperature ranges available - contact factory DESCRIPTION The OPA4872 offers a very wideband 4:1 multiplexer in an SO-14 package. Using only 10.6 mA, the OPA4872 provides a user-settable output amplifier gain with greater than 500-MHz large-signal bandwidth (2 VPP). The switching glitch is improved over earlier solutions using a new (patented) input stage switching approach. This technique uses current steering as the input switch while maintaining an overall closed-loop design. The OPA4872 exhibits an off isolation of 88dB in either Disable or Shutdown mode. With greater than 500-MHz small-signal bandwidth at a gain of 2, the OPA4872 gives a typical 0.1-dB gain flatness to greater than 120 MHz. System power may be optimized using the chip-enable feature for the OPA4872. Taking the chip enable (EN) line high powers down the OPA4872 to less than 3.4 mA total supply current. Further power reduction to 1.1mA quiescent current can be achieved by bringing the shutdown (SD) line high. Muxing multiple OPA4872s outputs together, then using the chip enable to select which channels are active, increases the number of possible inputs. 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. UNLESS OTHERWISE NOTED this document contains PRODUCTION DATA information current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2008, Texas Instruments Incorporated OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com +5V 50W OPA695 G = 1V/V -5V +5V 523W 50W IN0 +5V OPA4872 SD EN 50W OPA695 G = 2V/V -5V IN1 511W 50W 511W IN2 +5V 523W 50W To 50W Load 523W OPA695 G = 4V/V Logic -5V 453W IN3 A0 A1 -5V 149W +5V 50W OPA695 G = 8V/V -5V 402W 57.6W 2-Bit, High-Speed PGA, Greater Than 300MHz Channel Bandwidth 2 Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION (1) (1) (2) PACKAGE-LEAD PACKAGE DESIGNATOR (2) SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER TRANSPORT MEDIA, QUANTITY SO-14 D –55°C to 125°C OPA4872M OPA4872MDREP Tape and Reel, 2500 For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. Package drawings, standard packing quantities, thermal data, symbolization, and PCB design guidelines are available at www.ti.com/sc/package. ABSOLUTE MAXIMUM RATINGS (1) Over operating free-air temperature range, unless otherwise noted. Power supply OPA4872 UNIT ±6.5 V Internal power dissipation See Thermal Characteristics Input voltage range Storage temperature range ±VS V –65 to +125 °C Lead temperature (soldering, 10s) +260 °C Junction temperature (TJ) +150 °C Junction temperature: continuous operation, long-term reliability ESD rating (1) +140 °C Human body model (HBM) 1500 V Charged device model (CDM) 1000 V Machine model (MM) 200 V Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. PIN CONFIGURATION SO-14 Top View OPA4872 1 14 V+ GND 2 13 OUT IN1 3 12 FB GND 4 11 SD IN2 5 10 EN V- 6 9 A1 IN3 7 8 A0 Logic IN0 Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 3 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com ELECTRICAL CHARACTERISTICS: VS = ±5 V At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. TYP PARAMETER MIN/MAX OVER TEMPERATURE UNITS MIN/ MAX 500 MHz min B 120 MHz typ C CONDITIONS +25°C Small-signal bandwidth VO = 500 mVPP, RL = 150 Ω Bandwidth for 0.1 dB flatness VO = 500 mVPP, RL = 150 Ω +25°C (2) –55°C to +125°C (3) TEST LEVEL (1) AC PERFORMANCE VO = 2 VPP, RL = 150 Ω 500 MHz min B Slew rate 4 V step 2300 V/µs min B Rise time and fall time 4 V step 1.25 ns max B to 0.05% 2 V step 15 ns typ C to 0.1% 2 V step 14 ns max B 10 ns max B Large-signal bandwidth Settling time Channel switching time Harmonic distortion G = +2 V/V, f = 10 MHz, VO = 2 VPP 2nd-harmonic RL = 150 Ω -60 dBc max B 3rd-harmonic RL = 150 Ω -78 dBc max B Input voltage noise f > 100 kHz 4.5 nV/√Hz max B Noninverting input current noise f > 100 kHz 4.0 pA/√Hz max B Inverting input current noise f > 100 kHz 19 pA/√Hz max B G = +2 V/V, PAL, VO = 1.4 VP 0.035 % typ C Differential gain Differential phase All hostile crosstalk, input-referred G = +2 V/V, PAL, VO = 1.4 VP 0.005 ° typ C Three channels driven at 5 MHz, 1 VPP -80 dB typ C Three channels driven at 30 MHz, 1 VPP -66 dB typ C VO = 0 V, RL = 100 Ω 103 92 60 kΩ min A VCM = 0 V ±1 ±5 ±10.5 mV max A ±30 µV/°C max B DC PERFORMANCE Open-loop transimpedance (ZOL) Input offset voltage Average Input offset voltage drift VCM = 0 V Input offset voltage matching VCM = 0 V ±1 ±5 ±10.5 mV max A VCM = 0 V ±4 ±14 ±20 µA max A ±48 nA/°C max B ±35 µA max A ±125 nA/°C max B Noninverting input bias current Average noninverting input bias current Inverting bias current Average inverting input bias current VCM = 0 V VCM = 0 V ±4 ±18 VCM = 0 V INPUT Common-mode input range (CMIR) Common-mode rejection ratio (CMRR) Each noninverting input ±2.7 ±2.55 ±2.4 V min A VCM = 0 V, input-referred, noninverting input 56 50 43 dB min A Channel enabled 2.5 MΩ typ C open loop 70 Ω typ C Input resistance Noninverting Inverting Input capacitance Noninverting Channel selected 0.9 pF typ C Channel deselected 0.9 pF typ C Chip disabled 0.9 pF typ C OUTPUT RL ≥ 1 kΩ ±4 ±3.9 ±3.55 V min A RL = 150 Ω ±3.7 ±3.55 ±3.35 V min A VO = 0 V ±75 ±48 ±38 mA min A Short-circuit output current Output shorted to ground ±100 mA typ C Closed-Loop output impedance G = +2 V/V, f ≤ 100 kHz 0.03 Ω typ C Output voltage swing Output current (1) (2) (3) 4 Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. Junction temperature = ambient for +25°C tested specifications. Junction temperature = ambient at low temperature limit; junction temperature = ambient +9°C at high temperature limit for over temperature specifications. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 ELECTRICAL CHARACTERISTICS: VS = ±5 V (continued) At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. TYP PARAMETER MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C (2) –55°C to +125°C (3) UNITS MIN/ MAX TEST LEVEL (1) VEN = 0 V 3.4 3.6 4.3 mA max A VIN = ±0.25 VDC 25 ns typ C ENABLE (EN) Power-down supply current Disable time Enable time VIN = ±0.25 VDC 6 ns typ C Off isolation G = +2 V/V, f = 10 MHz 88 dB typ C Output resistance in disable 14 MΩ typ C Output capacitance in disable 2.5 pF typ C DIGITAL INPUTS Maximum logic 0 A0, A1, EN, SD 0.8 0.8 V max B Minimum logic 1 A0, A1, EN, SD 2.0 2.0 V min B 40 55 µA max A Logic input current Output switching glitch A0 , A1, EN, SD, input = 0V each line 32 Channel selection, at matched load ±20 mV typ C Channel disable, at matched load ±40 mV typ C Shutdown, at matched load ±40 mV typ C mA max A ns typ C SHUTDOWN Shutdown supply current VSD = 0 V 1.1 VIN = ±0.25 VDC 75 Enable time VIN = ±0.25 VDC 15 ns typ C Off isolation G = +2 V/V, f = 10 MHz 88 dB typ C Output resistance in shutdown 14 MΩ typ C Output capacitance in shutdown 2.5 pF typ C Shutdown time 1.3 2.0 POWER SUPPLY Specified operating voltage V typ C Minimum operating voltage ±5 ±3.5 ±3.5 V min B Maximum operating voltage ±6.0 ±6.0 V max A Maximum quiescent current VS = ±5 V 10.6 11 12.5 mA max A Minimum quiescent current VS = ±5 V 10.6 10 8.25 mA min A (+PSRR) Input-referred –56 –50 –42 dB min A (–PSRR) Input-referred –57 –51 –43 dB min A –55 to +125 °C typ C 80 °C/W typ C Power-supply rejection ratio THERMAL CHARACTERISTICS Specified operating range, D package Thermal resistance, θJA D Junction-to-ambient SO-14 Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 5 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com TYPICAL CHARACTERISTICS At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. SMALL-SIGNAL FREQUENCY RESPONSE 7 SMALL-SIGNAL FREQUENCY RESPONSE 0.3 6 0.1 Normalized 4 0 Flatness 3 -0.1 2 -0.2 VO = 500mVPP RL = 150W G = +2V/V -0.3 3 Normalized Gain (dB) -3 G = +2V/V -6 -9 G = +4V/V 0 -0.4 1M 0 10M 100M -12 1G 1M 10M Frequency (Hz) 100M Figure 1. LARGE-SIGNAL FREQUENCY RESPONSE NONINVERTING PULSE RESPONSE Small-Signal Output Voltage (V) 0.5 6 VO = 2VPP 5 4 VO = 1VPP 3 VO = 0.5VPP 2 1 0 2G Figure 2. 7 Gain (dB) 1G Frequency (Hz) VO = 4VPP 2.5 RL = 150W G = +2V/V 0.4 2.0 Large-Signal 4VPP Right Scale 0.3 1.5 0.2 1.0 Small-Signal 0.4VPP Left Scale 0.1 0.5 0 0 -0.1 -0.5 -0.2 -1.0 -0.3 -1.5 -2 -0.4 -2.0 -3 -0.5 -1 0 200M 400M 600M 800M 1G Large-Signal Output Voltage (V) Gain (dB) 5 Normalized Gain Flatness (dB) 0.2 1 G = +1V/V VO = 500mVPP Bandwidth 6 -2.5 Time (10ns/div) Frequency (Hz) Figure 3. Figure 4. RECOMMENDED RS vs CAPACITIVE LOAD FREQUENCY RESPONSE vs CAPACITIVE LOAD 8 90 7 Gain to Capacitive Load (dB) 100 80 RS (W) 70 60 50 40 30 20 10 5 4 3 VI 75W 2 1 75W 1 10 100 1000 + CL = 22pF RS VO - 0 75W -1 523W CL (1) 1kW CL = 47pF 523W NOTE: (1) Optional. -2 0 6 CL = 10pF 6 75W CL = 100pF -3 1 10 Capacitive Load (pF) Frequency (MHz) Figure 5. Figure 6. Submit Documentation Feedback 100 300 Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 TYPICAL CHARACTERISTICS (continued) At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. HARMONIC DISTORTION vs LOAD RESISTANCE HARMONIC DISTORTION vs SUPPLY VOLTAGE -55 VO = 2VPP f = 5MHz -60 2nd-Harmonic -65 -70 -75 -80 3rd-Harmonic 2nd-Harmonic -60 Harmonic Distortion (dBc) Harmonic Distortion (dBc) -55 -85 -65 -70 -75 3rd-Harmonic -80 -85 VO = 2VPP RL = 150W f = 5MHz -90 dBc = dB Below Carrier -90 100 2.5 1k 5.0 -55 Harmonic Distortion (dBc) 2nd-Harmonic -70 -75 3rd-Harmonic -80 -85 -90 10 6.0 RL = 150W f = 5MHz -60 2nd-Harmonic -65 -70 -75 3rd-Harmonic -80 -85 -90 -95 dBc = dB Below Carrier -95 1 5.5 HARMONIC DISTORTION vs OUTPUT VOLTAGE -50 -65 -100 100 dBc = dB Below Carrier 0.5 1.5 Frequency (MHz) 5 4.5 Figure 8. -55 6 4.0 Figure 7. -50 -60 3.5 Supply Voltage (±VS) VO = 2VPP RL = 150W -45 3.0 Load Resistance (W) HARMONIC DISTORTION vs FREQUENCY -40 Harmonic Distortion (dBc) dBc = dB Below Carrier -95 2.5 3.5 4.5 5.5 6.5 7.5 Output Voltage Swing (VPP) Figure 9. Figure 10. OUTPUT VOLTAGE AND CURRENT LIMITATIONS DISABLE AND SHUTDOWN FEEDTHROUGH vs FREQUENCY 1W Internal -20 Power Limit -30 Input-referred 100W Load 3 -40 Feedthrough (dB) Output Voltage (V) 4 2 1 0 50W Load 25W Load -1 -2 1W Internal -3 Power Limit -4 -5 -200 -50 Shutdown Feedthrough -60 -70 -80 Disable Feedthrough -90 -100 -100 0 100 200 300 -110 1M Output Current (mA) 10M 100M 1G Frequency (Hz) Figure 11. Figure 12. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 7 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com TYPICAL CHARACTERISTICS (continued) At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. VIN_Ch0 = 200MHz, 0.7VPP VIN_Ch1 = 0VDC A0 A0 Time (10ns/div) Output (mV) 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 VIN_Ch0 = 200MHz, 0.10VPP EN Enable Voltage (V) Output (V) DISABLE/ENABLE SWITCHING GLITCH Output 75 50 25 0 -25 -50 -75 At Matched Load EN Time (10ns/div) Time (10ns/div) Figure 15. Figure 16. Output (mV) SHUTDOWN GLITCH Output SD 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 Channel Select (V) Output (V) SHUTDOWN/START-UP TIME VIN_Ch0 = 200MHz, 0.7VPP 75 50 25 0 -25 -50 -75 Time (20ns/div) At Matched Load SD Time (20ns/div) Figure 17. 8 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 Figure 14. DISABLE/ENABLE TIME 0.75 0.50 0.25 0 -0.25 -0.50 -0.75 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 Time (10ns/div) Figure 13. 0.75 0.50 0.25 0 -0.25 -0.50 -0.75 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 Channel Select (V) 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 At Matched Load Enable Voltage (V) Output Voltage 75 50 25 0 -25 -50 -75 Shutdown Voltage (V) Output (mV) CHANNEL-TO-CHANNEL SWITCHING GLITCH Channel Select (V) Output (V) CHANNEL-TO-CHANNEL SWITCHING 0.75 0.50 0.25 0 -0.25 -0.50 -0.75 Figure 18. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 TYPICAL CHARACTERISTICS (continued) At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE vs FREQUENCY ALL HOSTILE CROSSTALK vs FREQUENCY 0 1M Input-referred 0 < ZOL -10 -30 -40 -50 -60 -70 -80 ½ZOL½ 10k -90 1k -135 100 -180 Phase (°) Transimpedance (W) Crosstalk (dB) -45 100k -20 -90 -100 -225 10 1M 10M 100M 10k 1G 100k 1M 10M 100M 1G Frequency (Hz) Frequency (Hz) Figure 19. Figure 20. CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY INPUT IMPEDANCE vs FREQUENCY 1M 100M Disabled 10M Disabled (or Shutdown) Input Impedance (W) Output Impedance (W) 100k 10k 1k 100 10 1M Enabled 100k 10k 1k 1 Enabled 100 1M 10M 100M 1G 10k 100k 1M Frequency (Hz) Figure 21. PSRR vs FREQUENCY 1G OUTPUT AND SUPPLY CURRENT vs TEMPERATURE Input-referred Output Current (mA) Power-Supply Rejection Ratio (dB) 100M Figure 22. 60 50 10M Frequency (Hz) -PSRR 40 30 +PSRR 20 10 78.00 15.00 76.75 13.75 75.50 12.50 Supply Current (IQ) 74.25 11.25 73.00 10.00 71.75 +IOUT 70.50 7.50 69.25 6.25 -IOUT 68.00 0 1k 10k 100k 1M 10M 100M 1G 8.75 Supply Current (mA) 0.1 100k 5.00 -50 -25 Frequency (Hz) Figure 23. 0 25 50 75 100 125 Ambient Temperature (°C) Figure 24. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 9 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com TYPICAL CHARACTERISTICS (continued) At TA = +25°C, G = +2 V/V, RF = 523 Ω, and RL = 150 Ω, unless otherwise noted. INPUT VOLTAGE AND CURRENT NOISE 1.5 8 1.0 7 VOS 0.5 6 0 5 Ibn -0.5 4 -1.0 3 -1.5 2 Ibi -2.0 1 -2.5 0 -3.0 -1 -50 -25 0 25 50 75 100 125 300 Voltage Noise Density (nV/ÖHz) Current Noise Density (pA/ÖHz) 9 Input Bias Current (mA) Output Offset Voltage (mV) TYPICAL DC DRIFT OVER TEMPERATURE 2.0 100 Inverting Input Current Noise (19pA/ÖHz) 10 Input Voltage Noise (4.5nV/ÖHz) Noninverting Input Current Noise (4pA/ÖHz) 1 10 Ambient Temperature (°C) 1k 10k 100k 1M 10M Frequency (Hz) Figure 25. 10 100 Figure 26. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 APPLICATION INFORMATION input of a current feedback amplifier. Depending on the logic applied to channel control pins A0 and A1, one switch is on at all times. Figure 27 represents the OPA4872 in this configuration. The truth table for channel selection is shown in Table 1. WIDEBAND MULTIPLEXER OPERATION The OPA4872 gives a new level of performance in wideband multiplexers. Figure 27 shows the dc-coupled, gain of +2 V/V, dual power-supply circuit used as the basis of the ±5-V Electrical Characteristics and Typical Characteristic curves. For test purposes, the input impedance is set to 75 Ω with a resistor to ground and the output impedance is set to 75 Ω with a series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins while load powers (in dBm) are defined at a matched 75-Ω load. For the circuit of Figure 27, the total effective load will be 150 Ω || 1046 Ω = 131 Ω. Logic pins A0 and A1 control which of the four inputs is selected while EN and SD allow for power reduction. One optional component is included in Figure 27. In addition to the usual power-supply decoupling capacitors to ground, a 0.01-µF capacitor is included between the two power-supply pins. In practical printed circuit board (PCB) layouts, this optional added capacitor typically improves the 2nd-harmonic distortion performance by 3 dB to 6 dB for bipolar supply operation. Table 1. TRUTH TABLE A0 A1 EN SD VOUT 0 0 0 0 IN0 1 0 0 0 IN1 0 1 0 0 IN2 1 1 0 0 IN3 X X 1 0 High-Z, IQ = 3.4 mA X X X 1 High-Z, IQ = 1.1 mA The OPA4872 is in disable mode, with a quiescent current of 3.4mA typical, when the EN pin is set to 0V. After being placed in disable mode, the OPA4872 is fully enabled in 6ns. For further power savings, the SD pin can be used. Setting the SD pin to 5V places the device in shutdown mode with a standing quiescent current of 1.1 mA. Note that in this shutdown mode, the OPA4872 requires 15ns to be fully powered again. The truth table for disable and shutdown modes can be found in Table 1. Even though the internal architecture of the OPA4872 includes current steering, it is advantageous to look at it as four switches looking into the noninverting +5V 0.1mF + 6.8mF OPA4872 SD IN0 EN VIN0 75W IN1 VIN1 75W 75W VOUT IN2 523W VIN2 To 75W Load 75W 523W IN3 VIN3 A0 75W A1 Optional 0.01mF 0.1mF + 6.8mF -5V Figure 27. DC-Coupled, G = +2V/V Bipolar Specification and Test Circuit (Channel 0 Selected) Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 11 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com 2-BIT HIGH-SPEED PGA The OPA4872 can be used as a 2-bit, high-speed programmable gain amplifier (PGA) when used in conjunction with another amplifier. Figure 28 shows one OPA695 used in series with each OPA4876 input and configured with gains of +1 V/V, +2 V/V, +4 V/V, and +8 V/V, respectively. When channel 0 is selected, the overall gain to the matched load of the OPA4872 is 0dB. When channel 1 is selected, this circuit delivers 6 dB of gain to the matched load. When channel 2 is selected, this circuit delivers 12 dB of gain to the matched load. When channel 3 is selected, this circuit delivers 18 dB of gain to the matched load. +5V 50W OPA695 G = 1V/V -5V +5V 523W 50W IN0 +5V OPA4872 SD EN 50W OPA695 G = 2V/V -5V IN1 511W 50W 511W IN2 +5V 523W 50W To 50W Load 523W OPA695 G = 4V/V Logic -5V 453W IN3 A0 A1 -5V 149W +5V 50W OPA695 G = 8V/V -5V 402W 57.6W Figure 28. 2-Bit, High-Speed PGA, Greater Than 300MHz Channel Bandwidth 12 Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 2-BIT, HIGH-SPEED ATTENUATOR 8-TO-1 VIDEO MULTIPLEXER In contrast to the PGA, a two-bit high-speed attenuator can be implemented by using an R-2R ladder together with the OPA4872. Figure 29 shows such an implementation. Two OPA4872s can be used together to form an 8-input video multiplexer. The multiplexer is shown in Figure 31. Channel 0 sees the full input signal amplitude, where as channel 1 sees 1/2 VIN, channel 2 see 1/4 VIN and channel 3 sees 1/8 VIN. OPA4872 EN SD VIN OPA4872 2R IN0 SD EN RO = 69W R 2R 523W IN1 VOUT IN2 523W R R 523W 50W R 2R To 75W Load To 50W Load Logic 523W IN3 Logic OPA4872 EN A0 A1 SD Figure 29. 2-Bit, High-Speed Attenuator, 500MHz Channel Bandwidth 4-INPUT RGB ROUTER RO = 69W Three OPA4872s can be used together to form a four-input RGB router. The router for the red component is shown in Figure 30. Identical stages would be used for the green and blue channels. 523W 523W Logic +5V OPA4872 EN Figure 31. 8-to-1 Video Multiplexer SD R1 R2 75W Red Out 523W R3 523W To 75W Load When connecting OPA4872 outputs together, maintain a gain of +1V/V at the load. The OPA4872 configuration shown is a gain of +6 dB; thus, the matching resistance must be selected to achieve –6 dB. The set of equations to solve is shown in Equation 1 and Equation 2. Here, the impedance of interest is 75 Ω. RO = ZO || (RO + RF + RG) R4 Logic 1+ A0 A1 RF RG =2 (1) -5V Figure 30. 4-Input RGB Router (Red Channel Shown) RF + RG = 1046W R F = RG (2) Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 13 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com Solving for RO, with n devices connected together, results in Equation 3: RO = 75 ´ (n - 1) + 804 1+ ´ 2 241200 [75 ´ (n - 1) + 804] 2 -1 (3) Results for n varying from 2 to 6 are given in Table 2. Table 2. Series Resistance vs Number of Parallel Outputs NUMBER OF OPA4872s RO (Ω) 2 69 3 63.94 4 59.49 5 55.59 6 52.15 The two major limitations of this circuit are the device requirements for each OPA4872 and the acceptable return loss resulting from the mismatch between the load and the matching resistor. OPERATING SUGGESTIONS SETTING RESISTOR VALUES TO OPTIMIZE BANDWIDTH The output stage of the OPA4872 is a current-feedback op amp, meaning it can hold an almost constant bandwidth over signal gain settings with the proper adjustment of the external resistor values. This performance is shown in the Typical Characteristic curves; the small-signal bandwidth decreases only slightly with increasing gain. These curves also show that the feedback resistor has been changed for each gain setting. The resistor values on the feedback path can be treated as frequency response compensation elements while the ratio sets the signal gain of the feedback resistor divided by the gain resistor. Figure 32 shows the small-signal frequency response analysis circuit for a current feedback amplifier. VI a VO DESIGN-IN TOOLS RI Z(S) iERR DEMONSTRATION FIXTURE A printed circuit board (PCB) is available to assist in the initial evaluation of circuit performance using the OPA4872. The fixture is offered free of charge as an unpopulated PCB, delivered with a user's guide. The summary information for this fixture is shown in Table 3. Table 3. OPA4872 Demonstration Fixture PACKAGE ORDERING NUMBER LITERATURE NUMBER SO-14 DEM-OPA-SO-1E SBOU045 The demonstration fixture can be requested at the Texas Instruments web site at (www.ti.com) through the OPA4872 product folder. iERR RF RG Figure 32. Recommended Feedback Resistor vs Noise Gain The key elements of this current-feedback op amp model are: α → Buffer gain from the noninverting input to the inverting input RI → Buffer output impedance MACROMODELS AND APPLICATIONS SUPPORT iERR → Feedback error current signal Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This practice is particularly true for video and RF amplifier circuits, where parasitic capacitance and inductance can have a major effect on circuit performance. A SPICE model for the OPA4872 is available through the Texas Instruments web site at www.ti.com. This model does a good job of predicting small-signal ac and transient performance under a wide variety of operating conditions. It does not do as well in predicting the harmonic distortion or dG/dP characteristics. Z(s) → Frequency-dependent transimpedance gain from iERR to VO 14 open-loop The buffer gain is typically very close to 1.00 and is normally neglected from signal gain considerations. It will, however, set the CMRR for a single op amp differential amplifier configuration. For a buffer gain α < 1.0, the CMRR = –20 × log (1 – α) dB. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 VO VI a 1+ = RF RG R RF + RI 1+ F RG 1+ Z(S) = aNG RF + RI NG 1+ Z(S) where: NG = 1+ RF RG (4) This formula is written in a loop-gain analysis format, where the errors arising from a noninfinite open-loop gain are shown in the denominator. If Z(S) were infinite over all frequencies, the denominator of Equation 4 would reduce to 1 and the ideal desired signal gain shown in the numerator would be achieved. The fraction in the denominator of Equation 4 determines the frequency response. Equation 5 shows this as the loop-gain equation: Z(S) = Loop Gain RF + RI NG (5) If 20 × log(RF + NG × RI) were drawn on top of the open-loop transimpedance plot, the difference between the two calculations would be the loop gain at a given frequency. Eventually, Z(S) rolls off to equal the denominator of Equation 5, at which point the loop gain reduces to 1 (and the curves intersect). This point of equality is where the amplifier closed-loop frequency response given by Equation 4 starts to roll off, and is exactly analogous to the frequency at which the noise gain equals the open-loop voltage gain for a voltage-feedback op amp. The difference here is that the total impedance in the denominator of Equation 5 may be controlled somewhat separately from the desired signal gain (or NG). The OPA4872 is internally compensated to give a maximally flat frequency response for RF = 523 Ω at NG = 2 on ±5-V supplies. Evaluating the denominator of Equation 5 (which is the feedback transimpedance) gives an optimal target of 663 Ω. As the signal gain changes, the contribution of the NG × RI term in the feedback transimpedance will change, but the total can be held constant by adjusting RF. Equation 6 gives an approximate equation for optimum RF over signal gain: RF = 663W - NG x RI (6) As the desired signal gain increases, this equation will eventually predict a negative RF. A somewhat subjective limit to this adjustment also can be set by holding RG to a minimum value of 20 Ω. Lower values load both the buffer stage at the input and the output stage, if RF gets too low, actually decreasing the bandwidth. Figure 33 shows the recommended RF versus NG for ±5-V operation. The values for RF versus gain shown here are approximately equal to the values used to generate the Typical Characteristics. They differ in that the optimized values used in the Typical Characteristics are also correcting for board parasitics not considered in the simplified analysis leading to Equation 5. The values shown in Figure 33 give a good starting point for design where bandwidth optimization is desired. 600 550 Feedback Resistor (W) RI, the buffer output impedance, is a critical portion of the bandwidth control equation. RI for the OPA4872 is typically about 30 Ω. A current-feedback op amp senses an error current in the inverting node (as opposed to a differential input error voltage for a voltage-feedback op amp) and passes this on to the output through an internal frequency dependent transimpedance gain. The Typical Characteristics show this open-loop transimpedance response. This open-loop response is analogous to the open-loop voltage gain curve for a voltage-feedback op amp. Developing the transfer function for the circuit of Figure 32 gives Equation 4: 500 450 400 350 300 250 200 150 100 0 5 10 15 20 Noise Gain Figure 33. Feedback Resistor vs Noise Gain The total impedance going into the inverting input may be used to adjust the closed-loop signal bandwidth. Inserting a series resistor between the inverting input and the summing junction increases the feedback impedance (denominator of Equation 4), decreasing the bandwidth. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 15 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com VOSO_envelope = VOS ´ G ± Ibi x RF ± (RS ´ Ib) ´ G DRIVING CAPACITIVE LOADS One of the most demanding, yet very common load conditions, is capacitive loading. Often, the capacitive load is the input of an analog-to-digital converter (ADC)—including additional external capacitance that may be recommended to improve ADC linearity. A high-speed device such as the OPA4872 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the device open-loop output resistance is considered, this capacitive load introduces an additional pole in the signal path that can decrease the phase margin. Several external solutions to this problem have been suggested. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This isolation resistor does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the phase margin and improving stability. The Typical Characteristics show the recommended RS versus capacitive load and the resulting frequency response at the load; see Figure 5. Parasitic capacitive loads greater than 2 pF can begin to degrade the performance of the OPA4872. Long PCB traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always consider this effect carefully, and add the recommended series resistor as close as possible to the OPA4872 output pin (see the Board Layout Guidelines section). DC ACCURACY The OPA4872 offers excellent dc signal accuracy. Parameters that influence the output dc offset voltage are: • Output offset voltage • Input bias current • Gain error • Power-supply rejection ratio • Temperature Leaving both temperature and gain error parameters aside, the output offset voltage envelope can be described as shown in Equation 7: 16 ± ½5 - (VS+)½ ´ 10 - ±½-5 - (VS-)½ ´ 10- PSRR+ 20 PSRR20 (7) Where: RS: Input resistance seen by R0, R1, G0, G1, B0, or B1. Ib: Noninverting input bias current Ibi: Inverting input bias current G: Gain VS+: Positive supply voltage VS–: Negative supply voltage PSRR+: Positive supply PSRR PSRR–: Negative supply PSRR VOS: Input Offset Voltage Evaluating the front-page schematic, using a worst-case, +25°C offset voltage, bias current and PSRR specifications and operating at ±6 V, gives a worst-case output equal to Equation 8: ±10mV + 75W ´ ±14mA ´ 2 50 +523W ´ ±18mA ±½5 - 6½ ´ 10 20 ±½-5 - (-6)½ ´ 10 - 51 20 = ±29.2mV (8) DISTORTION PERFORMANCE The OPA4872 provides good distortion performance into a 150-Ω load on ±5-V supplies. Relative to alternative solutions, it provides exceptional performance into lighter loads. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd harmonic dominates the distortion with a negligible 3rd harmonic component. Focusing then on the 2nd harmonic, increasing the load impedance directly improves distortion. Also, providing an additional supply decoupling capacitor (0.01 µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3 dB to 6 dB). In most op amps, increasing the output voltage swing increases harmonic distortion directly. The Typical Characteristics show the 2nd harmonic increasing at a little less than the expected 2X rate while the 3rd harmonic increases at a little less than the expected 3X rate. Where the test power doubles, the 2nd harmonic increases only by less than the expected 6 dB, whereas the 3rd harmonic increases by less than the expected 12 dB. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 NOISE PERFORMANCE The OPA4872 offers an excellent balance between voltage and current noise terms to achieve low output noise. The inverting current noise (19 pA/√Hz) is significantly lower than earlier solutions, while the input voltage noise (4.5 nV/√Hz) is lower than most unity-gain stable, wideband, voltage-feedback op amps. As long as the ac source impedance looking out of the noninverting node is less than 100 Ω, this current noise will not contribute significantly to the total output noise. The op amp input voltage noise and the two input current noise terms combine to give low output noise under a wide variety of operating conditions. Figure 34 shows the OPA4872 noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. ENI EO OPA4872 RS IBN ERS Ö( 2 2 ) 2 2 ENI + (IBNRS) + 4kTRS NG + (IBIRF) + 4kTRFNG (9) Dividing this expression by the noise gain (NG = (1 + RF/RG)) gives the equivalent input-referred spot noise voltage at the noninverting input, as shown in Equation 10. EO = Ö 2 2 ENI + (IBNRS) + 4kTRS + 2 ( INGR ) + 4kTR NG BI F F (10) Evaluating these two equations for the OPA4872 circuit and component values (see Figure 27) gives a total output spot noise voltage of 14.2 nV/√Hz and a total equivalent input spot noise voltage of 7.1 nV/√Hz. This total input-referred spot noise voltage is higher than the 4.5-nV/√Hz specification for the OPA4872 voltage noise alone. This voltage reflects the noise added to the output by the inverting current noise times the feedback resistor. If the feedback resistor is reduced in high-gain configurations, the total input-referred voltage noise given by Equation 10 approaches only the 4.5 nV/√Hz of the op amp itself. For example, going to a gain of +10 using RF = 178 Ω gives a total input-referred noise of 4.7 nV/√Hz. RF Ö4kTRS 4kT RG EO = RG IBI Ö4kTRF 4kT = 1.6 x 10 at 290K - 20 J Figure 34. Op Amp Noise Analysis Model The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 9 shows the general form for the output noise voltage using the terms shown in Figure 35. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 17 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com THERMAL ANALYSIS Heatsinking or forced airflow may be required under extreme operating conditions. Maximum desired junction temperature sets the maximum allowed internal power dissipation as discussed in this document. In no case should the maximum junction temperature be allowed to exceed +150°C. Operating junction temperature (TJ) is given by TA + PD × θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver en RS load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL depends on the required output signal and load; for a grounded resistive load, PDL is at a maximum when the output is fixed at a voltage equal to 1/2 of either supply voltage (for equal bipolar supplies). Under this condition PDL = VS2/(4 × RL), where RL includes feedback network loading. Note that it is the power in the output stage and not in the load that determines internal power dissipation. -20 OPA4872 4kT = 1.6 x 10 at 290K J ini VRS = Ö4kTRS eo RF VRF = Ö4kTRF iin RG iRG Ö 4kT RG Figure 35. OPA4872 Noise Analysis Model 18 Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP OPA4872-EP www.ti.com........................................................................................................................................................................................... SBOS444 – DECEMBER 2008 As a worst-case example, compute the maximum TJ using an OPA4872ID in the circuit of Figure 27 operating at the maximum specified ambient temperature of +85°C with its output driving a grounded 100-Ω load to +2.5 V: PD = 10V ´ 11.7mA + (5 /[4 ´ (150W || 1046W)]) = 165mW 2 Maximum TJ = +85°C + (165mW ´ 80°C/W) = 98°C This worst-case condition does not exceed the maximum junction temperature. Normally, this extreme case is not encountered. BOARD LAYOUT GUIDELINES Achieving optimum performance with a high-frequency amplifier such as the OPA4872 requires careful attention to board layout parasitics and external component types. Recommendations to optimize performance include: a) Minimize parasitic capacitance to any ac ground for all of the signal I/O pins. Parasitic capacitance on the output pin can cause instability; on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25") from the power-supply pins to high frequency 0.1-µF decoupling capacitors. At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections (on pins 9, 11, 13, and 15) should always be decoupled with these capacitors. An optional supply decoupling capacitor across the two power supplies (for bipolar operation) improves 2nd harmonic distortion performance. Larger (2.2 µF to 6.8 µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These capacitors may be placed somewhat farther from the device and may be shared among several devices in the same area of the PCB. c) Careful selection and placement of external components preserves the high-frequency performance of the OPA4872. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal-film and carbon composition, axially-leaded resistors can also provide good high-frequency performance. Again, keep their leads and PCB trace length as short as possible. Never use wirewound type resistors in a high-frequency application. Other network components, such as noninverting input termination resistors, should also be placed close to the package. d) Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of Figure 5. Low parasitic capacitive loads (greater than 5 pF) may not need an RS because the OPA4872 is nominally compensated to operate with a 2-pF parasitic load. If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50-Ω environment is normally not necessary on the board, and in fact, a higher impedance environment improves distortion as shown in the Distortion versus Load plot; see Figure 7. With a characteristic board trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the output of the OPA4872 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance is the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. The high output voltage and current capability of the OPA4872 allow multiple destination devices to be handled as separate transmission lines, each with its own series and shunt terminations. If the 6-dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in Figure 5. This configuration does not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation because of the voltage divider formed by the series output into the terminating impedance. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP 19 OPA4872-EP SBOS444 – DECEMBER 2008........................................................................................................................................................................................... www.ti.com e) Socketing a high-speed part like the OPA4872 is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network that can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA4872 onto the board. +VCC External Pin Internal Circuitry -VCC INPUT AND ESD PROTECTION The OPA4872 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins have limited ESD protection using internal diodes to the power supplies as shown in Figure 36. 20 Figure 36. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30-mA continuous current. Where higher currents are possible (for example, in systems with ±15-V supply parts driving into the OPA4872), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible because high values degrade both noise performance and frequency response. Submit Documentation Feedback Copyright © 2008, Texas Instruments Incorporated Product Folder Link(s): OPA4872-EP PACKAGE MATERIALS INFORMATION www.ti.com 16-Dec-2008 TAPE AND REEL INFORMATION *All dimensions are nominal Device OPA4872MDREP Package Package Pins Type Drawing SOIC D 14 SPQ Reel Reel Diameter Width (mm) W1 (mm) 2500 330.0 16.4 Pack Materials-Page 1 A0 (mm) B0 (mm) K0 (mm) P1 (mm) 6.5 9.0 2.1 8.0 W Pin1 (mm) Quadrant 16.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 16-Dec-2008 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) OPA4872MDREP SOIC D 14 2500 346.0 346.0 33.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. 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