AD ADF4150HVBCPZ

High Voltage, Fractional-N/
Integer-N PLL Synthesizer
ADF4150HV
FEATURES
GENERAL DESCRIPTION
Fractional-N synthesizer and integer-N synthesizer
High voltage charge pump: VP = 6 V to 30 V
Tuning range: 1.0 V to 29 V (or ±1 V from VP supply rails)
RF bandwidth to 3.0 GHz
Programmable divide-by-1/-2/-4/-8/-16 outputs
Synthesizer power supply: 3.0 V to 3.6 V
Programmable dual-modulus prescaler of 4/5 or 8/9
Programmable output power level
Programmable charge pump currents
RF output mute function
3-wire serial interface
Analog and digital lock detect
The ADF4150HV is a 3.0 GHz, fractional-N or integer-N
frequency synthesizer with an integrated high voltage charge
pump. The synthesizer can be used to drive external wideband
VCOs directly, eliminating the need for operational amplifiers
to achieve higher tuning voltages. This simplifies design and
reduces cost while improving phase noise, in contrast to active
filter topologies, which tend to degrade phase noise compared
to passive filter topologies.
APPLICATIONS
A simple 3-wire interface controls all on-chip registers. The
charge pump operates from a power supply ranging from 6 V to
30 V, whereas the rest of the device operates from 3.0 V to 3.6 V.
The ADF4150HV can be powered down when not in use.
The VCO frequency can be divided by 1, 2, 4, 8, or 16 to allow
the user to generate RF output frequencies as low as 31.25 MHz.
For applications that require isolation, the RF output stage can be
muted. The mute function is both pin- and software-controllable.
Wireless infrastructure
Microwave point-to-point/point-to-multipoint radios
VSAT radios
Test equipment
Private land mobile radios
FUNCTIONAL BLOCK DIAGRAM
SDVDD
REFIN
CLK
DATA
LE
×2
DOUBLER
AVDD
10-BIT R
COUNTER
DVDD
VP
RSET
MULTIPLEXER
÷2
DIVIDER
LOCK
DETECT
DATA REGISTER
LD
HIGH VOLTAGE
CHARGE
PUMP
FUNCTION
LATCH
PHASE
COMPARATOR
CURRENT
SETTING
INTEGER
VALUE
MUXOUT
FRACTION
VALUE
DIVIDE-BY-1/
-2/-4/-8/-16
MODULUS
VALUE
CPOUT
BOOST
MODE
OUTPUT
STAGE
RFOUT+
RFOUT–
PDBRF
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
RF
INPUT
MULTIPLEXER
RFIN+
RFIN–
ADF4150HV
CE
GND
CPGND
SDGND
09058-001
N COUNTER
Figure 1.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
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Fax: 781.461.3113
©2011 Analog Devices, Inc. All rights reserved.
ADF4150HV
TABLE OF CONTENTS
Features .............................................................................................. 1 Register 1 ..................................................................................... 17 Applications ....................................................................................... 1 Register 2 ..................................................................................... 17 General Description ......................................................................... 1 Register 3 ..................................................................................... 19 Functional Block Diagram .............................................................. 1 Register 4 ..................................................................................... 19 Revision History ............................................................................... 2 Register 5 ..................................................................................... 19 Specifications..................................................................................... 3 Register Initialization Sequence ............................................... 19 Timing Characteristics ................................................................ 5 RF Synthesizer—A Worked Example ...................................... 20 Absolute Maximum Ratings............................................................ 6 Reference Doubler and Reference Divider ............................. 20 Transistor Count ........................................................................... 6 12-Bit Programmable Modulus ................................................ 20 Thermal Resistance ...................................................................... 6 Spurious Optimization and Boost Mode ................................ 21 ESD Caution .................................................................................. 6 Spur Mechanisms ....................................................................... 21 Pin Configuration and Function Descriptions ............................. 7 Spur Consistency and Fractional Spur Optimization ........... 21 Typical Performance Characteristics ............................................. 9 Phase Resync ............................................................................... 22 Circuit Description ......................................................................... 11 Applications Information .............................................................. 23 Reference Input Section ............................................................. 11 Ultrawideband PLL .................................................................... 23 RF N Divider ............................................................................... 11 Microwave PLL ........................................................................... 23 Phase Frequency Detector (PFD) and High Voltage
Charge Pump .............................................................................. 11 Generating the High Voltage Supply ....................................... 24 MUXOUT and Lock Detect ...................................................... 12 PCB Design Guidelines for a Chip Scale Package ................. 25 Input Shift Registers ................................................................... 12 Output Matching ........................................................................ 26 Program Modes .......................................................................... 12 Outline Dimensions ....................................................................... 27 Output Stage ................................................................................ 12 Ordering Guide .......................................................................... 27 Interfacing to the ADuC702x and the ADSP-BF527 ............. 25 Register Maps .................................................................................. 13 Register 0 ..................................................................................... 17 REVISION HISTORY
8/11—Revision 0: Initial Version
Rev. 0 | Page 2 of 28
ADF4150HV
SPECIFICATIONS
AVDD = DVDD = SDVDD = 3.3 V ± 10%; VP = 6.0 V to 30 V; GND = 0 V; TA = TMIN to TMAX, unless otherwise noted. Operating temperature
range is −40°C to +85°C.
Table 1.
Parameter
REFIN CHARACTERISTICS
Input Frequency
Max
Unit
Test Conditions/Comments
10
10
300
30
MHz
MHz
Input Sensitivity
Input Capacitance
Input Current
RF INPUT CHARACTERISTICS
0.7
AVDD
5.0
±60
V p-p
pF
μA
For f < 10 MHz, ensure slew rate > 21 V/μs
Reference doubler enabled (DB25 bit in
Register 2 is set to 1)
Biased at AVDD/2; ac coupling ensures AVDD/2 bias
RF Input Frequency (RFIN)
Prescaler Output Frequency
PHASE DETECTOR
Phase Detector Frequency
0.5
HIGH VOLTAGE CHARGE PUMP
ICP Sink/Source
High Value
Low Value
RSET Range
High Value vs. RSET
Min
Typ
3.0
750
GHz
MHz
26
20
26
MHz
MHz
MHz
Low noise mode
Low spur mode
Integer-N mode
μA
μA
kΩ
μA
μA
%
%
%
%
nA
RSET = 5.1 kΩ
RSET = 5.1 kΩ
384
48
3.3
196
10
594
Sink and Source Current Matching
Absolute ICP Accuracy
ICP vs. VCP
ICP vs. Temperature
ICP Leakage
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
LOGIC OUTPUTS
Output High Voltage, VOH
Output High Current, IOH
Output Low Voltage, VOL
POWER SUPPLIES
AVDD
DVDD, SDVDD
VP
IP
DIDD + AIDD1
Current per Output Divider
IRFOUT2
Low Power Sleep Mode
6
3
2.5
2.5
2.5
2.0
0.6
±1
15.0
V
V
μA
pF
500
0.4
V
μA
V
DVDD − 0.4
3.0
3.6
AVDD
6.0
30
1
50
6 to 24
20
1
For lower RFIN frequencies, ensure slew
rate > 400 V/μs
−10 dBm ≤ RF input power ≤ +5 dBm
2.5
60
32
Rev. 0 | Page 3 of 28
V
V
V
mA
mA
mA
mA
μA
RSET = 10 kΩ
RSET = 3.3 kΩ
1.0 V ≤ VCP ≤ (VP − 1.0 V); VP = 6 V to 30 V
1.0 V ≤ VCP ≤ (VP − 1.0 V)
VCP = VP/2
VCP = VP/2
CMOS output selected
IOL = 500 μA
Set the VP supply at least 1 V above the
maximum desired tuning voltage
VP = 30 V
Each output divide-by-2 consumes 6 mA typ
RF output stage is programmable
ADF4150HV
Parameter
RF OUTPUT CHARACTERISTICS
Output Frequency Using RF Output
Dividers
Harmonic Content (Second)
Harmonic Content (Third)
Minimum RF Output Power 2
Maximum RF Output Power2
Output Power Variation vs. Supply
Output Power Variation vs. Temperature
Level of Signal with RF Mute Enabled
NOISE CHARACTERISTICS
Normalized In-Band Phase Noise
Floor (PNSYNTH) 3
Normalized 1/f Noise (PN1_f) 4
RF Output Divider Noise Floor
Spurious Signals Due to PFD Frequency
Min
Typ
Max
Unit
Test Conditions/Comments
MHz
500 MHz VCO input and divide-by-16 selected
−19
−20
−13
−10
−4
5
±1
dBc
dBc
dBc
dBc
dBm
dBm
dB
±1
−37
dB
dBm
Fundamental VCO output
Divided VCO output
Fundamental VCO output
Divided VCO output
Programmable in 3 dB steps
Programmable in 3 dB steps
Pull-up supply on Pin 18 and Pin 19 varied
from 3.0 V to 3.6 V
From −40°C to +85°C
PDBRF pin brought low; RF = 2 GHz
−213
dBc/Hz
Low noise mode
−203
−113
−108
−155
−70
−85
dBc/Hz
dBc/Hz
dBc/Hz
dBc/Hz
dBc
dBc
Low spur mode
Low noise mode
Low spur mode
Measured at 10 MHz offset
At RFOUT+/RFOUT− pins
At VCO output
31.25
1
TA = 25°C; AVDD = DVDD = 3.3 V; prescaler = 8/9; fREFIN = 100 MHz; fPFD = 25 MHz; fRF = 1.75 GHz.
Using 50 Ω resistors to AVDD, into a 50 Ω load.
3
This figure can be used to calculate phase noise for any application. To calculate in-band phase noise performance as seen at the VCO output, use the following formula:
PNSYNTH = PNTOT − 10 log(fPFD) − 20 log N.
4
The PLL phase noise is composed of flicker (1/f) noise plus the normalized PLL noise floor. The flicker noise is specified at a 10 kHz offset and normalized to 1 GHz. The
formula for calculating the 1/f noise contribution at an RF frequency (fRF) and at a frequency offset (f) is given by PN = PN1_f + 10 log(10 kHz/f) + 20 log(fRF/1 GHz). Both
the normalized phase noise floor and flicker noise are modeled in ADIsimPLL.
2
Rev. 0 | Page 4 of 28
ADF4150HV
TIMING CHARACTERISTICS
AVDD = DVDD = SDVDD = 3.3 V ± 10%; VP = 6.0 V to 30 V; GND = 0 V; TA = TMIN to TMAX, unless otherwise noted. Operating temperature
range is −40°C to +85°C.
Table 2.
Parameter
t1
t2
t3
t4
t5
t6
t7
Limit
20
10
10
25
25
10
20
Unit
ns min
ns min
ns min
ns min
ns min
ns min
ns min
Description
LE setup time
DATA to CLK setup time
DATA to CLK hold time
CLK high duration
CLK low duration
CLK to LE setup time
LE pulse width
Timing Diagram
t4
t5
CLK
t2
DATA
DB31 (MSB)
t3
DB30
DB2
(CONTROL BIT C3)
DB1
(CONTROL BIT C2)
DB0 (LSB)
(CONTROL BIT C1)
t7
LE
t1
09058-002
t6
LE
Figure 2. Timing Diagram
Rev. 0 | Page 5 of 28
ADF4150HV
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
TRANSISTOR COUNT
Table 3.
The transistor count for the ADF4150HV is 23,380 (CMOS)
and 809 (bipolar).
Parameter
AVDD to GND1
AVDD to DVDD
VP to GND1
Digital I/O Voltage to GND1
Analog I/O Voltage to GND1
REFIN to GND1
Operating Temperature Range
Storage Temperature Range
Maximum Junction Temperature
Reflow Soldering
Peak Temperature
Time at Peak Temperature
1
Rating
−0.3 V to +3.9 V
−0.3 V to +0.3 V
−0.3 V to +33 V
−0.3 V to AVDD + 0.3 V
−0.3 V to DVDD + 0.3 V
−0.3 V to AVDD + 0.3 V
−40°C to +85°C
−65°C to +125°C
150°C
THERMAL RESISTANCE
Thermal impedance (θJA) is specified for a device with the
exposed pad soldered to GND.
Table 4. Thermal Resistance
Package Type
32-Lead LFCSP (CP-32-11)
ESD CAUTION
260°C
40 sec
GND = CPGND = SDGND = 0 V.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rev. 0 | Page 6 of 28
θJA
27.3
Unit
°C/W
ADF4150HV
32
31
30
29
28
27
26
25
GND
RSET
GND
SDGND
SDVDD
MUXOUT
LD
REFIN
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
1
2
3
4
5
6
7
8
ADF4150HV
TOP VIEW
(Not to Scale)
24
23
22
21
20
19
18
17
GND
GND
DVDD
PDBRF
AVDD
RFOUT +
RFOUT –
GND
NOTES
1. THE LFCSP HAS AN EXPOSED PAD
THAT MUST BE CONNECTED TO GND.
09058-003
CPOUT
CPGND
AVDD
GND
AVDD
RFIN+
RFIN–
GND
9
10
11
12
13
14
15
16
GND
CLK
DATA
LE
CE
VP
GND
GND
Figure 3. Pin Configuration
Table 5. Pin Function Descriptions
Pin No.
Mnemonic
1, 7, 8, 12, 16, 17, GND
23, 24, 30, 32
2
CLK
3
DATA
4
LE
5
CE
6
VP
9
CPOUT
10
11, 13, 20
CPGND
AVDD
14
15
RFIN+
RFIN−
18
RFOUT−
19
RFOUT+
21
22
PDBRF
DVDD
25
REFIN
26
LD
Description
Ground. All ground pins should be tied together.
Serial Clock Input. Data is clocked into the 32-bit shift register on the CLK rising edge. This input is a high
impedance CMOS input.
Serial Data Input. The serial data is loaded MSB first with the three LSBs as the control bits. This input is a
high impedance CMOS input.
Load Enable. When LE goes high, the data stored in the 32-bit shift register is loaded into the register
that is selected by the three control bits. This input is a high impedance CMOS input.
Chip Enable. A logic low on this pin powers down the device and puts the charge pump into three-state
mode. A logic high on this pin powers up the device.
High Voltage Charge Pump Power Supply. Place decoupling capacitors to the ground plane as close to
this pin as possible. The decoupling capacitors should have the appropriate voltage rating (a value of
10 μF is recommended). Care should be taken to ensure that VP does not exceed the absolute maximum
ratings on power-up (see Table 3). A 10 Ω series resistor can help to significantly reduce voltage overshoot
with minimal IR drop.
High Voltage Charge Pump Output. When enabled, this output provides ±ICP to the external passive loop
filter. The output of the loop filter is connected to the voltage tuning port of the external VCO.
High Voltage Charge Pump Ground. All ground pins should be tied together.
Analog Power Supply. This pin ranges from 3.0 V to 3.6 V. Place decoupling capacitors to the ground
plane as close to this pin as possible. AVDD must have the same value as DVDD.
Positive RF Input. The output of the VCO or external prescaler should be ac-coupled to this pin.
Complementary RF Input. If a single-ended input is required, this pin can be tied to ground via a 100 pF
capacitor.
Divided-Down Output of RFIN−. This pin can be left unconnected if the divider functionality is not
required.
Divided-Down Output of RFIN+. This pin can be left unconnected if the divider functionality is not
required.
RF Power-Down. A logic low on this pin mutes the RF outputs. This function is also software controllable.
Digital Power Supply. Place decoupling capacitors to the ground plane as close to this pin as possible.
DVDD must have the same value as AVDD.
Reference Input. This CMOS input has a nominal threshold of AVDD/2 and a dc equivalent input resistance
of 100 kΩ. This input can be driven from a crystal oscillator, TCXO, or other reference.
Lock Detect Output. A logic high output on this pin indicates PLL lock. A logic low output indicates loss
of PLL lock.
Rev. 0 | Page 7 of 28
ADF4150HV
Pin No.
27
Mnemonic
MUXOUT
28
SDVDD
29
31
SDGND
RSET
EP
Exposed Pad
Description
Multiplexer Output. The multiplexer output allows the lock detect, the N divider value, or the R counter
value to be accessed externally.
Digital Σ-Δ Modulator Power Supply. Place decoupling capacitors to the ground plane as close to this
pin as possible. SDVDD must have the same value as AVDD.
Digital Σ-Δ Modulator Ground. All ground pins should be tied together.
Connecting a resistor between this pin and GND sets the charge pump output current. Place the resistor
as close to this pin as possible. The nominal voltage bias at the RSET pin is 0.55 V. The relationship between
ICP and RSET is as follows:
ICP = 1.96/RSET
where:
RSET = 5.1 kΩ.
ICP = 384 μA.
Exposed Pad. The LFCSP has an exposed pad that must be connected to GND.
Rev. 0 | Page 8 of 28
ADF4150HV
ICP MISMATCH (%)
ICP = 400µA SOURCE
ICP = 350µA SOURCE
ICP = 300µA SOURCE
ICP = 250µA SOURCE
ICP = 200µA SOURCE
ICP = 150µA SOURCE
ICP = 100µA SOURCE
ICP = 50µA SOURCE
ICP = 50µA SINK
ICP = 100µA SINK
ICP = 150µA SINK
ICP = 200µA SINK
ICP = 250µA SINK
ICP = 300µA SINK
ICP = 350µA SINK
ICP = 400µA SINK
2
4
6
8
10
12
14
16
18
20
22
24
26
28
VCP (V)
0
6
10
12
200kHz
400kHz
–50
SPUR LEVEL (dBc)
–60
14
16
18
20
22
24
26
28
600kHz
800kHz
BEAT NOTE
SPUR
BEAT NOTE
SPUR
–70
–80
–90
–100
–110
–120
1k
10k
100k
1M
10M
100M
–130
1500
1505
1510
1515
1520
1525
FREQUENCY (MHz)
Figure 8. Fractional Spur Levels vs. Frequency, Low Spur Mode;
Measured at VCO Output, PFD = 25 MHz, MOD = 125
Figure 5. Active Filter Phase Noise, ADF4150HV vs. ADF4156;
Active Filter Implemented Using OP27 Op Amp; PFD = 20 MHz, Loop
Bandwidth = 10 kHz, ICP = 300 μA, Carrier Frequency = 1.7 GHz, VP = 28 V
–40
2.0
1.9
8
–40
ADF4150HV
RMS NOISE = 0.28°
ADF4156
RMS NOISE = 0.36°
09058-005
200kHz
400kHz
–50
BOOST MODE ON
1.8
600kHz
800kHz
–60
SPUR LEVEL (dBc)
1.7
BOOST MODE OFF
1.6
1.5
1.4
1.3
–70
–80
–90
–100
–110
1.1
–120
1.0
0
50
100
150
TIME (µs)
200
250
300
09058-006
1.2
Figure 6. PLL Lock Time with Boost Mode On and Off;
Locking over Octave Range Jump (1 GHz to 2 GHz) for PFD = 20 MHz,
Loop Bandwidth = 100 kHz, ICP = 300 μA, VP = 28 V, VDD = 3.3 V, REFIN = 100 MHz
Rev. 0 | Page 9 of 28
–130
1500
1505
1510
1515
1520
1525
FREQUENCY (MHz)
Figure 9. Fractional Spur Levels vs. Frequency, Low Noise Mode;
Measured at VCO Output, PFD = 25 MHz, MOD = 125
09058-009
PHASE NOISE (dBc/Hz)
4
Figure 7. Charge Pump Output Mismatch vs. VP, ICP = 200 μA
FREQUENCY (Hz)
FREQUENCY (GHz)
2
VCP (V)
Figure 4. Charge Pump Output Characteristics, VP = 28 V,
ICP Varied from 50 μA to 400 μA, RSET = 5.1 kΩ
–80
–85
–90
–95
–100
–105
–110
–115
–120
–125
–130
–135
–140
–145
–150
–155
–160
–165
–170
100
VP = 6V MISMATCH (%)
VP = 9V MISMATCH (%)
VP = 12V MISMATCH (%)
VP = 15V MISMATCH (%)
VP = 18V MISMATCH (%)
VP = 21V MISMATCH (%)
VP = 24V MISMATCH (%)
VP = 28V MISMATCH (%)
09058-008
0
16
14
12
10
8
6
4
2
0
–2
–4
–6
–8
–10
–12
–14
–16
09058-007
600
550
500
450
400
350
300
250
200
150
100
50
0
–50
–100
–150
–200
–250
–300
–350
–400
–450
–500
09058-004
ICP (µA)
TYPICAL PERFORMANCE CHARACTERISTICS
ADF4150HV
–40
25MHz
50MHz
–50
–80
75MHz
100MHz
–85
PHASE NOISE (dBc/Hz)
–70
–80
–90
–100
–95
–100
LOW NOISE MODE
–105
–110
1200
1400
1600
1800
2000
FREQUENCY (MHz)
Figure 10. PFD and Reference Spur Levels vs. Frequency at VCO Output,
REFIN = 100 MHz, PFD = 25 MHz
–40
25MHz
50MHz
–50
–110
1000
09058-110
–120
1000
LOW SPUR MODE
–90
1050
1100
1150
1200
1250
1300
FREQUENCY (MHz)
09058-112
SPUR LEVEL (dBc)
–60
Figure 12. In-Band Phase Noise Measured at 3 kHz Offset for Low Noise Mode
and Low Spur Mode, PFD = 25 MHz, PLL Loop Bandwidth = 40 kHz
4
75MHz
100MHz
2
+5dBm
0
OUTPUT POWER (dBm)
SPUR LEVEL (dBc)
–60
–70
–80
–90
–100
+2dBm
–2
–4
–1dBm
–6
–4dBm
–8
–10
–12
–110
1400
1600
FREQUENCY (MHz)
1800
2000
Figure 11. PFD and Reference Spur Levels vs. Frequency at VCO Output
with ADL5541 Buffer Placed Between VCO Output and RF Input,
REFIN = 100 MHz, PFD = 25 MHz
Rev. 0 | Page 10 of 28
–16
0
500
1000
1500
2000
2500
FREQUENCY (MHz)
Figure 13. Single-Ended RF Output Power Level vs. Frequency and
Power Setting, RF Output Pins Pulled Up to 3.3 V via 27 nH||50 Ω
09058-113
1200
09058-111
–120
1000
–14
ADF4150HV
CIRCUIT DESCRIPTION
The PFD frequency (fPFD) equation is
REFERENCE INPUT SECTION
The reference input stage is shown in Figure 14. The SW1 and
SW2 switches are normally closed. The SW3 switch is normally
open. When power-down is initiated, SW3 is closed, and SW1
and SW2 are opened. In this way, no loading of the REFIN pin
occurs during power-down.
POWER-DOWN
CONTROL
NC
100kΩ
TO R COUNTER
09058-010
SW3
NO
R Counter
Figure 14. Reference Input Stage
RF N DIVIDER
The RF N divider allows a division ratio in the PLL feedback
path. The division ratio is determined by the INT, FRAC, and
MOD values, which build up this divider (see Figure 15).
RF N DIVIDER
FROM
VCO OUTPUT/
OUTPUT DIVIDERS
where:
REFIN is the reference input frequency.
D is the REFIN doubler bit.
R is the preset divide ratio of the binary 10-bit programmable
reference counter (1 to 1023).
T is the REFIN divide-by-2 bit (0 or 1).
If FRAC = 0 and the DB8 (LDF) bit in Register 2 is set to 1,
the synthesizer operates in integer-N mode. The DB8 bit in
Register 2 should be set to 1 for integer-N digital lock detect.
BUFFER
SW1
N = INT + FRAC/MOD
TO PFD
N COUNTER
THIRD-ORDER
FRACTIONAL
INTERPOLATOR
The 10-bit R counter allows the input reference frequency
(REFIN) to be divided down to produce the reference clock
to the PFD. Division ratios from 1 to 1023 are allowed.
PHASE FREQUENCY DETECTOR (PFD) AND HIGH
VOLTAGE CHARGE PUMP
The phase frequency detector (PFD) takes inputs from the
R counter and N counter and produces an output proportional
to the phase and frequency difference between them. Figure 16
is a simplified schematic of the phase frequency detector.
HIGH
MOD
VALUE
D1
Q1
UP
U1
FRAC
VALUE
+IN
CLR1
09058-011
INT
VALUE
(2)
Integer-N Mode
SW2
REFIN NC
fPFD = REFIN × [(1 + D)/(R × (1 + T))]
DELAY
Figure 15. RF N Divider
U3
CHARGE
PUMP
CPOUT
INT, FRAC, MOD, and R Counter Relationship
HIGH
U2
–IN
Figure 16. PFD Simplified Schematic
The RF VCO frequency (RFOUT) equation is
RFOUT = (fPFD/RF Divider) × [INT + (FRAC/MOD)]
CLR2
DOWN
D2
Q2
09058-012
The INT, FRAC, and MOD values, in conjunction with the
R counter, make it possible to generate output frequencies that
are spaced by fractions of the PFD frequency. For more information, see the RF Synthesizer—A Worked Example section.
(1)
where:
RFOUT is the output frequency of the external voltage controlled
oscillator (VCO).
RF Divider is the output divider that divides down the VCO
frequency.
INT is the preset divide ratio of the binary 16-bit counter (23 to
32,767 for the 4/5 prescaler, 75 to 65,535 for the 8/9 prescaler).
FRAC is the numerator of the fractional division (0 to MOD − 1).
MOD is the preset fractional modulus (2 to 4095).
The PFD includes a delay element that sets the width of the
antibacklash pulse to 4.2 ns. This pulse ensures that there is
no dead zone in the PFD transfer function and provides a
consistent reference spur level.
The high voltage charge pump is designed on an Analog
Devices, Inc., proprietary high voltage process and allows the
charge pump to output voltages as high as 29 V when powered
by a 30 V supply. The high voltage charge pump removes the
need for active filtering when interfacing to a high voltage VCO.
Rev. 0 | Page 11 of 28
ADF4150HV
MUXOUT AND LOCK DETECT
PROGRAM MODES
The multiplexer output on the ADF4150HV allows the user to
access various internal points on the chip. The state of MUXOUT is
controlled by the M3, M2, and M1 bits in Register 2 (see Figure 22).
Figure 17 shows the MUXOUT section in block diagram form.
Table 6 and Figure 19 through Figure 25 show how the program
modes are set up in the ADF4150HV.
R COUNTER INPUT
DVDD
THREE-STATE-OUTPUT
DVDD
1.
GND
R COUNTER OUTPUT
The following settings in the ADF4150HV are double buffered:
phase value, modulus value, reference doubler, reference divideby-2, R counter value, and charge pump current setting. Before
the part uses a new value for any double-buffered setting, the
following two events must occur:
MUX
MUXOUT
CONTROL
2.
N COUNTER OUTPUT
ANALOG LOCK DETECT
DIGITAL LOCK DETECT
GND
09058-013
RESERVED
Figure 17. MUXOUT Schematic
The new value is latched into the device by writing to the
appropriate register.
A new write is performed on Register 0 (R0).
For example, any time that the modulus value is updated,
Register 0 (R0) must be written to, to ensure that the modulus
value is loaded correctly. The divider select value in Register 4
(R4) is also double buffered, but only if the DB13 bit of
Register 2 (R2) is high.
INPUT SHIFT REGISTERS
OUTPUT STAGE
The ADF4150HV digital section includes a 10-bit RF R counter,
a 16-bit RF N counter, a 12-bit FRAC counter, and a 12-bit
modulus counter. Data is clocked into the 32-bit shift register
on each rising edge of CLK. The data is clocked in MSB first.
Data is transferred from the shift register to one of six latches
on the rising edge of LE. The destination latch is determined by
the state of the three control bits (C3, C2, and C1) in the shift
register. As shown in Figure 2, the control bits are the three LSBs:
DB2, DB1, and DB0. The truth table for these bits is shown in
Table 6. Figure 19 summarizes how the latches are programmed.
The RFOUT+ and RFOUT− pins of the ADF4150HV are connected
to the collectors of an NPN differential pair driven by buffered
outputs of the VCO, as shown in Figure 18. To allow the user to
optimize the power dissipation vs. output power requirements,
the tail current of the differential pair is programmable using
Bits[DB4:DB3] in Register 4 (R4). Four current levels can be set.
These levels give output power levels of −4 dBm, −1 dBm, +2 dBm,
and +5 dBm, respectively, using a 50 Ω resistor to AVDD and ac
coupling into a 50 Ω load. Alternatively, both outputs can be
combined in a 1 + 1:1 transformer or a 180° microstrip coupler
(see the Output Matching section). If the outputs are used
individually, the optimum output stage consists of a shunt
inductor to AVDD.
Table 6. Truth Table for C3, C2, and C1 Control Bits
Control Bits
C2
0
0
1
1
0
0
C1
0
1
0
1
0
1
Register
Register 0 (R0)
Register 1 (R1)
Register 2 (R2)
Register 3 (R3)
Register 4 (R4)
Register 5 (R5)
RFOUT+
VCO
RFOUT –
BUFFER/
DIVIDE-BY-1/-2/-4/-8/-16
09058-014
C3
0
0
0
0
1
1
Figure 18. Output Stage
Another feature of the ADF4150HV is that the supply current to
the RF output stage can be shut down until the part achieves lock,
as measured by the digital lock detect circuitry. This feature is
enabled by the mute-till-lock detect (MTLD) bit in Register 4 (R4).
Rev. 0 | Page 12 of 28
ADF4150HV
REGISTER MAPS
RESERVED
REGISTER 0
16-BIT INTEGER VALUE (INT)
CONTROL
BITS
12-BIT FRACTIONAL VALUE (FRAC)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
N16
N15
N14
N13
N12
N11
N10
N9
N8
N7
N6
N5
N4
N3
N2
N1
F12
F11
F10
F9
F8
F7
F6
F5
F4
F3
F2
F1
DB2
DB1
DB0
C3(0) C2(0) C1(0)
PRESCALER
REGISTER 1
RESERVED
DBR1
12-BIT PHASE VALUE (PHASE)
CONTROL
BITS
DBR 1
12-BIT MODULUS VALUE (MOD)
P10
P9
P8
P7
P6
P5
P4
P3
P2
P1
M12
M11
M10
M9
M8
M7
M6
M5
M4
M3
M2
M1
C3(0) C2(0) C1(1)
COUNTER
RESET
P11
CP THREESTATE
P12
POWER-DOWN
PR1
DB0
RESERVED
0
DB1
LDP
0
DB2
LDF
0
RESERVED
0
DOUBLE
BUFFER
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
CONTROL
BITS
MUXOUT
DBR 1
DBR 1
RDIV2
LOW
NOISE AND
LOW SPUR
MODES
REFERENCE
DOUBLER DBR 1
RESERVED
REGISTER 2
10-BIT R COUNTER
DBR 1
CHARGE
PUMP
CURRENT
SETTING
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
L2
L1
M3
M2
M1
RD2
RD1
R10
R9
R8
R7
R6
R5
R4
R3
R2
R1
D1
0
CP3
CP2
CP1
U6
U5
1
U3
U2
U1
DB2
DB1
DB0
C3(0) C2(1) C1(0)
RESERVED
RESERVED
BOOST EN
REGISTER 3
CLK
DIV
MODE
CONTROL
BITS
12-BIT CLOCK DIVIDER VALUE
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
0
0
0
0
0
0
0
0
0
0
0
0
B1
0
C2
C1
D12
D11
D10
D9
D8
D7
D6
D5
D4
D3
D2
D1
DB2
DB1
DB0
C3(0) C2(1) C1(1)
DBB 2
DIVIDER
SELECT
MTLD
RESERVED
RF OUTPUT
ENABLE
FEEDBACK
SELECT
REGISTER 4
RESERVED
RESERVED
OUTPUT
POWER
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
0
0
0
0
0
0
0
D13
D12
D11
D10
0
0
0
0
0
0
0
0
0
D8
0
0
0
0
D3
D2
D1
CONTROL
BITS
DB2
DB1
DB0
C3(1) C2(0) C1(0)
LD PIN
MODE
RESERVED
CONTROL
BITS
RESERVED
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
DB2
ABP2 ABP1 CE1
C3(1) C2(0) C1(1)
1
0
0
0
0
D15
D14
0
0
0
0
0
0
0
0
0
0
0
0
0
1DBR = DOUBLE BUFFERED REGISTER—BUFFERED BY THE WRITE TO REGISTER 0.
2DBB = DOUBLE BUFFERED BITS—BUFFERED BY THE WRITE TO REGISTER 0, IF AND ONLY IF DB13 OF REGISTER 2 IS HIGH.
Figure 19. Register Summary
Rev. 0 | Page 13 of 28
0
0
0
0
0
0
DB1
DB0
09058-015
RESERVED
CC ENABLE
ABP
WIDTH
REGISTER 5
RESERVED
ADF4150HV
16-BIT INTEGER VALUE (INT)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
N16
N15
N14
N13
N12
N11
N10
N9
N8
N7
N6
N5
N4
N3
N2
N1
F12
F11
F10
F9
F8
F7
F6
DB7 DB6
F5
F4
DB5 DB4
F3
F2
DB3
F1
DB2
DB1
DB0
C3(0) C2(0) C1(0)
N16
N15
...
N5
N4
N3
N2
N1
INTEGER VALUE (INT)
F12
F11
...
F2
F1
FRACTIONAL VALUE (FRAC)
0
0
...
0
0
0
0
0
NOT ALLOWED
0
0
...
0
0
0
0
0
...
0
0
0
0
1
NOT ALLOWED
0
0
...
0
1
1
0
0
...
0
0
0
1
0
NOT ALLOWED
0
0
...
1
0
2
.
.
...
.
.
.
.
.
...
0
0
...
1
1
3
0
0
...
1
0
1
1
0
NOT ALLOWED
.
.
...
.
.
.
0
0
...
1
0
1
1
1
23
.
.
...
.
.
.
0
0
...
1
1
0
0
0
24
.
.
...
.
.
.
.
.
...
.
.
.
.
.
...
1
1
...
0
0
4092
1
1
...
1
1
1
0
1
65,533
1
1
...
0
1
4093
1
1
...
1
1
1
1
0
65,534
1
1
...
1
0
4094
1
1
...
1
1
1
1
1
65,535
1
1
...
1
1
4095
09058-016
0
CONTROL
BITS
12-BIT FRACTIONAL VALUE (FRAC)
INTmin = 75 WITH PRESCALER = 8/9
PRESCALER
Figure 20. Register 0 (R0)
RESERVED
DBR
12-BIT PHASE VALUE (PHASE)
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
0
0
0
PR1
P12
P11
P10
P9
P8
P7
P6
P5
P4
P3
P2
P1
M12
M11
M10
M9
M8
M7
M6
DB7 DB6
M5
M4
P1
PRESCALER
P12
P11
...
P2
P1
PHASE VALUE (PHASE)
M12
M11
...
M2
M1
0
4/5
0
0
...
0
0
0
0
0
...
1
0
2
1
8/9
0
0
...
0
1
1 (RECOMMENDED)
0
0
...
1
1
3
0
0
...
1
0
2
.
.
...
.
.
.
0
0
...
1
1
3
.
.
...
.
.
.
.
.
...
.
.
.
.
.
...
.
.
.
1
1
0
0
4092
.
.
...
.
.
.
...
1
1
...
0
1
4093
.
.
...
.
.
.
1
1
...
1
0
4094
1
1
...
0
0
4092
1
1
...
1
1
4095
1
1
...
0
1
4093
1
1
...
1
0
4094
1
1
...
1
1
4095
DB5 DB4
M3
M2
DB3
M1
DB2
DB1
DB0
C3(0) C2(0) C1(1)
INTERPOLATOR MODULUS (MOD)
09058-017
0
CONTROL
BITS
DBR
12-BIT MODULUS VALUE (MOD)
Figure 21. Register 1 (R1)
Rev. 0 | Page 14 of 28
COUNTER
RESET
CP THREESTATE
POWER-DOWN
LDP
RESERVED
CHARGE
PUMP
CURRENT
SETTING
LDF
DBR
RESERVED
DBR
10-BIT R COUNTER
DOUBLE
BUFFER
MUXOUT
DBR
RDIV2
LOW
NOISE AND
LOW SPUR
MODES
REFERENCE
DOUBLER DBR
RESERVED
ADF4150HV
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
L2
L1
M3
M2
M1
RD2
RD1
R10
L2
L1
NOISE MODE
RD2
REFERENCE
DOUBLER
0
0
LOW NOISE MODE
0
DISABLED
0
1
RESERVED
1
ENABLED
1
0
RESERVED
1
1
LOW SPUR MODE
R9
R8
R7
R6
R5
1
0
2
.
.
...
.
.
.
.
.
...
.
.
.
.
.
...
.
.
.
1
1
...
0
0
1020
1
1
...
0
1
1021
1
1
...
1
0
1022
1
1
...
1
1
1023
THREE-STATE OUTPUT
1
DVDD
DB2
DB1
DB0
C3(0) C2(1) C1(0)
0
DISABLED
0
DISABLED
1
ENABLED
1
INT-N
1
ENABLED
48
96
144
192
240
288
336
384
1
U1
COUNTER
RESET
ICP (µA)
5.1kΩ
1
U2
U1
CP1
0
U3
FRAC-N
0
1
0
1
0
1
0
1
...
1
LDF
0
0
1
1
0
0
1
1
...
U5
0
CP2
0
U6
U6
0
0
0
0
1
1
1
1
0
CP1
DOUBLE BUFFER
R4[DB22:DB20]
CP3
0
0
CP2
ENABLED
0
0
CP3
DISABLED
R COUNTER (R)
0
0
1
R1
M1
D1
0
R2
M2
R1
REFERENCE DIVIDE-B Y-2
...
0
R2
RD1
R9
0
R3
D1
R10
M3
R4
CONTROL
BITS
U5
LDP
U2
CP
THREE-STATE
0
10ns
0
DISABLED
1
6ns
1
ENABLED
RESERVED
BIT
U3
POWER-DOWN
0
DISABLED
0
RESERVED
1
ENABLED
1
NORMAL
OPERATION
OUTPUT
0
1
0
GND
0
1
1
R COUNTER OUTPUT
1
0
0
N DIVIDER OUTPUT
1
0
1
1
1
0
DIGITAL LOCK DETECT
1
1
1
RESERVED
09058-018
ANALOG LOCK DETECT
RESERVED
CLK
DIV
MODE
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8
0
0
0
0
0
0
0
0
0
0
0
0
0
B1
0
C2
C1
CONTROL
BITS
12-BIT CLOCK DIVIDER VALUE
D12
D11
D10
D9
D8
D7
D6
DB7 DB6
D5
D4
DB5 DB4
D3
D2
DB3
D1
B1
BOOST
ENABLE
D12
D11
...
D2
D1
CLOCK DIVIDER VALUE
0
0
...
0
0
0
0
DISABLED
0
0
...
0
1
1
1
ENABLED
0
0
...
1
0
2
0
0
...
1
1
3
.
.
...
.
.
.
.
.
...
.
.
.
.
.
...
.
.
.
C2
C1
CLOCK DIVIDER MODE
0
0
CLOCK DIVIDER OFF
1
1
...
0
0
4092
0
1
RESERVED
1
1
...
0
1
4093
1
0
RESYNC ENABLE
1
1
...
1
0
4094
1
1
RESERVED
1
1
...
1
1
4095
Figure 23. Register 3 (R3)
Rev. 0 | Page 15 of 28
DB2
DB1
DB0
C3(0) C2(1) C1(1)
09058-019
BOOST EN
RESERVED
Figure 22. Register 2 (R2)
DBB
DIVIDER
SELECT
MTLD
RESERVED
RF OUTPUT
ENABLE
FEEDBACK
SELECT
ADF4150HV
RESERVED
RESERVED
OUTPUT
POWER
CONTROL
BITS
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3
0
0
0
0
0
0
0
D13
D12
D11
D10
0
0
0
0
0
0
0
0
0
D8
0
0
0
0
D3
D2
FEEDBACK
D13 SELECT
0
DIVIDED
FUNDAMENTAL
1
D12
D11
D10
RF DIVIDER SELECT
0
0
0
÷1
0
0
1
÷2
0
1
0
÷4
0
1
1
÷8
1
0
0
÷16
D1
DB2
DB1
DB0
C3(1) C2(0) C1(0)
D2
D1
OUTPUT POWER (dBm)
0
0
–4
0
1
–1
1
0
+2
1
1
+5
D8
MUTE TILL
LOCK DETECT
0
MUTE DISABLED
D3
RF OUT
1
MUTE ENABLED
0
DISABLED
1
ENABLED
09058-020
0
ABP
WIDTH
RESERVED
CC ENABLE
Figure 24. Register 4 (R4)
LD PIN
MODE
RESERVED
CONTROL
BITS
RESERVED
DB31 DB30 DB29 DB28 DB27 DB26 DB25 DB24 DB23 DB22 DB21 DB20 DB19 DB18 DB17 DB16 DB15 DB14 DB13 DB12 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4
ABP2 ABP1
CE1
1
0
0
0
0
D15
D14
0
0
D15
D14
LOCK DETECT PIN OPERATION
0
0
OUTPUT LOW
0
1
DIGITAL LOCK DETECT1
1
0
OUTPUT LOW
1
1
OUTPUT HIGH
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
DB3
0
DB2
DB1
DB0
C3(1) C2(0) C1(1)
RESERVED BIT
RESERVED
1
NORMAL OPERATION
CHARGE CANCELL ATION
0
DISABLED
1
ENABLED
ABP2
ABP1
ANTIBACKLASH PULSE WIDTH
0
0
0
1
4.2ns (RECOMMENDED)
RESERVED
1
0
RESERVED
1
1
RESERVED
1MUXOUT
09058-021
CE1
0
IN REGISTER 2 MUST ALSO BE SET TO DIGITAL LOCK DETECT FOR THE LOCK DETECT PIN TO OPERATE CORRECTLY.
Figure 25. Register 5 (R5)
Rev. 0 | Page 16 of 28
ADF4150HV
In most applications, the phase relationship between the RF
signal and the reference is not important. In such applications,
the phase value can be used to optimize the fractional and
subfractional spur levels. For more information, see the Spur
Consistency and Fractional Spur Optimization section.
REGISTER 0
Control Bits
When Bits[C3:C1] are set to 000, Register 0 is programmed.
Figure 20 shows the input data format for programming this
register.
If neither the phase resync nor the spurious optimization function
is used, it is recommended that the phase word be set to 1.
16-Bit Integer Value (INT)
The 16 INT bits (Bits[DB30:DB15]) set the INT value, which
determines the integer part of the feedback division factor. The
INT value is used in Equation 1 (see the INT, FRAC, MOD, and
R Counter Relationship section). Integer values from 23 to
32,767 are allowed for the 4/5 prescaler; for the 8/9 prescaler,
the minimum integer value is 75 and the maximum value is
65,535.
12-Bit Modulus Value (MOD)
The 12 MOD bits (Bits[DB14:DB3]) set the fractional modulus.
The fractional modulus is the ratio of the PFD frequency to the
channel step resolution on the RF output. For more information,
see the 12-Bit Programmable Modulus section.
REGISTER 2
12-Bit Fractional Value (FRAC)
Control Bits
The 12 FRAC bits (Bits[DB14:DB3]) set the numerator of the
fraction that is input to the Σ-Δ modulator. This fraction, along
with the INT value, specifies the new frequency channel that
the synthesizer locks to, as shown in the RF Synthesizer—A
Worked Example section. FRAC values from 0 to (MOD − 1)
cover channels over a frequency range equal to the PFD reference frequency.
When Bits[C3:C1] are set to 010, Register 2 is programmed.
Figure 22 shows the input data format for programming this
register.
REGISTER 1
Control Bits
When Bits[C3:C1] are set to 001, Register 1 is programmed.
Figure 21 shows the input data format for programming this
register.
Prescaler Value
The dual-modulus prescaler, along with the INT, FRAC, and
MOD values, determines the overall division ratio from the VCO
output to the PFD input. The PR1 bit (DB27) in Register 1 sets
the prescaler value.
Operating at CML levels, the prescaler takes the clock from the
VCO output and divides it down for the counters. The prescaler
is based on a synchronous 4/5 core. When the prescaler is set to
4/5, the maximum RF frequency allowed is 3 GHz. Therefore,
when operating the ADF4150HV above 3 GHz, the prescaler
must be set to 8/9. The prescaler limits the INT value as follows:
•
•
Prescaler = 4/5: NMIN = 23
Prescaler = 8/9: NMIN = 75
Low Noise and Low Spur Modes
The noise modes on the ADF4150HV are controlled by setting
Bits[DB30:DB29] in Register 2 (see Figure 22). The noise modes
allow the user to optimize a design either for improved spurious
performance or for improved phase noise performance.
When the low spur mode is chosen, dither is enabled. Dither
randomizes the fractional quantization noise so that it resembles
white noise rather than spurious noise. As a result, the part is
optimized for improved spurious performance. Low spur mode
is normally used for fast-locking applications when the PLL
closed-loop bandwidth is wide. Wide loop bandwidth is a loop
bandwidth greater than 1/10 of the RFOUT channel step resolution (fRES). A wide loop filter does not attenuate the spurs to the
same level as a narrow loop bandwidth.
For best noise performance, use the low noise mode option.
When the low noise mode is chosen, dither is disabled. This
mode ensures that the charge pump operates in an optimum
region for noise performance. Low noise mode is extremely
useful when a narrow loop filter bandwidth is available. The
synthesizer ensures extremely low noise, and the filter attenuates the spurs.
Figure 8 and Figure 9 show fractional spur levels when using
low spur mode and low noise mode. Figure 12 shows the in-band
phase noise when using low spur mode and low noise mode.
12-Bit Phase Value
Bits[DB26:DB15] control the phase word. The word must be
less than the MOD value programmed in Register 1. The phase
word is used to program the RF output phase from 0° to 360°
with a resolution of 360°/MOD. For more information, see the
Phase Resync section.
MUXOUT
The on-chip multiplexer is controlled by Bits[DB28:DB26] (see
Figure 22).
Rev. 0 | Page 17 of 28
ADF4150HV
Reference Doubler
Lock Detect Precision (LDP)
Setting the DB25 bit to 0 disables the doubler and feeds the
REFIN signal directly into the 10-bit R counter. Setting this bit to
1 multiplies the REFIN frequency by a factor of 2 before feeding
it into the 10-bit R counter. When the doubler is disabled, the
REFIN falling edge is the active edge at the PFD input to the
fractional synthesizer. When the doubler is enabled, both the
rising and falling edges of REFIN become active edges at the
PFD input.
The lock detect precision bit (Bit DB7) sets the comparison
window in the lock detect circuit. When DB7 is set to 0, the
comparison window is 10 ns; when DB7 is set to 1, the window
is 6 ns. The lock detect circuit goes high when n consecutive
PFD cycles are less than the comparison window value; n is set
by the LDF bit (DB8). For example, with DB8 = 0 and DB7 = 0,
40 consecutive PFD cycles of 10 ns or less must occur before
digital lock detect goes high. The recommended settings for
Bits[DB8:DB7] are listed in Table 7.
When the doubler is enabled and the low spur mode is chosen,
the in-band phase noise performance is sensitive to the REFIN duty
cycle. The phase noise degradation can be as much as 5 dB for
REFIN duty cycles outside a 45% to 55% range. The phase noise
is insensitive to the REFIN duty cycle in the low noise mode and
when the doubler is disabled.
The maximum allowable REFIN frequency when the doubler is
enabled is 30 MHz.
RDIV2
Setting the DB24 bit to 1 inserts a divide-by-2 toggle flip-flop
between the R counter and the PFD. This function allows a 50%
duty cycle signal to appear at the PFD input, which is necessary
when the charge pump boost mode is enabled (see the Boost
Enable section).
Table 7. Recommended LDF and LDP Bit Settings
Mode
Integer-N
Fractional-N, Low Noise Mode
Fractional-N, Low Spur Mode
The 10-bit R counter (Bits[DB23:DB14]) allows the input
reference frequency (REFIN) to be divided down to produce the
reference clock to the PFD. Division ratios from 1 to 1023 are
allowed.
Double Buffer
The DB13 bit enables or disables double buffering of
Bits[DB22:DB20] in Register 4. For information about how
double buffering works, see the Program Modes section.
DB7 (LDP)
1
1
0
Power-Down (PD)
The DB5 bit provides the programmable power-down mode.
Setting this bit to 1 performs a power-down. Setting this bit to 0
returns the synthesizer to normal operation. In software powerdown mode, the part retains all information in its registers. The
register contents are lost only if the supply voltages are removed.
When power-down is activated, the following events occur:
•
10-Bit R Counter
DB8 (LDF)
1
0
0
•
•
•
•
Synthesizer counters are forced to their load state
conditions.
Charge pump is forced into three-state mode.
Digital lock detect circuitry is reset.
RFOUT buffers are disabled.
Input registers remain active and capable of loading
and latching data.
Charge Pump Three-State
Setting the DB4 bit to 1 puts the charge pump into three-state
mode. This bit should be set to 0 for normal operation.
Charge Pump Current Setting
Counter Reset
Bits[DB11:DB9] set the charge pump current. This value
should be set to the charge pump current that the loop filter
is designed with (see Figure 22).
Lock Detect Function (LDF)
The DB8 bit configures the lock detect function (LDF). The LDF
controls the number of PFD cycles monitored by the lock detect
circuit to ascertain whether lock has been achieved. When DB8
is set to 0, the number of PFD cycles monitored is 40. When
DB8 is set to 1, the number of PFD cycles monitored is 5. It is
recommended that the DB8 bit be set to 0 for fractional-N mode
and 1 for integer-N mode.
The DB3 bit is the reset bit for the R counter and the N counter
of the ADF4150HV. When this bit is set to 1, the RF synthesizer
N counter and R counter are held in reset. For normal operation, this bit should be set to 0.
Rev. 0 | Page 18 of 28
ADF4150HV
REGISTER 3
Divider Select
Control Bits
Bits[DB22:DB20] select the value of the output divider (see
Figure 24).
When Bits[C3:C1] are set to 011, Register 3 is programmed.
Figure 23 shows the input data format for programming this
register.
Mute-Till-Lock Detect (MTLD)
Boost Enable
Setting the DB18 bit to 1 enables the charge pump boost mode.
If boost mode is enabled, the narrow loop bandwidth is maintained for spur attenuation, but faster lock times are still possible.
Boost mode speeds up locking significantly for higher values of
PFD frequencies that normally have many cycle slips.
When boost mode is enabled, an extra charge pump current cell
is turned on. This cell outputs a constant current to the loop filter
or removes a constant current from the loop filter (depending on
whether the VCO tuning voltage needs to increase or decrease
to acquire the new frequency) until VTUNE approaches the lock
voltage. The boost current is then disabled and the charge pump
current setting reverts to the user programmed value.
Loop stability is maintained because the current is constant and
is not pulsed, so there is no need to switch a compensating loop
filter resistor in and out, as in standard fast lock modes. Note that
the PFD requires a 45% to 55% duty cycle for the boost mode to
operate correctly. This duty cycle can be guaranteed by setting
the RDIV2 bit (DB24) in Register 2.
Clock Divider Mode
Bits[DB16:DB15] must be set to 10 to activate phase resync
(see the Phase Resync section). Setting Bits[DB16:DB15] to
00 disables the clock divider (see Figure 23).
When the DB10 bit is set to 1, the supply current to the RF output
stage is shut down until the part achieves lock, as measured by
the digital lock detect circuitry.
RF Output Enable
The DB5 bit enables or disables the primary RF output. If DB5
is set to 0, the primary RF output is disabled; if DB5 is set to 1,
the primary RF output is enabled.
Output Power
Bits[DB4:DB3] set the value of the primary RF output power
level (see Figure 24).
REGISTER 5
Control Bits
When Bits[C3:C1] are set to 101, Register 5 is programmed.
Figure 25 shows the input data format for programming this
register.
Antibacklash Pulse Width
Bits[DB31:DB30] set the PFD antibacklash pulse width.
The recommended value for all operating modes is 4.2 ns
(set Bits[DB31:DB30] to 00). Other antibacklash pulse width
settings are reserved and are not recommended.
Charge Cancellation
Setting the DB29 bit to 1 enables charge pump charge cancellation. This has the effect of reducing PFD spurs in integer-N
mode. In fractional-N mode, this bit should be set to 0.
12-Bit Clock Divider Value
Bits[DB14:DB3] set the 12-bit clock divider value. This value
is the timeout counter for activation of phase resync. For more
information, see the Phase Resync section.
REGISTER 4
Lock Detect Pin Operation
Bits[DB23:DB22] set the operation of the lock detect (LD) pin
(see Figure 25).
REGISTER INITIALIZATION SEQUENCE
Control Bits
At initial power-up, after the correct application of voltages to
the supply pins, the ADF4150HV registers should be started in
the following sequence:
When Bits[C3:C1] are set to 100, Register 4 is programmed.
Figure 24 shows the input data format for programming this
register.
Feedback Select
The DB23 bit selects the feedback from the VCO output to the
N counter. When this bit is set to 1, the signal is taken directly
from the VCO. When this bit is set to 0, the signal is taken from
the output of the output dividers. The dividers enable coverage
of the wide frequency band (31.25 MHz to 3.0 GHz). When the
dividers are enabled and the feedback signal is taken from the
output, the RF output signals of two separately configured PLLs
are in phase. This is useful in some applications where the positive interference of signals is required to increase the power.
1.
2.
3.
4.
5.
6.
Rev. 0 | Page 19 of 28
Register 5
Register 4
Register 3
Register 2
Register 1
Register 0
ADF4150HV
RF SYNTHESIZER—A WORKED EXAMPLE
The following equations are used to program the ADF4150HV
synthesizer:
RFOUT = [INT + (FRAC/MOD)] × (fPFD/RF Divider)
(3)
(4)
where:
REFIN is the reference frequency input.
D is the RF REFIN doubler bit (0 or 1).
R is the RF reference division factor (1 to 1023).
T is the reference divide-by-2 bit (0 or 1).
The choice of modulus (MOD) depends on the reference signal
(REFIN) available and the channel resolution (fRES) required at the
RF output. For example, a GSM system with 13 MHz REFIN sets
the modulus to 65. This means that the RF output resolution
(fRES) is the 200 kHz (13 MHz/65) necessary for GSM. With
dither off, the fractional spur interval depends on the modulus
values chosen (see Table 8).
1.5 GHz VCO in fundamental mode
3 GHz VCO with the RF divider set to 2
When enabling the RF divider, the user must decide whether to
close the PLL loop before the RF divider or after it. In this
example, the PLL loop is closed before the RF divider (see
Figure 26).
PFD
VCO
÷2
RFOUT
N
DIVIDER
Figure 26. PLL Loop Closed Before Output Divider
To minimize VCO feedthrough, the 3 GHz VCO is selected. A
channel resolution (fRESOUT) of 500 kHz is required at the output
of the RF divider. Therefore, the channel resolution at the output
of the VCO (fRES) needs to be 2 × fRESOUT, that is, 1 MHz.
MOD = REFIN/fRES
MOD = 25 MHz/1 MHz = 25
For example, consider an application that requires a 1.75 GHz
RF frequency output with a 200 kHz channel step resolution.
The system has a 13 MHz reference signal.
Another possible setup is to use the reference doubler to create
26 MHz from the 13 MHz input signal. The 26 MHz is then fed
into the PFD, and the modulus is programmed to divide by 130.
This setup also results in 200 kHz resolution but offers superior
phase noise performance over the first setup.
The programmable modulus is also very useful for multistandard
applications with different channel spacing requirements.
From Equation 4,
fPFD = [25 MHz × (1 + 0)/1] = 25 MHz
(5)
1500.5 MHz = 25 MHz × [(INT + (FRAC/25))/2]
(6)
where:
INT = 120.
FRAC = 1.
RF Divider = 2.
Unlike most other fractional-N PLLs, the ADF4150HV allows
the user to program the modulus over a 12-bit range. When
combined with the reference doubler and the 10-bit R counter,
the 12-bit modulus allows the user to set up the part in many
different configurations for the application.
One possible setup is to feed the 13 MHz reference signal
directly into the PFD and to program the modulus to divide
by 65. This setup results in the required 200 kHz resolution.
09058-022
fPFD
The reference divide-by-2 divides the reference signal by 2,
resulting in a 50% duty cycle PFD frequency. This is necessary
for the correct operation of the charge pump boost mode. For
more information, see the Boost Enable section.
12-BIT PROGRAMMABLE MODULUS
In this example, the user wants to program a 1.5 GHz RF
frequency output (RFOUT) with a 500 kHz channel resolution
(fRESOUT) required on the RF output. The reference frequency
input (REFIN) is 25 MHz. The VCO options available to the
user include the following:
•
•
REFERENCE DOUBLER AND REFERENCE DIVIDER
The on-chip reference doubler allows the input reference signal
to be doubled. Doubling the reference signal doubles the PFD
comparison frequency, which improves the noise performance
of the system. Doubling the PFD frequency usually improves
noise performance by 3 dB. Note that the PFD cannot operate
above 32 MHz due to a limitation in the speed of the Σ-Δ circuit
of the N divider.
where:
RFOUT is the RF frequency output.
INT is the integer division factor.
FRAC is the fractionality.
MOD is the modulus.
RF Divider is the output divider that divides down the VCO
frequency.
fPFD = REFIN × [(1 + D)/(R × (1 + T))]
The ADF4150HV evaluation software can be used to help
determine integer and fractional values for a given setup,
along with the actual register settings to be programmed.
It is important that the PFD frequency remain constant (in this
example, 13 MHz). This allows the user to design one loop filter
for both setups without encountering stability issues. Note that
the ratio of the RF frequency to the PFD frequency principally
affects the loop filter design, not the actual channel spacing.
Rev. 0 | Page 20 of 28
ADF4150HV
SPURIOUS OPTIMIZATION AND BOOST MODE
Integer Boundary Spurs
Narrow loop bandwidths can filter unwanted spurious signals,
but these bandwidths usually have a long lock time. A wider
loop bandwidth achieves faster lock times, but may lead to
increased spurious signals inside the loop bandwidth.
Another mechanism for fractional spur creation is the interactions between the RF VCO frequency and the reference
frequency. When these frequencies are not integer related (the
purpose of a fractional-N synthesizer), spur sidebands appear
on the VCO output spectrum at an offset frequency that corresponds to the beat note, or difference frequency, between an
integer multiple of the reference and the VCO frequency. These
spurs are attenuated by the loop filter and are more noticeable
on channels close to integer multiples of the reference where the
difference frequency can be inside the loop bandwidth (thus the
name integer boundary spurs).
The boost mode feature can achieve the same fast lock time
as the wider bandwidth, but with the advantage of a narrow
final loop bandwidth to keep spurs low (see the Boost Enable
section).
SPUR MECHANISMS
This section describes the three different spur mechanisms that
arise with a fractional-N synthesizer and how to minimize them
in the ADF4150HV.
Fractional Spurs
The fractional interpolator in the ADF4150HV is a third-order
Σ-Δ modulator with a modulus (MOD) that is programmable to
any integer value from 2 to 4095. In low spur mode (dither on),
the minimum allowable value of MOD is 50. The Σ-Δ modulator
is clocked at the PFD reference rate (fPFD), which allows PLL output frequencies to be synthesized at a channel step resolution
of fPFD/MOD.
In low noise mode (dither off), the quantization noise from the
Σ-Δ modulator appears as fractional spurs. The interval between
spurs is fPFD/L, where L is the repeat length of the code sequence
in the digital Σ-Δ modulator. For the third-order Σ-Δ modulator
used in the ADF4150HV, the repeat length depends on the value
of MOD, as listed in Table 8.
Table 8. Fractional Spurs with Dither Off (Low Noise Mode)
MOD Value (Dither Off)
MOD is divisible by 2, but not by 3
MOD is divisible by 3, but not by 2
MOD is divisible by 6
MOD is not divisible by 2, 3, or 6
Repeat
Length
2 × MOD
3 × MOD
6 × MOD
MOD
Spur Interval
Channel step/2
Channel step/3
Channel step/6
Channel step
Reference Spurs
Reference spurs are generally not a problem in fractional-N
synthesizers because the reference offset is far outside the loop
bandwidth. However, any reference feedthrough mechanism that
bypasses the loop may cause a problem. The PCB layout must
ensure adequate isolation between VCO traces and the input
reference to avoid a possible feedthrough path on the board.
SPUR CONSISTENCY AND FRACTIONAL SPUR
OPTIMIZATION
With dither off, the fractional spur pattern due to the quantization noise of the Σ-Δ modulator also depends on the particular
phase word with which the modulator is seeded.
The phase word can be varied to optimize the fractional and
subfractional spur levels on any particular frequency. Thus, a
lookup table of phase values corresponding to each frequency
can be constructed for use when programming the ADF4150HV.
If a lookup table is not used, keep the phase word at a constant
value to ensure consistent spur levels on any particular frequency.
In low spur mode (dither on), the repeat length is extended
to 221 cycles, regardless of the value of MOD, which makes the
quantization error spectrum look like broadband noise. This
may degrade the in-band phase noise at the PLL output by as
much as 10 dB. For lowest noise, dither off is a better choice,
particularly when the final loop bandwidth is low enough to
attenuate even the lowest frequency fractional spur.
Rev. 0 | Page 21 of 28
ADF4150HV
The output of a fractional-N PLL can settle to any one of the
MOD phase offsets with respect to the input reference, where
MOD is the fractional modulus. The phase resync feature of
the ADF4150HV produces a consistent output phase offset with
respect to the input reference. This is necessary in applications
where the output phase and frequency are important, such as
digital beamforming. For information about how to program
a specific RF output phase when using phase resync, see the
Phase Programmability section.
In the example shown in Figure 27, the PFD reference is 25 MHz
and MOD is 125 for a 200 kHz channel spacing. tSYNC is set to
400 μs by programming CLK_DIV_VALUE = 80.
LE
SYNC
(INTERNAL)
tSYNC
LAST CYCLE SLIP
FREQUENCY
PLL SETTLES TO
INCORRECT PHASE
Phase resync is enabled by setting Bits[DB16:DB15] in
Register 3 to 10. When phase resync is enabled, an internal
timer generates sync signals at intervals of tSYNC given by the
following formula:
PLL SETTLES TO
CORRECT PHASE
AFTER RESYNC
PHASE
tSYNC = CLK_DIV_VALUE × MOD × tPFD
–100
where:
CLK_DIV_VALUE is the decimal value programmed in
Bits[DB14:DB3] of Register 3 and can be any integer in the
range of 1 to 4095.
MOD is the modulus value programmed in Bits[DB14:DB3]
of Register 1.
tPFD is the PFD reference period.
0
100
200 300
400 500 600
TIME (µs)
700
800
900 1000
09058-025
PHASE RESYNC
Figure 27. Phase Resync Example
Phase Programmability
The phase word in Register 1 controls the RF output phase. As
this word is swept from 0 to MOD, the RF output phase sweeps
over a 360° range in steps of 360°/MOD.
When a new frequency is programmed, the second sync pulse
after the LE rising edge is used to resynchronize the output
phase to the reference. The tSYNC time must be programmed to
a value that is at least as long as the worst-case lock time. This
guarantees that the phase resync occurs after the last cycle slip
in the PLL settling transient.
Rev. 0 | Page 22 of 28
ADF4150HV
APPLICATIONS INFORMATION
ULTRAWIDEBAND PLL
MICROWAVE PLL
When paired with an octave tuning range VCO, the ADF4150HV
provides an ultrawideband PLL function using the on-board
RF dividers. With an octave tuning range at the fundamental
frequency, the RF dividers provide full frequency coverage with
no gaps down to much lower frequencies.
The ADF4150HV can be interfaced directly to a wide tuning
range microwave VCO without the need for an active filter.
Typically, most microwave VCOs have a maximum tuning range
of 15 V. In this case, set VP on the ADF4150HV to a value of
16 V or higher to ensure sufficient headroom in the charge
pump. An external prescaler, such as the ADF5001, is required
to divide down VCO frequencies that are above the maximum
RF input frequency of 3.0 GHz.
For example, using a 1 GHz to 2 GHz octave range VCO (such
as the Synergy DCYS100200-12), the user can obtain contiguous
output frequencies from 62.5 MHz to 2 GHz at the ADF4150HV
RF outputs, as shown in Figure 28. A broadband output match
is achieved using a 27 nH inductor in parallel with a 50 Ω resistor
(for more information, see the Output Matching section). With
such a wide output range, the same PLL hardware design can
generate different frequencies for each of the different hardware
platforms in the system.
In the application circuit shown in Figure 29, the ADF5001
divides down the 16 GHz VCO signal to 4 GHz, which can then
be input directly into the ADF4150HV RF inputs. The ADF5001
can be connected either single-ended or differentially to the
ADF4150HV. For best performance and to achieve maximum
power transfer, it is recommended that a differential connection
be used.
VDD
ZBIAS
RFOUT+
ZBIAS =
50Ω||27nH
ADF4150HV
PLL
RFIN+
RFOUT–
RFIN–
CPOUT
RFOUT =
62.5MHz TO 2GHz
SYNERGY DCYS100200-12
OCTAVE RANGE VCO
37Ω
VTUNE
150Ω
09058-026
150Ω
RFOUT
Figure 28. Ultrawideband PLL Using the ADF4150HV and an Octave Tuning Range VCO
Rev. 0 | Page 23 of 28
ADF4150HV
10pF
RFOUT
AC COUPLING INTEGRATED
ON ADF5001 DEVICE
RFIN+
ADF4150HV
ADF5001
PRESCALER
RFIN
GND
PLL
RFOUT
MICROWAVE
VCO
6dB PAD
37Ω
150Ω
CPOUT
RFIN–
18Ω
150Ω
18Ω
RFOUT
VTUNE
18Ω
16GHz OUT
09058-027
VDD1 VDD2
0.1µF
Figure 29. 16 GHz Microwave PLL
GENERATING THE HIGH VOLTAGE SUPPLY
The design of the boost converter is simplified using the
ADP161x Excel-based design tool. This tool is available from the
ADP1613 product page. Figure 30 shows the user inputs for an
example 5 V input to 20 V output design. To minimize voltage
ripple at the output of the converter stage, the Noise Filter
option is selected, and the Vout Ripple field is set to its
minimum value. The high voltage charge pump current draw is
2 mA maximum; therefore, a value of 10 mA is entered in the
Iout field to provide margin. When tested with the ADF4150HV
evaluation board, this design showed no evident switching
spurs at the VCO output.
Rev. 0 | Page 24 of 28
09058-028
It is possible to use a boost converter such as the Analog Devices
ADP1613 to generate the high voltage charge pump supply
from a lower voltage rail without degrading PLL performance.
To minimize any switching noise feedthrough, ensure that
sufficient decoupling is placed close to the charge pump supply
pin (Pin 6). Care should be taken to use capacitors with the
appropriate voltage rating; for example, if using a boost converter
to generate a 20 V VP supply, use capacitors with a rating of
20 V or higher.
Figure 30. ADP161x Designer Tool
ADF4150HV
Blackfin ADSP-BF527 Interface
INTERFACING TO THE ADuC702x AND
THE ADSP-BF527
The ADF4150HV has a simple SPI-compatible serial interface for
writing to the device. The CLK, DATA, and LE pins control the
data transfer. When LE goes high, the 32 bits that were clocked
into the appropriate register on each rising edge of CLK are
transferred to the appropriate latch. See Figure 2 for the timing
diagram and Table 6 for the register address table.
Figure 32 shows the interface between the ADF4150HV and
the Blackfin® ADSP-BF527 digital signal processor (DSP). The
ADF4150HV needs a 32-bit serial word for each latch write.
The easiest way to accomplish this using the Blackfin family
is to use the autobuffered transmit mode of operation with
alternate framing. This mode provides a means for transmitting
an entire block of serial data before an interrupt is generated.
ADuC702x Interface
ADF4150HV
The microcontroller is set up for SPI master mode with CPHA =
0. To initiate the operation, the I/O port driving LE is brought
low. Each latch of the ADF4150HV needs a 32-bit word, which
is accomplished by writing four 8-bit bytes from the microcontroller to the device. After the fourth byte is written, the
LE input should be brought high to complete the transfer.
ADF4150HV
MOSI
ADuC702x
I/O PORTS
CLK
DATA
LE
MUXOUT
(LOCK DETECT)
MOSI
DATA
GPIO
LE
ADSP-BF527
I/O PORTS
CE
MUXOUT
(LOCK DETECT)
Figure 32. ADSP-BF527 to ADF4150HV Interface
Set up the word length for eight bits and use four memory locations for each 32-bit word. To program each 32-bit latch, store
the 8-bit bytes, enable the autobuffered mode, and write to the
transmit register of the DSP. This last operation initiates the
autobuffer transfer. If using a faster SPI clock, make sure that
the SPI timing requirements listed in Table 2 are adhered to.
PCB DESIGN GUIDELINES FOR A CHIP SCALE
PACKAGE
CE
09058-030
SCLOCK
CLK
09058-031
Figure 31 shows the interface between the ADF4150HV and the
ADuC702x family of analog microcontrollers. The ADuC702x
family is based on an ARM7 core, but the same interface can be
used with any 8051-based microcontroller.
SCK
Figure 31. ADuC702x to ADF4150HV Interface
I/O port lines on the ADuC702x are also used to control the
power-down input (CE) and the lock detect (MUXOUT configured for lock detect and polled by the port input). When
operating in the mode described, the maximum SPI transfer
rate of the ADuC7023 is 20 Mbps. This means that the maximum rate at which the output frequency can be changed is
833 kHz. If using a faster SPI clock, make sure that the SPI
timing requirements listed in Table 2 are adhered to.
The lands on the chip scale package (CP-32-11) are rectangular.
The PCB pad for these lands must be 0.1 mm longer than the
package land length and 0.05 mm wider than the package land
width. Each land must be centered on the pad to ensure that the
solder joint size is maximized.
The bottom of the chip scale package has a central exposed
thermal pad. The thermal pad on the PCB must be at least as
large as the exposed pad. On the PCB, there must be a minimum
clearance of 0.25 mm between the thermal pad and the inner
edges of the pad pattern to ensure that shorting is avoided.
Thermal vias can be used on the PCB thermal pad to improve
the thermal performance of the package. If vias are used, they
must be incorporated into the thermal pad at 1.2 mm pitch grid.
The via diameter must be between 0.3 mm and 0.33 mm, and
the via barrel must be plated with 1 oz. of copper to plug the via.
Rev. 0 | Page 25 of 28
ADF4150HV
OUTPUT MATCHING
The output of the ADF4150HV can be matched in a number of
ways for optimum operation; the most basic is to connect a 50 Ω
resistor to AVDD. A dc bypass capacitor of 100 pF is connected
in series, as shown in Figure 33. Because the resistor is not frequency dependent, this method provides a good broadband
match. When connected to a 50 Ω load, this circuit typically
gives a differential output power equal to the values chosen by
Bits[DB4:DB3] in Register 4.
The circuit shown in Figure 34 provides a good broadband
match to 50 Ω for frequencies from 250 MHz to 3.0 GHz. The
maximum output power in this case is approximately 5 dBm.
The inductor can be increased for operation below 250 MHz.
Both single-ended architectures can be examined using the
EVAL-ADF4150HVEB1Z evaluation board.
AVDD
22nH
1nF
50Ω
RFOUT+
AV DD
50Ω
50Ω
100Ω
1nF
50Ω
RFOUT–
50Ω
Figure 33. Simple ADF4150HV Output Stage
Another solution is to connect a shunt inductor (acting as an RF
choke) to AVDD. This solution can help provide a better narrowband match and, therefore, more output power. However, because
the output stage is open-collector, it is recommended that a
termination resistor be used in addition to the RF choke to give
a defined output impedance. The termination resistor can be
either 50 Ω in parallel with the RF choke or 100 Ω connected
across the RF output pins.
22nH
AVDD
09058-032
50Ω
09058-029
100pF
RFOUT
Figure 34. Optimum ADF4150HV Output Stage
If differential outputs are not needed, the unused output can be
terminated, or both outputs can be combined using a balun.
Rev. 0 | Page 26 of 28
ADF4150HV
OUTLINE DIMENSIONS
0.30
0.25
0.18
32
25
0.50
BSC
0.80
0.75
0.70
0.50
0.40
0.30
8
16
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.20 REF
SEATING
PLANE
3.65
3.50 SQ
3.45
EXPOSED
PAD
17
TOP VIEW
PIN 1
INDICATOR
1
24
9
BOTTOM VIEW
0.25 MIN
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-WHHD.
112408-A
PIN 1
INDICATOR
5.10
5.00 SQ
4.90
Figure 35. 32-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
5 mm × 5 mm Body, Very Very Thin Quad
(CP-32-11)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
ADF4150HVBCPZ
ADF4150HVBCPZ-RL7
EVAL-ADF4150HVEB1Z
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
Package Description
32-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
32-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
Evaluation Board
Z = RoHS Compliant Part.
Rev. 0 | Page 27 of 28
Package Option
CP-32-11
CP-32-11
ADF4150HV
NOTES
©2011 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D09058-0-8/11(0)
Rev. 0 | Page 28 of 28