Agilent ATF-521P8 High Linearity Enhancement Mode [1] Pseudomorphic HEMT in 2x2 mm2 LPCC [3] Package Data Sheet Features • Single voltage operation • High linearity and P1dB Agilent Technologies’s ATF-521P8 is a single-voltage high linearity, low noise E-pHEMT housed in an 8-lead JEDECstandard leadless plastic chip carrier (LPCC[3]) package. The device is ideal as a medium-power, high-linearity amplifier. Its operating frequency range is from 50 MHz to 6 GHz. • Low noise figure Pin Connections and Package Marking Pin 8 Pin 7 (Drain) Pin 6 Pin 5 Source (Thermal/RF Gnd) Description • Excellent uniformity in product specifications Pin 1 (Source) Pin 2 (Gate) Pin 3 Pin 4 (Source) Bottom View The thermally efficient package measures only 2mm x 2mm x 0.75mm. Its backside metalization provides excellent thermal dissipation as well as visual evidence of solder reflow. The device has a Point MTTF of over 300 years at a mounting temperature of +85°C. All devices are 100% RF & DC tested. Pin 3 • Point MTTF > 300 years[2] • MSL-1 and lead-free • Tape-and-reel packaging option available Pin 8 Pin 1 (Source) Pin 2 (Gate) • Small package size: 2.0 x 2.0 x 0.75 mm3 2Px Pin 4 (Source) Pin 7 (Drain) Specifications Pin 6 2 GHz; 4.5V, 200 mA (Typ.) Pin 5 • 42 dBm output IP3 Top View Note: Package marking provides orientation and identification • 26.5 dBm output power at 1 dB gain compression • 1.5 dB noise figure • 17 dB Gain Note: 1. Enhancement mode technology employs a single positive Vgs, eliminating the need of negative gate voltage associated with conventional depletion mode devices. 2. Refer to reliability datasheet for detailed MTTF data 3. Conform to JEDEC reference outline MO229 for DRP-N 4. Linearity Figure of Merit (LFOM) is essentially OIP3 divided by DC bias power. “2P” = Device Code “x” = Month code indicates the month of manufacture. • 12.5 dB LFOM[4] Applications • Front-end LNA Q2 and Q3, driver or pre-driver amplifier for Cellular/ PCS and WCDMA wireless infrastructure • Driver amplifier for WLAN, WLL/RLL and MMDS applications • General purpose discrete E-pHEMT for other high linearity applications ATF-521P8 Absolute Maximum Ratings [1] Symbol Parameter Units Absolute Maximum VDS Drain – Source Voltage [2] V 7 VGS Gate – Source Voltage [2] V -5 to 1 VGD Gate Drain Voltage [2] V -5 to 1 IDS Drain Current [2] mA 500 IGS Gate Current mA 46 Pdiss Total Power Dissipation [3] W 1.5 Pin max. RF Input Power dBm 27 TCH Channel Temperature °C 150 TSTG Storage Temperature °C -65 to 150 θch_b Thermal Resistance [4] °C/W 45 Notes: 1. Operation of this device in excess of any one of these parameters may cause permanent damage. 2. Assumes DC quiescent conditions. 3. Board (package belly) temperatureTB is 25°C. Derate 22 mW/°C for TB > 83°C. 4. Channel to board thermal resistance measured using 150°C Liquid Crystal Measurement method. 5. Device can safely handle +27dBm RF Input Power provided IGS is limited to 46mA. IGS at P1dB drive level is bias circuit dependent. Product Consistency Distribution Charts [5, 6] 150 180 600 Stdev = 0.19 0.8V 500 0.7V IDS (mA) 400 4 VDS (V) 0 6 8 0 0 0.5 1 1.5 2 2.5 3 300 Cpk = 2.13 Stdev = 0.21 Cpk = 4.6 Stdev = 0.11 250 120 200 -3 Std 90 +3 Std 60 100 30 50 15 16 17 18 GAIN (dB) Figure 4. Gain @ 2 GHz, 4.5 V, 200 mA. Nominal = 17.2 dB, LSL = 15.5 dB, USL = 18.5 dB. -3 Std 150 19 0 25 25.5 26 +3 Std 26.5 27 27.5 P1dB (dBm) Figure 5. P1dB @ 2 GHz, 4.5 V, 200 mA. Nominal = 26.5 dBm, LSL = 25 dBm. Notes: 5. Distribution data sample size is 500 samples taken from 5 different wafers. Future wafers allocated to this product may have nominal values anywhere between the upper and lower limits. 6. Measurements are made on production test board, which represents a trade-off between optimal OIP3, P1dB and VSWR. Circuit losses have been de-embedded from actual measurements. 2 39 41 43 45 47 Figure 3. OIP3 @ 2 GHz, 4.5 V, 200 mA. Nominal = 41.9 dBm, LSL = 38.5 dBm. Figure 2. NF @ 2 GHz, 4.5 V, 200 mA. Nominal = 1.5 dB. 180 37 OIP3 (dBm) NF (dB) Figure 1. Typical I-V Curves. (VGS = 0.1 V per step) 150 +3 Std 30 30 0.4V 2 -3 Std 60 0.5V 0 90 +3 Std 60 Vgs = 0.6V 0 -3 Std 90 200 0 120 120 300 100 Cpk = 0.86 Stdev = 1.32 150 49 ATF-521P8 Electrical Specifications TA = 25°C, DC bias for RF parameters is Vds = 4.5V and Ids = 200 mA unless otherwise specified. Symbol Parameter and Test Condition Units Min. Typ. Max. Vgs Operational Gate Voltage Vds = 4.5V, Ids = 200 mA V — 0.62 — Vth Threshold Voltage Vds = 4.5V, Ids = 16 mA V — 0.28 — Idss Saturated Drain Current Vds = 4.5V, Vgs = 0V µA — 14.8 — Gm Transconductance Vds = 4.5V, Gm = ∆Idss/∆Vgs; Vgs = Vgs1 - Vgs2 Vgs1 = 0.55V, Vgs2 = 0.5V mmho — 1300 — Igss Gate Leakage Current Vds = 0V, Vgs = -4V µA -20 0.49 — NF Noise Figure [1] f = 2 GHz f = 900 MHz dB dB — — 1.5 1.2 — — G Gain [1] f = 2 GHz f = 900 MHz dB dB 15.5 — 17 17.2 18.5 — OIP3 Output 3rd Order Intercept Point [1] f = 2 GHz f = 900 MHz dBm dBm 38.5 — 42 42.5 — — P1dB Output 1dB Compressed[1] f = 2 GHz f = 900 MHz dBm dBm 25 — 26.5 26.5 — — PAE Power Added Efficiency f = 2 GHz f = 900 MHz % % 45 — 60 56 — — ACLR Adjacent Channel Leakage Power Ratio[1,2] Offset BW = 5 MHz Offset BW = 10 MHz dBc dBc — — -51.4 -61.5 — — Notes: 1. Measurements obtained using production test board described in Figure 6. 2. ACLR test spec is based on 3GPP TS 25.141 V5.3.1 (2002-06) – Test Model 1 – Active Channels: PCCPCH + SCH + CPICH + PICH + SCCPCH + 64 DPCH (SF=128) – Freq = 2140 MHz – Pin = -5 dBm – Chan Integ Bw = 3.84 MHz Input 50 Ohm Transmission Line Including Gate Bias T (0.3 dB loss) Input Matching Circuit Γ_mag = 0.55 Γ_ang = -166° (1.1 dB loss) DUT Output Matching Circuit Γ_mag = 0.35 Γ_ang = 168° (0.9 dB loss) 50 Ohm Transmission Line and Drain Bias T (0.3 dB loss) Output Figure 6. Block diagram of the 2 GHz production test board used for NF, Gain, OIP3 , P1dB and PAE and ACLR measurements. This circuit achieves a trade-off between optimal OIP3, P1dB and VSWR. Circuit losses have been de-embedded from actual measurements. 3 1 pF 3.9 nH 50 Ohm .02 λ 1.5 pF RF Input 110 Ohm .03 λ 110 Ohm .03 λ 50 Ohm .02 λ 1.5 pF RF Output DUT 12 nH 47 nH 15 Ohm 2.2 µF Drain Supply 2.2 µF Gate Supply Figure 7. Simplified schematic of production test board. Primary purpose is to show 15 Ohm series resistor placement in gate supply. Transmission line tapers, tee intersections, bias lines and parasitic values are not shown. Gamma Load and Source at Optimum OIP3 and P1dB Tuning Conditions The device’s optimum OIP3 and P1dB measurements were determined using a Maury load pull system at 4.5V, 200 mA quiesent bias: Freq (GHz) Gamma Source Mag Ang (deg) Optimum OIP3 Gamma Load OIP3 Mag Ang (deg) (dBm) Gain (dB) P1dB (dBm) PAE (%) 0.9 0.413 10.5 0.314 179.0 42.7 16.0 27.0 54.0 2 0.368 162.0 0.538 -176.0 42.5 15.8 27.5 55.3 2.4 0.318 169.0 0.566 -169.0 42.0 14.1 27.4 53.5 3.9 0.463 -134.0 0.495 -159.0 40.3 9.6 27.3 43.9 Freq (GHz) Gamma Source Mag Ang (deg) Optimum P1dB Gamma Load OIP3 Mag Ang (deg) (dBm) Gain (dB) P1dB (dBm) PAE (%) 0.9 0.587 12.7 0.613 -172.1 39.1 14.5 29.3 49.6 2 0.614 126.1 0.652 -172.5 39.5 12.9 29.3 49.5 2.4 0.649 145.0 0.682 -171.5 40.0 12.0 29.4 46.8 3.9 0.552 -162.8 0.670 -151.2 38.1 9.6 27.9 39.1 4 ATF-521P8 Typical Performance Curves (at 25°C unless specified otherwise) Tuned for Optimal OIP3 50 45 50 45 40 45 30 25 15 150 200 250 300 350 30 25 20 4.5V 4V 3V 20 OIP3 (dBm) 35 10 100 40 35 OIP3 (dBm) OIP3 (dBm) 40 4.5V 4V 3V 15 10 100 400 150 200 4.5V 4V 3V 15 400 10 100 150 30 30 30 20 4.5V 4V 3V 15 150 200 250 300 350 P1dB (dBm) 35 25 25 20 4.5V 4V 3V 15 10 100 400 150 200 Idq (mA) 250 300 350 10 100 400 11 15 15 10 250 300 350 14 13 12 4.5V 4V 3V 11 GAIN (dBm) 16 GAIN (dBm) 16 200 150 200 Id (mA) Figure 14. Small Signal Gain vs Ids and Vds at 2 GHz. Note: Bias current for the above charts are quiescent conditions. Actual level may increase depending on amount of RF drive. 10 100 150 200 250 300 350 400 300 350 9 8 7 4.5V 4V 3V 11 400 250 Figure 13. P1dB vs. Idq and Vds at 3.9 GHz. 12 150 4.5V 4V 3V Idq (mA) 17 12 400 20 17 13 350 15 Figure 12. P1dB vs. Idq and Vds at 900 MHz. 14 300 25 Idq (mA) Figure 11. P1dB vs. Idq and Vds at 2 GHz. 10 100 250 Figure 10. OIP3 vs. Ids and Vds at 3.9 GHz. 35 10 100 200 Id (mA) 35 P1dB (dBm) P1dB (dBm) 350 25 20 Figure 9. OIP3 vs. Ids and Vds at 900 MHz. Figure 8. OIP3 vs. Ids and Vds at 2 GHz. GAIN (dBm) 300 30 Id (mA) Id (mA) 5 250 35 4.5V 4V 3V 6 400 Id (mA) Figure 15. Small Signal Gain vs Ids and Vds at 900 MHz. 5 100 150 200 250 300 350 400 Id (mA) Figure 16. Small Signal Gain vs Ids and Vds at 3.9 GHz. ATF-521P8 Typical Performance Curves, continued (at 25°C unless specified otherwise) Tuned for Optimal OIP3 70 70 60 60 50 50 50 45 40 40 4.5V 4V 3V 20 150 4.5V 4V 3V 20 200 250 300 350 10 100 400 150 250 300 350 10 100 400 150 200 27 40 25 35 30 85°C 25°C -40°C 250 300 350 400 Idq (mA) Figure 19. PAE @ P1dB vs. Idq and Vds at 3.9 GHz. 20 15 GAIN (dB) 45 P1dB (dBm) OIP3 (dBm) 29 23 21 85°C 25°C -40°C 19 20 10 85°C 25°C -40°C 5 17 1 1.5 2 2.5 3 3.5 4 FREQUENCY (GHz) 60 50 40 30 85°C 25°C -40°C 20 10 1.5 2 2.5 1 1.5 2 2.5 3 3.5 Figure 21. P1dB vs. Temp and Freq tuned for optimal OIP3 at 4.5V, 200 mA. 70 1 15 0.5 FREQUENCY (GHz) Figure 20. OIP3 vs. Temp and Freq tuned for optimal OIP3 at 4.5V, 200 mA. PAE (%) 15 Figure 18. PAE @ P1dB vs. Idq and Vds at 900 MHz. 50 25 3 3.5 4 FREQUENCY (GHz) Figure 23. PAE vs Temp and Freq tuned for optimal OIP3 at 4.5V, 200 mA. Note: Bias current for the above charts are quiescent conditions. Actual level may increase depending on amount of RF drive. 6 4.5V 4V 3V Idq (mA) Figure 17. PAE @ P1dB vs. Idq and Vds at 2 GHz. 0 0.5 30 20 200 Idq (mA) 15 0.5 35 25 30 30 10 100 PAE (%) PAE (%) PAE (%) 40 4 0 0.5 1 1.5 2 2.5 3 3.5 FREQUENCY (GHz) Figure 22. Gain vs. Temp and Freq tuned for optimal OIP3 at 4.5V, 200 mA. 4 45 45 50 40 40 45 35 35 30 25 20 10 100 150 200 250 300 30 25 20 4.5V 4V 3V 15 40 OIP3 (dBm) OIP3 (dBm) OIP3 (dBm) ATF-521P8 Typical Performance Curves (at 25°C unless specified otherwise) Tuned for Optimal P1dB 4.5V 4.5V 4V 4V 3V 3V 15 350 10 100 400 150 200 10 100 400 30 30 25 20 4.5V 4V 3V 15 250 300 350 P1db (dBm) 30 200 25 20 4.5V 4V 3V 15 10 100 400 150 200 250 300 350 10 100 400 15 13 13 13 GAIN (dBm) 15 GAIN (dBm) 15 11 9 7 150 200 250 4.5V 4V 3V 150 200 300 350 Id (mA) Figure 30. Gain vs Ids and Vds at 2 GHz. Note: Bias current for the above charts are quiescent conditions. Actual level may increase depending on amount of RF drive. 5 100 300 350 400 4.5V 4V 3V 11 9 4.5V 4V 3V 7 400 250 Figure 29. P1dB vs. Idq and Vds at 3.9 GHz. 17 4.5V 4V 3V 400 Idq (mA) 17 9 350 20 17 11 300 15 Figure 28. P1dB vs. Idq and Vds at 900 MHz. Figure 27. P1dB vs. Idq and Vds at 2 GHz. 250 25 Idq (mA) Idq (mA) 5 100 200 Figure 26. OIP3 vs. Ids and Vds at 3.9 GHz. 35 150 150 Id (mA) 35 10 100 4.5V 4V 3V 15 35 P1dB (dBm) P1dB (dBm) 350 25 20 Figure 25. OIP3 vs. Ids and Vds at 900 MHz. Figure 24. OIP3 vs. Ids and Vds at 2 GHz. GAIN (dBm) 300 30 Id (mA) Id (mA) 7 250 35 150 200 250 300 350 7 400 Id (mA) Figure 31. Gain vs Ids and Vds at 900 MHz. 5 100 150 200 250 300 350 400 Id (mA) Figure 32. Gain vs Ids and Vds at 3.9 GHz. ATF-521P8 Typical Performance Curves, continued (at 25°C unless specified otherwise) Tuned for Optimal P1dB 60 55 55 50 35 45 40 PAE (%) 45 PAE (%) PAE (%) 50 40 40 35 30 35 4.5V 4V 3V 30 30 25 25 20 100 150 200 4.5V 4V 3V 250 300 350 20 100 400 150 200 50 32 45 30 150 200 300 250 350 400 Idq (mA) Figure 35. PAE @ P1dB vs. Idq and Vds at 3.9 GHz. 20 35 30 85°C 25°C -40°C 1.5 2 2.5 3 3.5 4 60 50 40 30 20 85°C 25°C -40°C 10 2 2.5 85°C 25°C -40°C 20 0.5 1 1.5 2 2.5 3 3.5 Figure 37. P1dB vs. Temp and Freq (tuned for optimal P1dB at 4.5V, 200 mA). Figure 36. OIP3 vs. Temp and Freq tuned for optimal P1dB at 4.5V, 200 mA. 1.5 24 FREQUENCY (GHz) FREQUENCY (GHz) 1 26 22 20 1 GAIN (dB) P1dB (dBm) OIP3 (dBm) 20 100 400 28 25 PAE (%) 350 15 40 3 3.5 4 FREQUENCY (GHz) Figure 39. PAE vs Temp and Freq tuned for optimal P1dB at 4.5V. Note: Bias current for the above charts are quiescent conditions. Actual level may increase depending on amount of RF drive. 8 300 Figure 34. PAE @ P1dB vs. Idq and Vds at 900 MHz. Figure 33. PAE @ P1dB vs. Idq and Vds at 2 GHz. 0 0.5 250 Idq (mA) Idq (mA) 15 0.5 4.5V 4V 3V 25 10 85°C 25°C -40°C 5 4 0 0.5 1 1.5 2 2.5 3 3.5 FREQUENCY (GHz) Figure 38. Gain vs. Temp and Freq tuned for optimal P1dB at 4.5V, 200 mA. 4 ATF-521P8 Typical Scattering Parameters at 25°C, VDS = 4.5V, IDS = 280 mA Freq. GHz Mag. S11 Ang. dB S21 Mag. Ang. dB S12 Mag. Ang. S22 Mag. Ang. MSG/MAG dB 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.5 2.0 2.5 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 13.0 14.0 15.0 16.0 17.0 18.0 0.613 0.780 0.831 0.855 0.860 0.878 0.888 0.887 0.894 0.886 0.892 0.883 0.890 0.884 0.890 0.893 0.896 0.906 0.882 0.887 0.887 0.882 0.878 0.894 0.888 0.884 0.830 0.708 0.790 -96.9 -131.8 -147.2 -156.4 -162.0 -166.7 -170.2 -172.6 -174.5 -177.2 175.0 168.7 162.8 157.2 146.6 137.0 127.9 119.5 105.6 96.4 84.6 72.3 62.2 52.0 42.0 34.6 24.7 11.0 -12.7 33.2 30.0 27.3 25.1 23.5 22.0 20.8 19.7 18.7 17.9 14.3 12.1 10.2 8.6 6.1 4.1 2.3 0.9 -0.8 -1.7 -2.9 -3.9 -5.0 -6.4 -7.6 -8.3 -9.5 -9.0 -10.3 45.79 31.50 23.26 18.04 14.98 12.62 10.95 9.63 8.65 7.82 5.20 4.01 3.24 2.71 2.02 1.60 1.31 1.11 0.92 0.82 0.72 0.64 0.56 0.48 0.42 0.38 0.34 0.35 0.31 141.7 121.6 111.0 104.1 99.7 95.6 92.8 90.0 87.9 85.4 76.3 68.4 61.5 54.5 40.6 27.6 15.4 3.7 -9.8 -22.2 -33.6 -45.8 -57.0 -67.8 -76.2 -84.3 -92.8 -99.5 -93.1 -39.5 -36.7 -36.2 -35.4 -35.2 -35.0 -34.6 -34.3 -33.7 -33.8 -32.8 -31.2 -30.0 -28.9 -27.0 -25.5 -24.2 -22.9 -21.3 -20.1 -19.3 -18.5 -18.0 -17.8 -17.3 -16.6 -16.1 -15.4 -16.4 0.011 0.015 0.015 0.017 0.017 0.018 0.019 0.019 0.021 0.020 0.023 0.027 0.032 0.036 0.045 0.053 0.061 0.071 0.086 0.098 0.109 0.119 0.126 0.130 0.137 0.147 0.156 0.169 0.152 51.3 37.1 30.6 28.2 27.4 26.1 27.4 28.9 28.5 30.3 34.6 36.7 36.8 39.2 36.1 32.4 28.2 22.9 14.5 7.2 -1.0 -10.5 -19.8 -28.6 -36.1 -42.9 -52.4 -63.8 -82.8 0.317 0.423 0.466 0.483 0.488 0.496 0.497 0.500 0.501 0.502 0.502 0.492 0.490 0.494 0.505 0.529 0.551 0.570 0.567 0.585 0.593 0.617 0.636 0.662 0.697 0.732 0.752 0.816 0.660 36.2 33.2 31.9 30.3 29.5 28.5 27.6 27.0 26.1 25.9 23.5 20.2 18.5 16.2 13.8 11.9 10.4 9.6 6.8 6.2 5.0 3.9 2.8 2.1 0.9 0.3 -1.8 -2.2 -4.3 -108.3 -138.5 -152.4 -159.9 -163.8 -167.0 -169.9 -171.7 -173.6 -175.7 178.8 173.6 169.8 165.7 157.8 150.3 142.9 135.5 127.3 117.8 107.3 97.1 86.0 74.7 67.5 58.7 51.9 46.1 41.2 Typical Noise Parameters at 25°C, VDS = 4.5V, IDS = 280 mA Fmin dB Γopt Mag. Γopt Ang. Rn 0.5 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 1.20 1.30 1.61 1.68 2.12 2.77 2.58 2.85 3.35 0.47 0.53 0.61 0.69 0.67 0.71 0.79 0.82 0.73 170.00 -177.00 -166.34 -155.85 -146.98 -134.35 -125.22 -115.35 -105.76 2.8 2.6 2.7 4.0 8.4 19.0 26.7 47.2 65.2 Ga dB 22.8 20.1 17.3 14.4 11.6 9.9 8.8 7.5 5.7 40.0 MSG/MAG and |S21|2 (dB) Freq GHz 30.0 MSG 20.0 10.0 MAG 0.0 S21 -10.0 -20.0 0 5 10 15 20 FREQUENCY (GHz) Figure 40. MSG/MAG and |S21|2 vs. Frequency at 4.5V, 280 mA. Notes: 1. Fmin values at 2 GHz and higher are based on measurements while the Fmins below 2 GHz have been extrapolated. The Fmin values are based on a set of 16 noise figure measurements made at 16 different impedances using an ATN NP5 test system. From these measurements a true Fmin is calculated. Refer to the noise parameter application section for more information. 2. S and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. The input reference plane is at the end of the gate lead. The output reference plane is at the end of the drain lead. 9 ATF-521P8 Typical Scattering Parameters, VDS = 4.5V, IDS = 200 mA Freq. GHz Mag. 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 1.5 2 2.5 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 0.823 0.873 0.879 0.885 0.883 0.897 0.895 0.894 0.900 0.893 0.894 0.889 0.888 0.892 0.884 0.891 0.889 0.902 0.881 0.891 0.876 0.885 0.885 0.893 0.889 0.894 0.840 0.719 0.794 S11 Ang. -89.9 -128.7 -145.5 -155.1 -161.1 -165.9 -169.5 -171.9 -174.7 -176.6 175.3 168.5 162.6 157.0 146.5 137.0 127.9 119.6 105.6 96.0 83.9 73.1 60.9 53.0 42.2 34.3 25.0 9.1 -8.1 dB Mag. S21 Ang. dB S12 Mag. Ang. Mag. 34.4 30.5 27.6 25.2 23.6 22.1 20.8 19.6 18.7 17.8 14.3 12.0 10.2 8.6 6.0 4.0 2.3 0.9 -0.9 -1.7 -2.9 -3.6 -4.8 -6.3 -7.2 -7.8 -8.4 -10.0 -12.2 52.21 33.39 23.90 18.25 15.12 12.66 10.95 9.59 8.64 7.78 5.17 4.00 3.22 2.69 2.00 1.59 1.30 1.11 0.90 0.83 0.72 0.66 0.57 0.48 0.44 0.41 0.38 0.32 0.25 135.6 115.7 106.3 100.5 96.6 92.9 90.5 88.0 86.2 83.7 75.7 67.8 61.3 54.5 40.7 28.3 16.4 4.8 -8.8 -20.1 -32.1 -43.7 -54.1 -66.2 -74.0 -80.6 -83.4 -90.1 -102.3 -37.9 -35.6 -34.9 -34.7 -34.4 -34.1 -33.7 -33.6 -33.1 -33.1 -32.1 -30.8 -29.8 -28.6 -26.8 -25.2 -24.0 -22.8 -21.3 -20.2 -19.3 -18.5 -18.0 -17.7 -17.2 -16.9 -16.2 -15.4 -16.7 0.013 0.017 0.018 0.018 0.019 0.020 0.021 0.021 0.022 0.022 0.025 0.029 0.032 0.037 0.046 0.055 0.063 0.072 0.086 0.098 0.108 0.119 0.126 0.131 0.138 0.143 0.154 0.171 0.147 46.2 32.0 27.0 25.8 24.8 24.2 24.2 25.3 26.2 27.6 32.6 33.6 35.2 35.6 34.4 30.5 26.4 21.0 13.3 5.6 -3.2 -12.1 -21.6 -29.9 -36.7 -44.1 -54.3 -64.8 -84.1 0.388 0.478 0.507 0.518 0.519 0.525 0.526 0.528 0.528 0.529 0.527 0.516 0.514 0.517 0.526 0.548 0.568 0.584 0.580 0.594 0.600 0.622 0.641 0.663 0.698 0.732 0.750 0.815 0.655 S22 Ang. -113.0 -143.2 -156.0 -163.1 -166.7 -169.6 -172.2 -174.0 -175.6 -177.7 177.2 172.1 168.1 164.0 156.0 148.3 141.0 133.5 124.9 115.8 105.3 95.0 84.1 73.1 65.7 57.4 51.0 44.5 40.4 MSG/MAG dB 36.0 32.9 31.2 30.1 29.0 28.0 27.2 26.6 25.9 25.5 23.2 21.4 18.4 16.7 13.5 11.9 10.1 9.4 6.7 6.4 4.6 4.2 3.0 2.1 1.2 1.0 -0.8 -3.2 -5.9 Typical Noise Parameters, VDS = 4.5V, IDS = 200 mA Fmin dB Γopt Mag. Γopt Ang. Rn 0.5 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 0.60 0.72 0.96 1.11 1.44 1.75 1.99 2.12 2.36 0.30 0.35 0.47 0.57 0.62 0.69 0.74 0.80 0.69 130.00 150.00 -175.47 -162.03 -150.00 -136.20 -127.35 -116.83 -108.38 2.8 2.6 1.9 2.1 4.5 10.0 17.0 28.5 35.6 Ga dB 20.2 18.4 16.5 13.8 11.2 9.8 8.7 7.5 5.7 40.0 MSG/MAG and |S21|2 (dB) Freq GHz 30.0 MSG 20.0 10.0 MAG 0.0 S21 -10.0 -20.0 0 5 10 15 20 FREQUENCY (GHz) Figure 41. MSG/MAG and |S21|2 vs. Frequency at 4.5V, 200 mA. Notes: 1. Fmin values at 2 GHz and higher are based on measurements while the Fmins below 2 GHz have been extrapolated. The Fmin values are based on a set of 16 noise figure measurements made at 16 different impedances using an ATN NP5 test system. From these measurements a true Fmin is calculated. Refer to the noise parameter application section for more information. 2. S and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. The input reference plane is at the end of the gate lead. The output reference plane is at the end of the drain lead. 10 ATF-521P8 Typical Scattering Parameters, VDS = 4.5V, IDS = 120 mA Freq. GHz Mag. 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 1.5 2 2.5 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 0.913 0.900 0.896 0.893 0.882 0.895 0.893 0.895 0.897 0.895 0.893 0.889 0.882 0.888 0.883 0.885 0.892 0.894 0.880 0.876 0.879 0.889 0.881 0.893 0.891 0.888 0.845 0.828 0.827 S11 Ang. -84.6 -125.0 -142.0 -152.3 -158.4 -164.2 -167.8 -170.8 -173.0 -175.5 176.0 169.2 163.6 157.9 146.8 137.7 128.0 120.4 105.7 96.5 84.4 72.8 62.4 54.0 42.1 34.1 25.3 13.2 -10.2 dB Mag. S21 Ang. dB S12 Mag. Ang. Mag. 34.2 30.3 27.4 25.1 23.4 21.8 20.6 19.5 18.5 17.6 14.1 11.8 10.0 8.4 5.9 3.8 2.1 0.6 -1.0 -1.9 -3.0 -3.8 -5.2 -6.3 -7.2 -8.3 -9.1 -11.2 -11.0 51.26 32.80 23.39 17.89 14.75 12.36 10.71 9.39 8.44 7.59 5.07 3.89 3.15 2.62 1.97 1.55 1.28 1.08 0.89 0.81 0.71 0.65 0.55 0.48 0.44 0.39 0.35 0.28 0.28 135.4 115.4 106.1 100.3 96.3 92.9 90.5 88.0 86.1 83.6 75.3 67.8 61.2 54.6 40.7 28.2 16.7 5.1 -8.7 -20.8 -32.7 -44.3 -56.0 -66.6 -72.6 -79.2 -89.6 -95.9 -92.5 -36.4 -33.9 -33.4 -32.9 -32.6 -32.7 -32.4 -32.3 -32.2 -31.8 -31.1 -30.0 -29.0 -28.2 -26.5 -25.2 -24.0 -22.8 -21.2 -20.1 -19.3 -18.6 -18.1 -17.7 -17.3 -16.8 -16.1 -15.6 -16.6 0.015 0.020 0.021 0.023 0.023 0.023 0.024 0.024 0.025 0.026 0.028 0.032 0.036 0.039 0.047 0.055 0.063 0.072 0.087 0.099 0.108 0.118 0.125 0.130 0.136 0.144 0.157 0.167 0.147 49.0 31.2 25.3 23.5 22.5 20.6 20.4 21.1 22.1 23.0 25.5 27.9 30.2 30.2 29.7 26.3 21.9 18.2 10.6 3.2 -5.2 -13.5 -23.1 -31.4 -38.4 -45.9 -55.0 -64.2 -86.1 0.423 0.499 0.522 0.530 0.531 0.537 0.537 0.539 0.539 0.540 0.538 0.528 0.526 0.528 0.536 0.556 0.576 0.591 0.585 0.602 0.605 0.624 0.642 0.664 0.697 0.732 0.751 0.821 0.654 S22 Ang. -106.6 -139.4 -153.4 -161.1 -165.0 -168.4 -171.2 -173.1 -174.8 -176.9 177.4 172.2 168.1 163.9 155.7 148.1 140.5 133.1 124.3 114.9 104.5 94.2 83.4 72.4 65.1 56.7 50.4 44.0 39.9 MSG/MAG dB 35.3 32.1 30.5 28.9 28.1 27.3 26.5 25.9 25.3 24.7 22.6 20.8 19.4 16.9 13.6 11.6 10.2 8.9 6.6 5.7 4.7 4.3 2.7 2.2 1.2 0.4 -1.5 -3.9 -4.3 Typical Noise Parameters, VDS = 4.5V, IDS = 120 mA Fmin dB Γopt Mag. Γopt Ang. Rn 0.5 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 0.60 0.72 0.81 0.92 1.24 1.50 1.60 1.88 2.02 0.19 0.30 0.44 0.56 0.59 0.70 0.75 0.81 0.68 162.00 164.00 176.97 -164.98 -155.51 -136.55 -128.59 -117.31 -109.54 3.0 2.6 2.0 2.0 3.4 11.1 16.0 24.0 28.8 Ga dB 20.0 18.3 15.9 13.6 11.1 9.7 8.7 7.6 5.6 MSG/MAG and |S21|2 (dB) 40.0 Freq GHz 30.0 MSG 20.0 10.0 MAG 0.0 S21 -10.0 -20.0 0 5 10 15 20 FREQUENCY (GHz) Figure 42. MSG/MAG and |S21|2 vs. Frequency at 4.5V, 120 mA. Notes: 1. Fmin values at 2 GHz and higher are based on measurements while the Fmins below 2 GHz have been extrapolated. The Fmin values are based on a set of 16 noise figure measurements made at 16 different impedances using an ATN NP5 test system. From these measurements a true Fmin is calculated. Refer to the noise parameter application section for more information. 2. S and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. The input reference plane is at the end of the gate lead. The output reference plane is at the end of the drain lead. 11 ATF-521P8 Typical Scattering Parameters, VDS = 4V, IDS = 200 mA Freq. GHz Mag. 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 1.5 2 2.5 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 0.843 0.879 0.888 0.892 0.886 0.896 0.897 0.898 0.896 0.896 0.898 0.887 0.893 0.886 0.887 0.894 0.898 0.896 0.879 0.888 0.872 0.880 0.875 0.908 0.898 0.888 0.815 0.725 0.792 S11 Ang. -90.5 -129.3 -146.1 -155.6 -161.5 -165.7 -169.5 -172.2 -174.9 -176.7 175.2 168.0 162.8 156.9 146.6 136.8 127.4 119.7 105.4 95.0 84.1 72.4 60.4 52.4 41.3 34.1 24.1 11.3 -9.8 dB Mag. S21 Ang. dB S12 Mag. Ang. Mag. 34.3 30.3 27.4 25.1 23.4 21.8 20.6 19.5 18.6 17.6 14.1 11.8 10.0 8.4 5.9 3.9 2.1 0.7 -0.9 -1.7 -2.9 -3.8 -4.8 -6.2 -7.1 -8.2 -8.9 -9.9 -10.2 51.89 32.88 23.48 17.91 14.80 12.37 10.74 9.39 8.47 7.61 5.06 3.91 3.15 2.63 1.97 1.57 1.28 1.09 0.90 0.82 0.72 0.65 0.58 0.49 0.44 0.39 0.36 0.32 0.31 134.8 115.0 105.8 100.1 96.3 92.7 90.5 88.1 85.9 84.0 75.7 68.1 61.7 55.1 41.5 29.4 17.7 6.3 -7.1 -19.3 -30.9 -42.8 -53.3 -63.4 -73.5 -80.2 -85.3 -90.9 -95.1 -37.7 -35.4 -35.1 -34.4 -34.2 -34.2 -33.6 -33.5 -33.3 -32.9 -32.1 -30.7 -29.5 -28.4 -26.7 -25.1 -23.9 -22.6 -21.1 -20.1 -19.2 -18.6 -18.0 -17.7 -17.2 -16.8 -16.2 -15.5 -16.6 0.013 0.017 0.018 0.019 0.020 0.020 0.021 0.021 0.022 0.023 0.025 0.029 0.034 0.038 0.046 0.056 0.064 0.074 0.088 0.099 0.110 0.118 0.126 0.130 0.138 0.144 0.156 0.167 0.147 46.5 32.1 26.0 25.1 24.6 24.1 24.7 24.4 26.5 26.3 29.9 35.2 35.8 35.8 33.2 29.6 25.5 20.4 12.4 4.7 -4.3 -12.9 -22.8 -31.4 -38.0 -45.6 -54.7 -66.0 -84.8 0.408 0.507 0.539 0.549 0.551 0.556 0.557 0.559 0.559 0.560 0.558 0.547 0.545 0.547 0.554 0.572 0.590 0.603 0.594 0.609 0.610 0.629 0.647 0.666 0.699 0.734 0.750 0.809 0.652 Typical Noise Parameters, VDS = 4V, IDS = 200 mA Fmin dB Γopt Mag. Γopt Ang. Rn 0.5 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 0.67 0.74 0.96 1.24 1.44 1.62 1.83 1.99 2.21 0.21 0.30 0.46 0.57 0.62 0.69 0.74 0.82 0.71 155.00 164.00 -176.61 -162.19 -152.18 -135.43 -127.94 -117.20 -108.96 2.8 2.6 2.1 2.8 4.5 10.0 17.0 27.7 35.3 -118.1 -146.1 -158.3 -164.8 -168.2 -170.9 -173.5 -175.2 -176.9 -178.7 176.0 170.9 166.9 162.6 154.3 146.6 139.0 131.6 122.7 113.2 102.9 92.6 81.9 71.0 64.0 55.9 49.3 43.5 39.7 MSG/MAG dB 36.0 32.9 31.2 29.7 28.7 27.9 27.1 26.5 25.9 25.2 23.1 21.3 18.9 16.3 13.6 11.9 10.3 8.9 6.6 6.1 4.4 3.8 2.8 2.6 1.5 0.5 -1.7 -3.1 -4.2 40.0 Ga dB 20.1 18.4 16.4 13.9 11.4 10.0 8.7 7.7 5.9 MSG/MAG and |S21|2 (dB) Freq GHz S22 Ang. 30.0 MSG 20.0 10.0 MAG 0.0 S21 -10.0 -20.0 0 5 10 15 20 FREQUENCY (GHz) Figure 43. MSG/MAG and |S21|2 vs. Frequency at 4V, 200 mA. Notes: 1. Fmin values at 2 GHz and higher are based on measurements while the Fmins below 2 GHz have been extrapolated. The Fmin values are based on a set of 16 noise figure measurements made at 16 different impedances using an ATN NP5 test system. From these measurements a true Fmin is calculated. Refer to the noise parameter application section for more information. 2. S and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. The input reference plane is at the end of the gate lead. The output reference plane is at the end of the drain lead. 12 ATF-521P8 Typical Scattering Parameters, VDS = 3V, IDS = 200 mA Freq. GHz Mag. 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 1.5 2 2.5 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 0.867 0.894 0.899 0.896 0.892 0.910 0.906 0.902 0.907 0.902 0.900 0.896 0.896 0.887 0.890 0.898 0.896 0.904 0.877 0.883 0.877 0.875 0.863 0.910 0.868 0.863 0.835 0.720 0.780 S11 Ang. -94.6 -132.9 -148.2 -157.2 -162.8 -167.4 -170.8 -173.6 -175.2 -177.7 174.2 168.1 162.3 156.7 145.7 136.3 127.4 119.4 104.9 94.8 83.1 71.7 60.6 51.6 40.9 33.4 25.2 11.2 -7.7 dB Mag. S21 Ang. dB S12 Mag. Ang. Mag. 33.7 29.4 26.5 24.1 22.4 20.8 19.6 18.4 17.5 16.6 13.1 10.8 9.0 7.4 4.9 3.0 1.3 -0.2 -1.6 -2.4 -3.5 -4.4 -5.4 -6.5 -7.5 -8.1 -9.6 -9.5 -11.6 48.20 29.66 21.06 16.00 13.20 11.00 9.51 8.35 7.51 6.76 4.50 3.49 2.82 2.35 1.76 1.41 1.16 0.98 0.83 0.76 0.67 0.60 0.54 0.47 0.42 0.39 0.33 0.33 0.26 132.4 113.2 104.4 99.1 95.6 92.3 90.2 87.8 86.3 84.2 76.4 69.1 63.0 56.9 43.8 32.1 21.6 10.3 -2.3 -13.0 -26.0 -36.3 -47.4 -57.9 -62.8 -74.7 -78.2 -90.8 -92.8 -36.8 -34.9 -34.1 -34.0 -33.6 -33.2 -33.2 -33.0 -32.9 -32.5 -31.5 -29.9 -29.0 -27.7 -26.1 -24.5 -23.4 -22.1 -20.7 -19.8 -18.9 -18.3 -17.8 -17.6 -17.2 -16.8 -16.3 -15.8 -17.0 0.014 0.018 0.020 0.020 0.021 0.022 0.022 0.022 0.023 0.024 0.027 0.032 0.036 0.041 0.050 0.059 0.068 0.078 0.092 0.102 0.113 0.121 0.128 0.132 0.138 0.144 0.154 0.161 0.142 45.1 28.5 23.2 23.7 24.5 22.9 23.9 24.6 27.0 26.9 32.7 32.9 34.3 35.0 32.2 28.3 23.5 17.7 9.0 1.3 -7.3 -16.6 -25.1 -33.6 -40.4 -47.6 -56.8 -67.6 -85.1 0.482 0.601 0.636 0.647 0.650 0.655 0.657 0.658 0.660 0.659 0.656 0.647 0.642 0.643 0.645 0.659 0.671 0.677 0.651 0.661 0.657 0.670 0.680 0.694 0.721 0.748 0.758 0.818 0.655 S22 Ang. -132.4 -154.2 -163.8 -169.2 -171.9 -174.4 -176.7 -178.2 -179.5 178.6 173.4 167.9 163.7 159.2 150.4 142.1 134.3 126.6 117.0 107.2 96.8 86.7 76.2 65.9 59.3 51.3 44.9 39.4 37.1 MSG/MAG dB 35.4 32.2 30.2 29.0 28.0 27.0 26.4 25.8 25.1 24.5 22.2 20.4 18.6 15.6 12.9 11.3 9.5 8.5 5.9 5.3 4.0 3.1 1.9 2.3 0.2 -0.2 -2.1 -2.6 -5.7 Typical Noise Parameters, VDS = 3V, IDS = 200 mA Fmin dB Γopt Mag. Γopt Ang. Rn 0.5 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 0.66 0.72 0.87 1.00 1.32 1.49 1.59 1.79 1.96 0.22 0.30 0.42 0.59 0.63 0.72 0.74 0.78 0.70 147.00 160.00 -179.94 -163.63 -153.81 -135.10 -128.97 -117.68 -110.04 2.9 2.6 1.9 1.6 3.7 10.0 15.0 25.1 29.2 Ga dB 20.0 18.3 16.0 13.7 11.3 9.9 8.5 7.6 5.6 40.0 MSG/MAG and |S21|2 (dB) Freq GHz 30.0 MSG 20.0 10.0 MAG 0.0 S21 -10.0 -20.0 0 5 10 15 20 FREQUENCY (GHz) Figure 44. MSG/MAG and |S21|2 vs. Frequency at 3V, 200 mA. Notes: 1. Fmin values at 2 GHz and higher are based on measurements while the Fmins below 2 GHz have been extrapolated. The Fmin values are based on a set of 16 noise figure measurements made at 16 different impedances using an ATN NP5 test system. From these measurements a true Fmin is calculated. Refer to the noise parameter application section for more information. 2. S and noise parameters are measured on a microstrip line made on 0.025 inch thick alumina carrier. The input reference plane is at the end of the gate lead. The output reference plane is at the end of the drain lead. 13 ATF-521P8 Applications Information 14 3 dB RF Input & Output Matching In order to achieve maximum linearity, the appropriate input (Γs) and output (ΓL) impedances must be presented to the device. Correctly matching from these impedances to 50Ωs will result in maximum linearity. Although ATF-521P8 may be used in other impedance systems, data collected for this data sheet is all referenced to a 50Ω system. S11* 5 dB 9 dB Application Guidelines The ATF-521P8 device operates as a normal FET requiring input and output matching as well as DC biasing. Unlike a depletion mode transistor, this enhancement mode device only requires a single positive power supply, which means a positive voltage is placed on the drain and gate in order for the transistor to turn on. This application note walks through the RF and DC design employed in a single FET amplifier. Included in this description is an active feedback scheme to accomplish this DC biasing. rules but will have different locations. Also, the location of these points is largely due to the manufacturing process and partly due to IC layout, but in either case beyond the scope of this application note. ΓS B 16 d Description Agilent’s ATF-521P8 is an enhancement mode PHEMT designed for high linearity and medium power applications. With an OIP3 of 42 dBm and a 1dB compression point of 26 dBm, ATF-521P8 is well suited as a base station transmit driver or a first or second stage LNA in a receive chain. Whether the design is for a W-CDMA, CDMA, or GSM basestation, this device delivers good linearity in the form of OIP3 or ACLR, which is required for standards with high peak to average ratios. The input load pull parameter at 2 GHz is shown in Figure 1 along with the optimum S11 conjugate match. ΓL s n Los Retur S22* Figure 1. Input Match for ATF-521P8 at 2 GHz. Thus, it should be obvious from the illustration above that if this device is matched for maximum return loss i.e. S11*, then OIP3 will be sacrificed. Conversely, if ATF-521P8 is matched for maximum linearity, then return loss will not be greater than 10 dB. For most applications, a designer requires VSWR greater than 2:1, hence limiting the input match close to S11*. Normally, the input return loss of a single ended amplifier is not critical as most basestation LNA and driver amplifiers are in a balanced configuration with 90° (quadrature) couplers. Proceeding from the same premise, the output match of this device becomes much simpler. As background information, it is important to note that OIP3 is largely dependant on the output match and that output return loss is also required to be greater than 10 dB. So, Figure 2 shows how both good output return loss and good linearity could be achieved simultaneously with the same impedance point. Figure 2. Output Match at 2 GHz. Once a designer has chosen the proper input and output impedance points, the next step is to choose the correct topology to accomplish this match. For example to perform the above output impedance transformation from 50Ω to the given load parameter of 0.53∠-176°, two possible solutions exist. The first potential match is a high pass configuration accomplished by a shunt inductor and a series capacitor shown in Figure 3 along with its frequency response in Figure 4. RFin C1 RFout L1 Figure 3. High Pass Circuit Topology. Amp Frequency Of course, these points are valid only at 2 GHz, and other frequencies will follow the same design Figure 4. High Pass Frequency Response. The second solution is a low pass configuration with a shunt capacitor and a series inductor shown in Figure 5 and 6. RFin L1 RFout C1 Figure 5. Low Pass Circuit Topology. Amp Frequency Figure 6. Low Pass Frequency Response. The actual values of these components may be calculated by hand on a Smith Chart or more accurately done on simulation software such as ADS. There are some advantages and disadvantages of choosing a high pass versus a low pass. For instance, a high pass circuit cuts off low frequency gain, which narrows the usable bandwidth of the amplifier, but consequently helps avoid potential low frequency instability problems. A low pass match offers a much broader frequency response, but it has two major disadvantages. First it has the potential for low frequency instability, and second it creates the need for an extra DC blocking capacitor on the input in order to isolate the device gate from the preceding stages. precipitously giving a narrow band frequency response, yet still wide enough to accommodate a CDMA or WCDMA transmit band. For more information on RF matching techniques refer to MGA-53543 application note. Passive Bias [1] Once the RF matching has been established, the next step is to DC bias the device. A passive biasing example is shown in Figure 8. In this example the voltage drop across resistor R3 sets the drain current (Id) and is calculated by the following equation: A voltage divider network with R1 and R2 establishes the typical gate bias voltage (Vg). R1 = R2 = Vg Ibb (2) p (Vdd – Vg) x R1 Vg Often the series resistor, R4, is added to enhance the low frequency stability. The complete passive bias example may be found in reference [1]. C4 C1 INPUT Q1 Zo R3 = Vdd – Vds Ids + Ibb (3) p L1 C2 (1) OUTPUT Zo L4 C5 p R4 where, Vdd is the power supply voltage; Vds is the device drain to source voltage; Ids is the device drain to source current; C3 R3 C6 Ib R5 R1 R2 Vdd Figure 8. Passive Biasing. Ibb for DC stability is 10X the typical gate current; RFin RFout C3 C1 Zo Zo 52 L1 C2 Total Response Figure 7 displays the input and output matching selected for ATF-521P8. In this example the input and output match both essentially function as high pass filters, but the high frequency gain of the device rolls off 15 Output Match ATF-521P8 Input Match Amp Amp Amp + Frequency Amp + Frequency Figure 7. Input and Output Match for ATF-521P8 at 2 GHz. = Frequency Frequency Active Bias [2] Due to very high DC power dissipation and small package constraints, it is recommended that ATF-521P8 use active biasing. The main advantage of an active biasing scheme is the ability to hold the drain to source current constant over a wide range of temperature variations. A very inexpensive method of accomplishing this is to use two PNP bipolar transistors arranged in a current mirror configuration as shown in Figure 9. Due to resistors R1 and R3, this circuit is not acting as a true current mirror, but if the voltage drop across R1 and R3 is kept identical then it still displays some of the more useful characteristics of a current mirror. For example, transistor Q1 is configured with its base and collector tied together. This acts as a simple PN junction, which helps temperature compensate the EmitterBase junction of Q2. R2 Q1 To calculate the values of R1, R2, R3, and R4 the following parameters must be know or chosen first: Ids is the device drain-to-source current; IR is the Reference current for active bias; Vdd is the power supply voltage available; Vds is the device drain-to-source voltage; Vg is the typical gate bias; Vbe1 is the typical Base-Emitter turn on voltage for Q1 & Q2; Therefore, resistor R3, which sets the desired device drain current, is calculated as follows: R3 = Vdd – Vds Ids + IC2 (4) p IC2 is chosen for stability to be 10 times the typical gate current R1 Vdd R4 Vg C6 R3 Vds Q2 C5 C4 C3 R6 R5 C8 L3 L2 RFin C1 L1 C2 Figure 9. Active Bias Circuit. 16 C7 2 2PL RFout 7 ATF-521P8 L4 The next three equations are used to calculate the rest of the biasing resistors for Figure 9. Note that the voltage drop across R1 must be set equal to the voltage drop across R3, but with a current of IR. R1 = Vdd – Vds IR (5) p R2 sets the bias current through Q1. R2 = Vds – Vbe1 IR (6) p R4 sets the gate voltage for ATF-521P8. V R4 = g (7) IC2 p where, VE and also equal to the reference current IR. Thus, by forcing the emitter voltage (VE) of transistor Q1 equal to Vds, this circuit regulates the drain current similar to a current mirror. As long as Q2 operates in the forward active mode, this holds true. In other words, the Collector-Base junction of Q2 must be kept reversed biased. PCB Layout A recommended PCB pad layout for the Leadless Plastic Chip Carrier (LPCC) package used by the ATF-521P8 is shown in Figure 10. This layout provides plenty of plated through hole vias for good thermal and RF grounding. It also provides a good transition from microstrip to the device package. For more detailed dimensions refer to Section 9 of the data sheet. Figure 10. Microstripline Layout. Source Gate Pin 8 Drain P Pin 6 Pin 5 Pin 3 Source Bottom View Figure 11. LPCC Package for ATF-521P8. This simplifies RF grounding by reducing the amount of inductance from the source to ground. It is also recommended to ground pins 1 and 4 since they are also connected to the device source. Pins 3, 5, 6, and 8 are not connected, but may be used to help dissipate heat from the package or for better alignment when soldering the device. This three-layer board (Figure 12) contains a 10-mil layer and a 52-mil layer separated by a ground plane. The first layer is Getek RG200D material with dielectric constant of 3.8. The second layer is for mechanical RF Grounding Unlike SOT packages, ATF-521P8 is housed in a leadless package with the die mounted directly to the lead frame or the belly of the package shown in Figure 11. rigidity and consists of FR4 with dielectric constant of 4.2. High Linearity Tx Driver The need for higher data rates and increased voice capacity gave rise to a new third generation standard know as Wideband CDMA or UMTS. This new standard requires higher performance from radio components such as higher dynamic range and better linearity. For example, a WCDMA waveform has a very high peak to average ratio which forces amplifiers in a transmit chain to have very good Adjacent Channel Leakage power Ratio or ACLR, or else operate in a backed off mode. If the amplifier is not backed off then the waveform is compressed and the signal becomes very nonlinear. This application example presents a highly linear transmit drive for use in the 2.14GHz frequency range. Using the RF matching techniques described earlier, ATF-521P8 is matched to the following input and output impedances: Input Match BCV62B S11* = 0.89∠ -169 Figure 13. ATF-521P8 Matching. R3 C6 R6 R5 C4 C3 C7 C1 C2 L1 short Figure 12. ATF-521P8 demoboard. 17 J2 0 L4 L3 L2 J1 ΓL = 0.53∠ -176 R1 0 R4 50 Ohm 50 Ohm C5 R2 Output Match 2PL C8 Resistor Calculated Actual R1 50Ω 49.9Ω R2 385Ω 383Ω R3 2.38Ω 2.37Ω R4 62Ω 61.9Ω 20 Gain 15 GAIN and NF (dB) As described previously the input impedance must be matched to S11* in order to guarantee return loss greater than 10 dB. A high pass network is chosen for this match. The output is matched to ΓL with another high pass network. The next step is to choose the proper DC biasing conditions. From the data sheet, ATF-521P8 produces good linearity at a drain current of 200mA and a drain to source voltage of 4.5V. Thus to construct the active bias circuit described, the following parameters are given: 10 Table 1. Resistors for Active Bias. 5 NF Ids = 200 mA IR = 10 mA Vdd = 5 V Vds = 4.5V Vg = 0.62 V Vbe1 = 0.65V Performance of ATF-521P8 at 2140 MHz ATF-521P8 delivers excellent performance in the WCDMA frequency band. With a drain-tosource voltage of 4.5V and a drain current of 200 mA, this device has 16.5 dB of gain and 1.55 dB of noise figure as show in Figure 15. Using equations 4, 5, 6, and 7, the biasing resistor values are calculated in column 2 of table 1, and the actual values used are listed in column 3. 0 1.6 1.8 2.0 2.2 2.4 2.6 FREQUENCY (GHz) Figure 15. Gain and Noise Figure vs. Frequency. Input and output return loss are both greater that 10 dB. Although somewhat narrowband, the response is adequate in the frequency range of 2110 MHz to 2170 MHz for the WCDMA downlink. If wider band response is need, using a balanced configuration improves return loss and doubles OIP3. 0 INPUT AND OUTPUT RETURN LOSS (dB) The entire circuit schematic for a 2.14 GHz Tx driver amplifier is shown below in Figure 14. Capacitors C4, C5, and C6 are added as a low frequency bypass. These terminate second order harmonics and help improve linearity. Resistors R5 and R6 also help terminate low frequencies, and can prevent resonant frequencies between the two bypass capacitors. S11 -5 -10 S22 IR R1=49.9Ω R2=383Ω Q1 Vg Vbe1+ C4=1µF C3=4.7pF C1=1.2pF L1=1.0nH C2=1.5nH Figure 14. 2140 MHz Schematic. 18 2.2 2.4 2.6 Figure 16. Input and Output Return Loss vs. Frequency. C6=.1µF R6=1.2Ω C7=150pF L3=39nH C8=1.5pF 2 2.0 R3=2.37Ω IC2 R5=10Ω 1.8 FREQUENCY (GHz) C5=1µF L2=12nH RFin +5V Vds Q2 R4=61.9Ω -15 1.6 2PL 7 ATF-521P8 RFout L4=3.9nH Perhaps the most critical system level specification for the ATF-521P8 lies in its distortionless output power. Typically, amplifiers are characterized for linearity by measuring OIP3. This is a two-tone harmonic measurement using CW signals. But because WCDMA is a modulated waveform spread across 3.84 MHz, it is difficult to correlated good OIP3 to good ACLR. Thus, both are measured and presented to avoid ambiguity. OIP3 (dBm) 40 35 30 25 2060 2080 2100 2120 2140 2160 2180 2200 FREQUENCY (MHz) Figure 17. OIP3 vs. Frequency in WCDMA Band (Pout = 12 dBm). -30 Using the 3GPP standards document Release 1999 version 2002-6, the following channel configuration was used to test ACLR. This table contains the power levels of the main channels used for Test Model 1. Note that the DPCH can be made up of 16, 32, or 64 separate channels each at different power levels and timing offsets. For a listing of power levels, channelization codes and timing offset see the entire 3GPP TS 25.141 V3.10.0 (2002-06) standards document at: http://www.3gpp.org/specs/ specs.htm -35 ACLR (dB) -40 -45 3GPP TS 25.141 V3.10.0 (2002-06) Type Pwr (dB) -50 P-CCPCH+SCH -10 -55 Primary CPICH -10 -60 PICH -18 S-CCPCH containing PCH (SF=256) -18 DPCH-64ch (SF=128) -1.1 -65 -3 2 7 12 17 22 Pout (dBm) Figure 18. ACLR vs. Pout at 5 MHz Offset. Table 3. ACLR Channel Power Configuration. C1=1.2 pF Phycomp 0402CG129C9B200 C2,C8=1.5 pF Phycomp 0402CG159C9B200 C3=4.7 pF Phycomp 0402CG479C9B200 C4,C6=.1 µF Phycomp 06032F104M8B200 C5=1 µF AVX 0805ZC105KATZA C7=150 pF Phycomp 0402CG151J9B200 L1=1.0 nH TOKO LL1005-FH1n0S L2=12 nH TOKO LL1005-FS12N L3=39 nH TOKO LL1005-FS39 L4=3.9 nH TOKO LL1005-FH3N9S R1=49.9Ω RohmRK73H1J49R9F R2=383Ω Rohm RK73H1J3830F R3=2.37Ω Rohm RK73H1J2R37F R4=61.9Ω Rohm RK73H1J61R9F R5=10Ω Rohm RK73H1J10R0F R6=1.2Ω Rohm RK73H1J1R21F Q1, Q2 Philips BCV62C J1, J2 142-0701-851 Table 2. 2140 MHz Bill of Material. 19 Thermal Design When working with medium to high power FET devices, thermal dissipation should be a large part of the design. This is done to ensure that for a given ambient temperature the transistor’s channel does not exceed the maximum rating, TCH, on the data sheet. For example, ATF-521P8 has a maximum channel temperature of 150°C and a channel to board thermal resistance of 45°C/W, thus the entire thermal design hinges from these key data points. The question that must be answered is whether this device can operate in a typical environment with ambient temperature fluctuations from -25°C to 85°C. From Figure 19, a very useful equation is derived to calculate the temperature of the channel for a given ambient temperature. These calculations are all incorporated into Agilent Technologies AppCAD. Tch (channel) Pdiss = Vds x Ids 45 θch-b Tb (board or belly of the part) θb-s Ts (sink) θs-a Ta (ambient) Figure 19. Equivalent Circuit for Thermal Resistance. Hence very similar to Ohms Law, the temperature of the channel is calculated with equation 8 below. TCH = Pdiss (θch–b + θ b–s + θs–a ) + Tamb (8) If no heat sink is used or heat sinking is incorporated into the PCB board then equation 8 may be reduced to: TCH = Pdiss (θch–b + θ b–a ) + Tamb (9) where, θb–a is the board to ambient thermal resistance; θch–b is the channel to board thermal resistance. The board to ambient thermal resistance thus becomes very important for this is the designer’s major source of heat control. To demonstrate the influence of θb-a, thermal resistance is measured for two very different scenarios using the ATF-521P8 demoboard. The first case is done with just the demoboard by itself. The second case is the ATF demoboard mounted on a chassis or metal casing, and the results are given below: ATF Demoboard θ b-a PCB 1/8" Chassis 10.4°C/W PCB no HeatSink 32.9°C/W Table 4. Thermal resistance measurements. Therefore calculating the temperature of the channel for these two scenarios gives a good indication of what type of heat sinking is needed. Case 1: Chassis Mounted @ 85°C Tch = P x (θch-b + θb-a) + Ta =.9W x (45+10.4)°C/W +85°C Tch = 135°C Case 2: No Heatsink @ 85°C Tch = P x (θch-b + θb-a) + Ta =.9W x (45+32.9)°C/W + 85°C Tch = 155°C In other words, if the board is mounted to a chassis, the channel temperature is guaranteed to be 135°C safely below the 150°C maximum. But on the other hand, if no heat sinking is used and the θb-a is above 27°C/W (32.9°C/W in this case), then the power must be derated enough to 20 lower the temperature below 150°C. This can be better understood with Figure 20 below. Note power is derated at 13 mW/°C for the board with no heat sink and no derating is required for the chassis mounted board until an ambient temperature of 100°C. Pdiss (W) Mounted on Chassis (18 mW/°C) 0.9W 0 81 100 Summary A high linearity Tx driver amplifier for WCDMA has been presented and designed using Agilent’s ATF-521P8. This includes RF, DC and good thermal dissipation practices for reliable lifetime operation. A summary of the typical performance for ATF-521P8 demoboard at 2140 MHz is as follows: Demo Board Results at 2140 MHz No Heatsink (13 mW/°C) Gain 16.5 dB Tamb (°C) OIP3 41.2 dBm ACLR -58 dBc P1dB 24.8 dBm NF 1.55 dB 150 Figure 20. Derating for ATF- 521P8. Thus, for reliable operation of ATF-521P8 and extended MTBF, it is recommended to use some form of thermal heatsinking. This may include any or all of the following suggestions: • Maximize vias underneath and around package; • Maximize exposed surface metal; • Use 1 oz or greater copper clad; • Minimize board thickness; • Metal heat sinks or extrusions; • Fans or forced air; • Mount PCB to Chassis. References [1] Ward, A. (2001) Agilent ATF-54143 Low Noise Enhancement Mode Pseudomorphic HEMT in a Surface Mount Plastic Package, 2001 [Internet], Available from: <http://www.agilent.com/view/rf> [Accessed 22 August, 2002]. [2] Biasing Circuits and Considerations for GaAs MESFET Power Amplifiers, 2001 [Internet], Available from: <http://www.rf-solutions.com/ pdf/AN-0002_ajp.pdf> [Accessed 22 August, 2002] Device Models Refer to Agilent’s Web Site www.agilent.com/view/rf Ordering Information Part Number No. of Devices Container ATF-521P8-TR1 3000 7” Reel ATF-521P8-TR2 10000 13”Reel ATF-521P8-BLK 100 antistatic bag 2 x 2 LPCC (JEDEC DFP-N) Package Dimensions D1 D pin1 P pin1 8 1 2 e E1 3 R 2PX 4 5 Top View Bottom View A1 A A2 End View End View DIMENSIONS SYMBOL A A1 A2 b D D1 E E1 e MIN. 0.70 0 0.225 1.9 0.65 1.9 1.45 NOM. 0.75 0.02 0.203 REF 0.25 2.0 0.80 2.0 1.6 0.50 BSC DIMENSIONS ARE IN MILLIMETERS 21 E 6 b L A 7 MAX. 0.80 0.05 0.275 2.1 0.95 2.1 1.75 PCB Land Pattern and Stencil Design 2.72 (107.09) 2.80 (110.24) 0.70 (27.56) 0.63 (24.80) 0.25 (9.84) 0.22 (8.86) 0.25 (9.84) PIN 1 φ0.20 (7.87) 0.50 (19.68) 0.50 (19.68) Solder mask RF transmission line 0.32 (12.79) PIN 1 1.54 (60.61) 1.60 (62.99) 0.28 (10.83) + 0.60 (23.62) 0.25 (9.74) 0.63 (24.80) 0.72 (28.35) 0.80 (31.50) 0.15 (5.91) 0.55 (21.65) Stencil Layout (top view) PCB Land Pattern (top view) Device Orientation 4 mm REEL 8 mm CARRIER TAPE USER FEED DIRECTION COVER TAPE 22 2PX 2PX 2PX 2PX Tape Dimensions P0 P D P2 E F W + + D1 Tt t1 K0 10° Max 10° Max A0 DESCRIPTION CAVITY PERFORATION CARRIER TAPE COVER TAPE DISTANCE 23 B0 SYMBOL SIZE (mm) SIZE (inches) LENGTH A0 2.30 ± 0.05 0.091 ± 0.004 WIDTH B0 2.30 ± 0.05 0.091 ± 0.004 DEPTH K0 1.00 ± 0.05 0.039 ± 0.002 PITCH P 4.00 ± 0.10 0.157 ± 0.004 BOTTOM HOLE DIAMETER D1 1.00 + 0.25 0.039 + 0.002 DIAMETER D 1.50 ± 0.10 0.060 ± 0.004 PITCH P0 4.00 ± 0.10 0.157 ± 0.004 POSITION E 1.75 ± 0.10 0.069 ± 0.004 WIDTH W THICKNESS t1 8.00 + 0.30 8.00 – 0.10 0.254 ± 0.02 0.315 ± 0.012 0.315 ± 0.004 0.010 ± 0.0008 WIDTH C 5.4 ± 0.10 0.205 ± 0.004 TAPE THICKNESS Tt 0.062 ± 0.001 0.0025 ± 0.0004 CAVITY TO PERFORATION (WIDTH DIRECTION) F 3.50 ± 0.05 0.138 ± 0.002 CAVITY TO PERFORATION (LENGTH DIRECTION) P2 2.00 ± 0.05 0.079 ± 0.002 www.agilent.com/semiconductors For product information and a complete list of distributors, please go to our web site. For technical assistance call: Americas/Canada: +1 (800) 235-0312 or (916) 788 6763 Europe: +49 (0) 6441 92460 China: 10800 650 0017 Hong Kong: (+65) 6271 2451 India, Australia, New Zealand: (+65) 6271 2394 Japan: (+81 3) 3335-8152(Domestic/International), or 0120-61-1280(Domestic Only) Korea: (+65) 6271 2194 Malaysia, Singapore: (+65) 6271 2054 Taiwan: (+65) 6271 2654 Data subject to change. Copyright © 2003 Agilent Technologies, Inc. Obsoletes 5988-8403 July 29, 2003 5988-9974EN