APW8811 System Power PWM Controller for Notebook Computers Features Simplified Application Circuit • • VIN 6V~28V Wide Input Voltage Range from 6V to 25V Provide 4 Independent Outputs with ±1.5% Accuracy Over-Temperature VLDO5 - PWM1 Controller with Adjustable (2V to 5.5V) Out- VOUT1 put - PWM2 Controller with Adjustable (2V to 5.5V) Out- EN LDO Q1 LDO3 L1 Q2 put LDO5 EN PWM Q3 L2 PWM1 PWM2 VLDO3 VOUT2 Q4 - 100mA Low Dropout Regulator (LDO5) with Fixed 5V Output - 100mA Low Dropout Regulator (LDO3) with Fixed 3.3V Output General Description Excellent Line/Load Regulations about ±1.5% Over- The APW8811 integrates dual step-down, constant-on- • • • Temperature Range ±1%, (±1.5%, 50µA) 2.0V Reference Voltage Output time, synchronous PWM controllers (that drives dual Nchannel MOSFETs for each channel) and two low drop- Built-In POR Control Scheme Implemented Selectable Forced-PWM or Automatic PFM/PWM out regulators as well as various protections into a chip. The PWM controllers step down high voltage of a battery • (with Selectable Ultrasonic Operation) Constant-On-Time Control Scheme with Frequency to generate low-voltage for NB applications. The output of PWM1 and PWM2 can be adjusted from 2V to 5.5V by Compensation for PWM Mode Selectable Switching Frequency in PWM Mode setting a resistive voltage-divider from VOUTx to GND. The linear regulators provide 5V and 3.3V output for Built-in Digital Soft-Start for PWM Outputs and SoftStop for PWM Outputs and LDO Outputs standby power supply. The linear regulators provide up to 100mA output current. When the PWMx output voltage Integrated Bootstrap Forward P-CH MOSFET High Efficiency over Light to Full Load Range is higher than LDOx bypass threshold, the related LDOx regulator is shut off and its output is connected to VOUTx (PWMs) Built-in Power Good Indicators (PWMs) by internal switchover MOSFET. It can save power dissipation. Independent Enable Inputs (PWMs, LDO) 70% Under-Voltage and 125% Over-Voltage Protec- The APW8811 provides excellent transient response and accurate DC output voltage in either PFM or PWM Mode. tions (PWM) Adjustable Current-Limit Protection (PWMs) In Pulse-Frequency Mode (PFM), the APW8811 provides very high efficiency over light to heavy loads with loading- - Using Sense Low-Side MOSFET’s RDS(ON) Over-Temperature Protection modulated switching frequencies. The Forced-PWM mode works nearly at constant frequency for low-noise 4mmx4mm Thin QFN-24 (TQFN4x4-24A) package Lead Free and Green Device Available (RoHS requirements. The unique ultrasonic mode maintains the switching frequency above 25KHz, which eliminates Compliant) noise in audio applications. The APW8811 is equipped with accurate sourcing cur- • • • • • • • • • • • • rent-limit, output under-voltage and output over-voltage protections, being perfect for NB applications. A 1.7ms ANPEC reserves the right to make changes to improve reliability or manufacturability without notice, and advise customers to obtain the latest version of relevant information to verify before placing orders. Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 20121 1 www.anpec.com.tw APW8811 General Description (Cont.) Game Consoles Telecommunications POK BOOT1 UGATE1 PHASE1 LGATE1 22 21 20 19 17 LDO5 REF 3 16 VIN TON 4 15 PGND FB2 5 14 SKIP# ILIM2 6 13 EN LDO 11 12 PHASE2 LGATE2 Bottom View Thermal Pad 10 3-Cell and 4-Cell Li+ Battery-Powered Devices Graphic Cards 2 UGATE2 DDR1, DDR2, and DDR3 Power Supplies FB1 9 Portable Devices EN PWM BOOT2 Notebook and Sub-Notebook Computers 18 8 • • • • • • • 1 LDO3 Applications ILIM1 7 whole chip with low quiescent current close to zero. The APW8811 is available in a TQFN4x4-24A package. VOUT2 enable controls for PWM channels and LDOs. Pulling both ENPWM pin and ENLDO pin low shuts down the 23 soft-stop function actively discharges the output capacitors by the discharge device. The APW8811 has individual VOUT1 (typ.) digital soft-start can reduce the start-up current. A 24 Pin Configuration TQFN4x4-24A Top View = Thermal Pad (connected to GND plane for better heat dissipation) Ordering and Marking Information Package Code QB : TQFN4x4-24A Operating Ambient Temperature Range I : -40 to 85 oC Handling Code TR : Tape & Reel Assembly Material G : Halogen and Lead Free Device APW8811 Assembly Material Handling Code Temperature Range Package Code APW8811 QB: XXXXX - Date Code APW8811 XXXXX Note: ANPEC lead-free products contain molding compounds/die attach materials and 100% matte tin plate termination finish; which are fully compliant with RoHS. ANPEC lead-free products meet or exceed the lead-free requirements of IPC/JEDEC J-STD-020D for MSL classification at lead-free peak reflow temperature. ANPEC defines “Green” to mean lead-free (RoHS compliant) and halogen free (Br or Cl does not exceed 900ppm by weight in homogeneous material and total of Br and Cl does not exceed 1500ppm by weight). Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 2 www.anpec.com.tw APW8811 Absolute Maximum Ratings Symbol VIN VBOOT VBOOT-GND (Note 1) Rating Unit Input Power Voltage (VIN to GND) Parameter -0.3 ~ 28 V BOOT Supply Voltage (BOOT to PHASE) -0.3 ~ 7 V BOOT Supply Voltage (BOOT to GND) -0.3 ~ 35 V <400ns pulse width >400ns pulse width -5 ~ VBOOT+0.3 -0.3 ~ VBOOT+0.3 V <400ns pulse width >400ns pulse width -5 ~ VLDO5+0.3 -0.3 ~ VLDO5+0.3 V -5 ~ 35 -0.3 ~ 28 V -0.3 ~ 6 V UGATE Voltage (UGATE to PHASE) VUG-PHASE LGATE Voltage (LGATE to GND) VLG-GND PHASE Voltage (PHASE to GND) VPHASE <400ns pulse width >400ns pulse width All Other Pins (LDOx, FBx, VOUTx, LDO5, LDO3, REF, ILIMx, EN LDO, EN PWM to GND) TJ Maximum Junction Temperature TSTG Storage Temperature TSDR Maximum Lead Soldering Temperature, 10 Seconds 150 o -65 ~ 150 o 260 o C C C Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Thermal Characteristics (Note 2) Symbol Parameter Typical Value θJA Thermal Resistance - Junction to Ambient 52 θJC Thermal Resistance - Junction to Case 7 Unit o C/W Note 2: θJA and θJC are measured with the component mounted on a high effective thermal conductivity test board in free air. The thermal pad of package is soldered directly on the PCB. Recommended Operating Conditions Symbol Range Unit VIN PWM1/2 Converter Input Voltage 6 ~ 25 V VOUT1 PWM1 Converter Output Voltage 2 ~ 5.5 V VOUT2 PWM2 Converter Output Voltage CIN CLDO Parameter 2 ~ 5.5 V PWM1/2 Converter Input Capacitor (MLCC) 10 ~ µF LDO Output Capacitor (MLCC) 2.2 ~ µF TA Ambient Temperature -40 ~ 85 o TJ Junction Temperature -40 ~ 125 o Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 3 C C www.anpec.com.tw APW8811 Electrical Characteristics Refer to the typical application circuits. These specifications apply over VIN=12V and TA= -40 ~ 85 °C, unless otherwise specified. Typical values are at TA=25°C. Symbol Parameter APW8811 Test Conditions Unit Min. Typ. Max. - 0.55 1.3 mA - 5 7 mW - 200 - - 20 40 INPUT SUPPLY POWER IVN VIN Supply Current Supply current1, VOUT1 = VOUT2 = 0V, SKIP# = GND, EN LDO = open, EN PWM = 5V, VFB1 = VFB2 = 2.05V Supply current2, VOUT1 = 5V, VOUT2 = 3.3V, SKIP# = GND, EN LDO = open, EN PWM = 5V, VFB1 = VFB2 = 2.05V, PVIN+PLDO5 Standby current, EN LDO=open, EN PWM=0V Shutdown current, EN LDO= EN PWM= 0V µA UNDER-VOLTAGE LOCK OUT PROTECTION (UVLO) LDO5 UVLO threshold LDO3 UVLO threshold Rising Edge 4.1 4.2 4.3 V Hysteresis - 0.1 - V Shutdown - 2.5 - V 2 - 5.5 V 1.98 2.0 2.02 V - -1.7 - % PWM CONTROLLERS Output Voltage Adjust Ranve VFB FBx Reference Voltage VOUT1, VOUT2 o o IREF = 0A, TA = -40 C to 85 C SKIP# = LDO5, IOUT = 0A to 5A PWM 1/2 Load Regulation PWM1/2 Line Regulation IFB SKIP# = REF, IOUT = 0A to 5A - -1.5 - % SKIP# = GND, IOUT = 0A to 5A - -0.1 - % VIN = 6V to 25V - 0.005 - %/V o FBx input current VFBX = 2.0V, TA = 25 C -20 - 20 nA Soft-Start Ramp Time ENPWM High to VOUT Full Regulation - 1.7 - ms TON11 PWM1 On Time seting1 TON = GND, SKIP# = GND, VIN = 12V, PWM1 = 5V - 2080 - TON12 PWM1 On Time seting2 TON = REF, SKIP# = GND, VIN=12V, PWM1=5V - 1700 - TON13 PWM1 On Time seting3 TON = LDO3, SKIP# = GND, VIN = 12V, PWM1 = 5V - 1390 - TON14 PWM1 On Time seting4 TON = LDO5, SKIP# = GND, VIN=12V, PWM1=5V - 1140 - TON21 PWM2 On Time seting1 TON = GND, SKIP# = GND, VIN=12V, PWM2 = 3.3V - 1100 - TON22 PWM2 On Time seting2 TON = REF, SKIP# = GND, VIN=12V, PWM2 = 3.3V - 900 - TON23 PWM2 On Time seting3 TON = LDO3, SKIP# = GND, VIN = 12V, PWM2 = 3.3V - 730 - TON24 PWM2 On Time seting4 TON = LDO5, SKIP# = GND, VIN = 12V, PWM2 = 3.3V - 600 - 350 450 550 UGATEx Minimum Off-Time Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 4 ns ns www.anpec.com.tw APW8811 Electrical Characteristics (Cont.) Refer to the typical application circuits. These specifications apply over VIN=12V and TA= -40 ~ 85 °C, unless otherwise specified. Typical values are at TA=25°C. Symbol Parameter APW8811 Test Conditions Unit Min. Typ. Max. UGATEx Minimum On-Time 80 110 140 ns Minimum Ultrasonic SKIP Operating Frequency 25 37 - kHz PWM CONTROLLERS (CONT.) LOW DROUPUT LINEAR REGULATORS (LDO5/LDO3) LDO5 Output Voltage VOUT1 = GND, 6V < VIN < 25V, 0< ILDO < 100mA 4.8 5.0 5.2 V LDO3 Output Voltage VOUT2 = GND, 6V < VIN < 25V, 0< ILDO3 < 100mA 3.2 3.33 3.46 V VTHBYP5 LDO5 Bypass Threshold for VOUT1-to-LDO5 Switch On VOUT1 Regulation Voltage Rising 4.55 4.7 4.85 Hysteresis 0.15 0.25 0.3 VTHBYP3 LDO3 Bypass Threshold for VOUT2-to-LDO3 Switch On VOUT2 Regulation Voltage Rising 3.05 3.15 3.25 Hysteresis 0.1 0.2 0.25 VOUTx-to-LDOx Switch On Resistance VOUTx to LDOx, 10mA LDOx Current-Limit VOUTx = GND, LDOx = GND LDOx Discharge On Resistance V V - 1.5 3 Ω 150 - - mA - 40 65 Ω 1.98 2.00 2.02 V - 10 - mV REFFERENCE REF Output Voltage IREF = 0A REF Load Regulation ILOAD = 0 to 50µA REF Sink Current REF in Regulation 10 - - µA Over-Voltage Protection Threshold VOUTX RIsing 120 125 130 % Over-Voltage Fault Propagation Delay Delta voltage = 10mV - 1.5 - µs 9.4 10 10.6 µA - 4500 2 ppm/ oC 0.515 - 2 V 205 250 - mV PWM 1/2 PROTECTIONS o Current-Limit Current Source VILIM = 920mV, TA = 25 C On the basis of 25 oC ILIMx Adjustment Range VILIMx-GND Maximum setting voltage VILIMx= 5V, Setting Current-Limit Threshold Current-limit comparator offset (VILIMx-GND-VPGND-PHASEx), VILIMx = 920mV -8 0 8 mV Zero-Crossing Threshold SKIP# = REF or LDOx, VPGND-PHASE -5 0 5 mV Under-Voltage Protection Threshold 65 70 75 % Under-Voltage Protection Hysteresis - 3 - % Under-Voltage Protection Debounce Interval 22 32 42 µs From EN signal go high to UVP workable - 2 2.6 ms TJ Rising - 160 - Hysteresis - 25 - Under-Voltage Protection Enable Blanking Time Over-Temperature Protection THreshold Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 5 o C www.anpec.com.tw APW8811 Electrical Characteristics (Cont.) Refer to the typical application circuits. These specifications apply over VIN=12V and TA= -40 ~ 85 °C, unless otherwise specified. Typical values are at TA=25°C. Symbol Parameter APW8811 Test Conditions Unit Min. Typ. Max. POK in from Lower (POK goes high) 87 90 93 POK Threshold POK in from higher (POK goes low) 120 125 130 POK Propagation Delay VFBX falling and rising POK Enable Delay ENILMx goes high to POK goes High POK Sink Current VPOK = 500mV 2.5 POK Leakage Current VPOK = 5V - Automatic PFM/PWM Mode - Forced PWM Mode Auto Skip with Ultrasonic POWER GOOD POK hysteresis % - 3 - 43 63 85 µs - 2 - ms 7.5 - mA 0.1 1 µA - 1.5 V 1.9 - 2.1 V 2.7 - - V 200kHz/250kHz - - 1.5 V 245kHz/305kHz 1.9 - 2.1 V 300kHz/375kHz 2.7 - 3.6 V 365kHz/460kHz 4.7 - - V Shutdown - - 0.6 V Enable 2 - - V Shutdown - - 0.4 V Enable 2.4 - - V VSKIP# = VTON = 0V or 5V -1 - 1 µA IEN LDO 0.5 1 3 µA LOGIC LEVELS SKIP# Input Voltage TON Input Voltage EN PWM Input Voltage EN LDO Input Voltage Input Leakage Current GATE DRIVERS UG Pull-Up Resistance VBOOTx – VUGATEx = 100mV - 4 8 Ω UG Sink Resistance VUGATEx – VPHASEx = 100mV - 1.5 4 Ω LG Pull-Up Resistance VLDO5 – VLGATEx = 100mV - 4 8 Ω LG Sink Resistance VLGATEx – VPGND = 100mV - 1.5 4 Ω UG falling to LG rising - 40 - ns LG falling to UG rising - 40 - ns - 40 80 Ω Dead-Time VOUT1/2 Discharge On Resistance BOOTSTRAP SWITCH VF IR Forward Voltage VLDO5x – VBOOTx-GND, IF = 10mA - 0.5 0.8 V Reverse Leakage VBOOTx-GND = 30V, VPHASEx = 25V, VLDO5 = 5V - - 0.5 µA Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 6 www.anpec.com.tw APW8811 Pin Description PIN NO. FUNCTION NAME 1 ILIM1 Current-Limit Adjustment. There is an internal 10µA current source from LDO5 to ILIM1 and connected a resistor from ILIM1 to GND to set the current-limit threshold. The PGND-PHASE1 current-limit threshold is 1/10th the voltage set at ILIM1 over a 0.515 to 2V range. The logic current-limit threshold is default to 250mV value if ILIM1 is 5V. 2 FB1 Output voltage feedback pin (PWM1). It can use a resistive divider from VOUT1 to GND to adjust the output from 2V to 5.5V. 3 REF 2V Reference Output. Bypass to GND with a 0.1µF (minimum) capacitor. REF can supply external loads for 50µA (maximum). REF load-regulation error will degrade feedback and output accuracy. 4 TON Frequency Selection Input. Connect to LDO5 for 365kHz(PWM1)/460kHz(PWM2) operation and to LDO3 for 300kHz/375kHz operation. Connect to REF for 245kHz/305kHz operation and to GND for 200kHz/250kHz operation. 5 FB2 Output voltage feedback pin (PWM2). It can use a resistive divider from VOUT2 to GND to adjust the output from 2V to 5.5V. 6 ILIM2 Current-Limit Adjustment. There is an internal 10µA current source from LDO5 to ILIM2 and connected a resistor from ILIM2 to GND to set the current-limit threshold. The PGND-PHASE2 current-limit threshold is 1/10th the voltage set at ILIM2 over a 0.515 to 2V range. The logic current-limit threshold is default to 250mV value if ILIM2 is 5V. 7 VOUT2 PWM2 Output Voltage-Sense Input. The VOUT2 pin makes a direct measurement of the PWM2 output voltage. VOUT2 is an input to the constant-on-time PWM one-time one-shot circuit. 8 LDO3 3.3V Linear Regulator Output. LDO3 can provide a total of 100mA, 3.3V external loads. When LDO3 is at 3.3V and PWM2 output voltage is over 3.15V bypass threshold, the internal LDO will shut down, and LDO3 output pin connects to VOUT2 through a 1.5Ω switch. Bypass to GND with a minimum of 2.2µF ceramic capacitor for stability. 9 BOOT2 Supply Input for The UGATE2 Gate Driver and an internal level-shift circuit. Connect to an external capacitor to create a boosted voltage suitable to drive a logic-level N-channel MOSFET. 10 UGATE2 Output of The High-Side MOSFET Driver for PWM2. Connect this pin to Gate of the high-side MOSFET. 11 Junction Point of The High-Side MOSFET Source, Output Filter Inductor and The Low-Side MOSFET Drain PHASE2 for PWM2. Connect this pin to the Source of the high-side MOSFET. PHASE2 serves as the lower supply rail for the UGATE2 high-side gate driver. PHASE2 is the current-sense input for the PWM2. 12 LGATE2 Output of The Low-Side MOSFET Driver for PWM2. Connect this pin to Gate of the low-side MOSFET. Swings from PGND to LDO5. 13 EN LDO Master Enable Input. The LDOx is enabled when EN LDO=1. When ENLDO=0, the LDO is shutdown. See the table 2 “Power-Up Control Logics.” 14 SKIP# PWM1 and 2 Controller Operation Mode Control. Connect SKIP# to GND for auto PWM/PFM mode, to REF for forced-PWM mode and to LDO3 or LDO5 for ultra-sonic mode. 15 PGND Power Ground of The LGATE Low-Side MOSFET Drivers. Connect the pin to the source of the low-side MOSFETs. 16 VIN Battery voltage input pin. VIN powers linear regulators and is also used for the constant-on-time PWM on-time one-shot circuits. Connect VIN to the battery input and bypass with a 1µF capacitor for noise interference. 17 LDO5 5V Linear Regulator Output. LDO5 can provide a total of 100mA, 5V external loads. When LDO5 is at 5V and PWM1 output voltage is over 4.7V bypass threshold, the internal LDO will shut down, and LDO5 output pin connects to VOUT1 through a 1.5Ω switch. Bypass to GND with a minimum of 2.2µF ceramic capacitor for stability. 18 EN PWM PWMx Enable Input. Both PWMx are enabled when EN PWM=1. When EN PWM=0, both PWMx are in shutdown. See the table 2 “Power-Up Control Logics.” 19 LGATE1 Output of The Low-Side MOSFET Driver for PWM1. Connect this pin to Gate of the low-side MOSFET. Swings from PGND to LDO5. 20 Junction Point of The High-Side MOSFET Source, Output Filter Inductor and The Low-Side MOSFET Drain PHASE1 for PWM1. Connect this pin to the Source of the high-side MOSFET. PHASE1 serves as the lower supply rail for the UGATE1 high-side gate driver. PHASE1 is the current-sense input for the PWM1. Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 7 www.anpec.com.tw APW8811 Pin Description (Cont.) PIN NO. FUNCTION NAME 21 UGATE1 Output of The High-Side MOSFET Driver for PWM1. Connect this pin to Gate of the high-side MOSFET. 22 BOOT1 Supply Input for The UGATE1 Gate Driver and an internal level-shift circuit. Connect to an external capacitor to create a boosted voltage suitable to drive a logic-level N-channel MOSFET. 23 POK Power-Good Output Pin of Both PWMs.(Logic AND) POK is an open-drain output used to indicate the status of the PWMx output voltage. Connect the POK in to +5V through a pull-high resistor. 24 VOUT1 PWM1 Output Voltage-Sense Input. The VOUT1 pin makes a direct measurement of the PWM1 output voltage. VOUT1 is an input to the constant-on-time PWM one-time one-shot circuit. Thermal Pad GND Signal Ground for The IC. Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 8 www.anpec.com.tw APW8811 Block Diagram TON BOOT2 ADAPTIVE DEAD-TIME DIODE EMULATION PWM/PFM TRANSITION UGATE2 PHASE2 PWM FREQUENCY CONTROL VOUT2 SKIP# LDO5 LGATE2 TON Generator VOUT1 SMPS1 PWM2 CONTROLLER SMPS2 PWM2 CONTROLLER BOOT1 ADAPTIVE DEAD-TIME DIODE EMULATION PWM/PFM TRANSITION UGATE1 PHASE1 SKIP# LDO5 LGATE1 PGND SKIP# VIN LDO UVLO LDO3 LDO5 THERMAL SHUTDOWN LDO3 LDO5 EN ENABLE POWER ON SEQUENCE CLEAR FAULT LATCH VOUT2 VTHBYP3 VTHBYP5 PHASE1 PHASE2 ILIM2 ILIM1 REF ENABLE CURRENT-LIMIT CONTROLLER REF VOUT1 REF EN LDO EN PWM SOFT-START POK POK2 POK1 90% VFB2 90% VFB1 125% VFB2 OV2 FB2 FAULT LATCH LOGIC UV2 OV1 FB1 UV1 70% VFB2 70% VFB1 VOUT2 SOFTSTOP Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 125% VFB1 VOUT1 EN LDO or EN PWM EN LDO or EN PWM 9 SOFTSTOP GND www.anpec.com.tw APW8811 Typical Application Circuit VIN : 6V to 25V LDO5 VIN LDO3 CLDO5 4.7µF CIN1 10µF VOUT1 5V/7A COUT1 330µF/6.3V 9mΩ Q1 APM4810 LOUT1 4.7µH POK RPOK 200k CBOOT1 0.1µF RBOOT1 0 CLDO3 4.7µF TON BOOT1 BOOT2 UGATE1 UGATE2 RBOOT2 0 CBOOT2 0.22µF PHASE1 PHASE2 Q2 APM4810 Q4 APM4810 LGATE1 LGATE2 PGND PGND RTOP1 30k ON OFF VOUT1 EN LDO EN PWM FB1 RGND1 20k RILIM1 200k ON VOUT2 3.3V/11A COUT2 330µF/6.3Vx2 4mΩ RTOP2 13k OFF FB2 ILIM2 ILIM1 SKIP# REF GND Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 VOUT2 CIN2 10µF Q3 APM4810 LOUT2 2.2µH GND 10 RILIM2 200k CREF 0.1µF RGND2 20k www.anpec.com.tw APW8811 Function Description Where FSW is the nominal switching frequency of the converter in PWM mode. Similarly, the on-time of ultrasonic Constant-On-Time PWM Controller with Input Feed-Forward The constant-on-time control architecture is a pseudo- mode is the same with PFM mode. The description of ultrasonic mode will be illustrated later. fixed frequency with input voltage feed-forward. This architecture relies on the output filter capacitor’s effective The load current at handoff from PFM to PWM mode is given by: series resistance (ESR) to act as a current-sense resistor, so the output ripple voltage provides the PWM ramp signal. ILOAD(PFM to PWM) = In PFM operation, the high-side switch on-time controlled by the on-time generator is determined solely by a one- = shot whose pulse width is inversely proportional to input voltage and directly proportional to output voltage. In PWM operation, the high-side switch on-time is determined by a switching frequency control circuit in the on-time gen- 1 VIN − VOUT × × TON −PFM 2 L VIN − VOUT V 1 × × OUT 2L F SW VIN Forced-PWM Mode Connect SKIP# to REF for normal Forced-PWM operation. erator block. The switching frequency control circuit senses the switching frequency of the high-side switch The Forced-PWM mode disables the zero-crossing comparator, which truncates the low-side switch on-time and keeps regulating it at a constant frequency in PWM mode. The design improves the frequency variation and at the inductor current zero crossing. This causes the low-side gate-drive waveform to become the complement is more outstanding than a conventional constant-ontime controller, which has large switching frequency varia- of the high-side gate-drive waveform. This in turn causes the inductor current to reverse at light loads while UGATE tion over input voltage, output current and temperature. Both in PFM and PWM, the on-time generator, which maintains a duty factor of VOUT/VIN. The benefit of ForcedPWM mode is to keep the switching frequency fairly senses input voltage on VIN pin, provides very fast ontime response to input line transients. constant. The Forced-PWM mode is the most useful for reducing audio frequency noise, improving load-transient Another one-shot sets a minimum off-time (typ.: 300ns). The on-time one-shot is triggered if the error comparator response, and providing sink-current capability for dynamic output voltage adjustment. is high, the low-side switch current is below the currentlimit threshold, and the minimum off-time one-shot has Ultrasonic Mode timed out. Connecting SKIP# to LDO3 or LDO5 for ultrasonic mode. Pulse-Frequency Modulation (PFM) Mode The ultrasonic mode activates a unique PFM mode with a minimum switching frequency of 37kHz. The minimum Connect SKIP# to GND for normal PFM operation. In PFM frequency 37KHz of ultrasonic mode eliminates audiofrequency interference in light load condition. It will transit mode, an automatic switchover to pulse-frequency modulation (PFM) takes place at light loads. This switchover is to unique PFM mode when output loading makes the frequency bigger than ultrasonic frequency. In ultrasonic affected by a comparator that truncates the low-side switch on-time at the inductor current zero crossing. This mode, the controller automatically transits to fixed-frequency PWM operation when the load reaches the same mechanism causes the threshold between PFM and PWM operation to coincide with the boundary between critical conduction point (ILOAD(PFM to PWM)). When the controller detects that no switching has oc- continuous and discontinuous inductor-current operation (also known as the critical conduction point). The on- curred within about 27µs (typ.), an ultrasonic pulse will occurre. The ultrasonic controller turns on the low-side time of PFM is given by: TON - PFM = V 1 × OUT FSW VIN Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 MOSFET first to reduce the output voltage. After feedback voltage drops below the internal reference voltage, the 11 www.anpec.com.tw APW8811 Function Description (Cont.) Ultrasonic Mode (Cont.) Digital Soft-Start controller turns off the low-side MOSFET and triggers a constant-on-time. When the constant-on-time has The APW8811 integrates digital soft-start circuit to ramp up the PWMx output voltage of the converter to the pro- expired, the controller turns on the low-side MOSFET again until the inductor current is below the zero-cross- grammed regulation set point at a predictable slew rate. The slew rate of PWMx output voltage is internally con- ing threshold. The behavior is the same with PFM mode. trolled to limit the inrush current through the output capacitors during soft-start process. When the ENPWM pin Reference Voltages and Linear Regulator (REF and LDO3/5) is pulled above the rising threshold voltage, the both of PWM1 and PWM2 initiate a soft-start process to ramp up The 2V reference, REF, is accurate to ±1% over- the output voltage at the same time. The soft-start interval is 1.7ms(typ.) and independent of the UGATE switching temperature. Bypass to GND with a 0.1µF (minimum) capacitor. REF can source up to 50µA for external loads. frequency. However, avoid loading REF if extremely accurate specifications for both the main output voltages and REF are Enable Controls essential. In addition, REF voltage must be bigger than its rising enable threshold, and then the LDO output starts The APW8811 has two independent enable controls for PWM and LDO. When the ENLDO pin is high (EN=1), the REF, LDO3 and LDO5 are enabled to standby mode. It to rise up. The LDO3 and LDO5 regulators can supply up to 100mA means that the PWM1 and PWM2 are ready to enable at this mode. When the ENPWM pin is high (EN=1) at standby for external loads. Bypass to GND with a minimum of 2.2 µF ceramic capacitor for stability. When ENLDO is mode, the both of PWM1 and PWM2 initiate a soft-start process to ramp up the output voltage at the same time. enabled, the VLDO3 is fixed 3.33V and the VLDO5 is fixed 5V in standby mode. When PWMx output voltage is over whose When both ENLDO and ENPWM are low, the chip is in its low-power shutdown state. The APW8811 only consumes bypass threshold(PWM1 is 4.7V and PWM2 is 3.15V), the switchover between the internal LDOx and VOUTx is 20µA of quiescent current while in shutdown. Both PWM outputs are discharged to 0V through a 25Ω workable. These actions change the current path to power the loads from the PWMx regulateon voltage, rather than from the internal linear regulator. switch and both LDO outputs is discharged to 0V through a 40Ω switch in shutdown mode. Driving ENPWM or Power-On-Reset ENLDO below 0.4V clears the over-voltage, under-voltage and over-temperature fault latches. A Power-On-Reset (POR) function is designed to prevent wrong logic controls. The POR function continually moni- Soft-Stop (PWMs) tors the supply voltage on the LDO5 pins. LDO5 POR circuitry inhibits wrong switching. When the rising VLDO5 In the event of PWM under-voltage or shutdown, the chip enables the soft-stop function. The soft-stop function discharges the PWM output voltages to GND through an voltage reaches the rising POR threshold (4.2V typ.), the output voltages begin to ramp up. When the LDO5 volt- internal 25Ω switch. The reference remains active to provide an accurate threshold and to provide over-voltage age is lower than 4.1V(typ.) or LDO3 voltage is lower than 2.5V(typ.), both switch power supplies are shut off. This protection. is non-latch protection. LDO5 POR threshold could reset the under-voltage, over-voltage. Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 12 www.anpec.com.tw APW8811 Function Description (Cont.) Power Good Indicator (PWMs) continuously high output from low-side MOSFET driver. POK is actively held low in shutdown, standby, and soft- It’s a common problem for OVP schemes with a latch. Once an over-voltage fault condition is set, it can be reset start. In the soft-start process, the POK is an open-drain output, and it is released with enable delay after ENPWM by toggling ENLDO or ENPWM signal. goes high (about 2ms typ.). In normal operation, the POK window is from 90% to its OVP threshold of the converter Over-Temperature Protection reference voltage. Both of VOUT1 and VOUT2 have to stay within this window for POK to be high (AND gated). In When the junction temperature increases above the rising threshold temperature 160°C, the IC will enter the over -temperature protection (OTP). When the OTP occurs, order to prevent false POK drop, capacitors need to parallel at the output to confine the voltage deviation with severe load step transient. REF, LDO and PWM controllers circuitry shuts down. It is non-latch protection. Under-Voltage Protection (PWMs) Current-Limit (PWMs) In the process of operation, if a short-circuit occurs, the The current-limit circuit employs a “valley” current-sensing algorithm (See Figure 1). The APW8811 uses the low- output voltage will drop quickly. When load current is bigger than the value of current-limit threshold, the output side MOSFET’s RDS(ON) of the synchronous rectifier as a current-sensing element. If the magnitude of the current- voltage will fall out of the required regulation range. The under-voltage continually monitors the setting output volt- sense signal at PHASE pin is above the current-limit threshold, the PWM is not allowed to initiate a new cycle. age after soft-start is completed. If a load step is strong enough to pull the output voltage lower than the under- The actual peak current is greater than the current-limit threshold by an amount equal to the inductor ripple voltage threshold for at least 2µs, the PWM controller starts a soft-stop process to shut down the output gradually. As current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the sense long as either of PWM channels triggers under-voltage, both of PWM channels active under-voltage protection resistance, inductor value, and input voltage. put voltage. Under-voltage protection is ignored for at least 2ms(typ.) after a rising edge on EN. Toggling ENLDO or ENPWM signal will clear the latch and bring the chip back to operation. Over-Voltage Protection (OVP) Should the output voltage of VOUT1 and VOUT2 increase over IOUT ∆I ILIMIT 0 25% of the setting voltage due to the high-side MOSFET failure or for other reasons, the over-voltage protection Time Figure 1. Current-Limit Algorithm will active. As long as either of PWM channels triggers over-voltage, both of PWM channels active overvoltage Both PWM controllers use the low-side MOSFETs onresistance RDS(ON) to monitor the current for protection protection. Over-voltage protection will force the low-side MOSFET gate driver fully turn on. This action actively pulls against shorted outputs. The MOSFET’s R is varied DS(ON) by temperature and gate to source voltage, the user down the output voltage. When the OVP occurs, the POK pin will pull down and latch-off the converter. This OVP should determine the maximum R in manufacture’s DS(ON) datasheet. scheme only clamps the voltage overshoot, and does not invert the output voltage when otherwise activated with a Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 IPEAK INDUCTOR CURRENT and latched off when the soft-stop process is completed. The under-voltage threshold is 70% of the nominal out- The current-Limit threshold of APW8811 is adjusted with 13 www.anpec.com.tw APW8811 Function Description (Cont.) Current-Limit (PWMs) (Cont.) an external resistor. The ILIMx pin adjustment range is from 515mV to 2V. In the adjustable mode, the currentlimit threshold voltage is 1/10th the voltage at ILIM pin. As shown in Figure 2, The ILIM pin can source 10µA. The voltage at ILIM pin is equal to 10µA x RILIM. Connect ILIM to REF for a fixed 200mV threshold. The logic current-limit threshold is default to 250mV value if ILIM is 5V. The relationship between the sampled voltage VILIM and the current-limit threshold ILIMIT is given by: 1 × VILIM = ILIMIT × R DS ( ON ) 10 Where V ILIM is the voltage at the ILIM pin. RDS(ON) is the low side MOSFETs conducive resistance. ILIMIT is the setting current-limit threshold. ILIMIT can be expressed as IOUT minus half of peak-to-peak inductor current. The PCB layout guidelines should ensure that noise and DC errors do not corrupt the current-sense signals at PHASE. Place the hottest power MOSEFTs as close to the IC as possible for best thermal coupling. When combined with the under-voltage protection circuit, this current-limit method is effective in almost every circumstance. ILIM VILIM RILIM 10µA 9R VCC TO CURRENT LIMIT LOGIC R Figure 2. Current-Limit Setting Block Diagram Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 14 www.anpec.com.tw APW8811 Function Description (Cont.) Table 1. Operating Mode Truth Table MODE Run CONDITION COMMENT ENLDO = 1, EN PWM =1 PWM is in normal operation. If PWMx is in shutdown, discharge switch (25 Ω) connects their VOUTx to GND. LDOx and REF active. Standby & Soft-Stop ENPWM=0, ENLDO=1 Shutdown PWMx output discharge switch (25 Ω) connects VOUTx to GND. LDOx discharge switch (40 Ω) connects LDOx to GND. In this mode, all circuitry is off. EN PWM=0 and EN LDO=0 UVP The internal 25Ω switch turns on to pull low output voltage. Either VOUT1, or VOUT2 < 70% of nominal LDOx and REF are active. Reset by toggling EN PWM or EN output voltage LDO single. OVP Either VOUT1 and VOUT2 > 125% of normal LGATE of two PWM channel are forced high. LDO and REF output voltage active. Reset by toggling ENPWM or ENLDO single. OTP TJ > +160 oC All circuitry off. It is non-latch protection after the junction temperature cools by 25oC. Table 2. Power-Up Control Logics VENLDO VEN PWM LDO5 LDO3 PWM1 PWM2 Low Don’t Care OFF OFF OFF OFF High Low ON ON OFF OFF High High ON ON ON ON Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 15 www.anpec.com.tw APW8811 Application Information Output Voltage Selection approximately 30% of the maximum output current. The output voltage of PWM1 can be adjusted from2V to 5.5V with a resistor-driver at FB1 between VOUT1 and Once the inductance value has been chosen, selecting an inductor is capable of carrying the required peak cur- GND. Using 1% or better resistors for the resistive divider is recommended. The FB1 pin is the inverter input rent without going into saturation. In some types of inductors, especially core that is made of ferrite, the ripple of the error amplifier, and the reference voltage is 2V. Take the example, the output voltage of PWM1 is deter- current will increase abruptly when it saturates. This will result in a larger output ripple voltage. mined by: VOUTI R = 2 × 1 + TOP1 R GND1 Output Capacitor Selection Output voltage ripple and the transient voltage deviation are factors that have to be taken into consid- Where RTOP1 is the resistor connected from VOUTI to VFB1 and RGND1 is the resistor connected from FB1 to GND. eration when selecting an output capacitor. Higher capacitor value and lower ESR reduce the output ripple Similarly, the output voltage of PWM2 can be also adjusted from 2V to 5.5V. and the load transient drop. Therefore, selecting high performance low ESR capacitors is intended for switch- Output Inductor Selection ing regulator applications. In addition to high frequency noise related MOSFET turn-on and turn-off, the output The duty cycle of a buck converter is the function of the voltage ripple includes the capacitance voltage drop and ESR voltage drop caused by the AC peak-to-peak current. input voltage and output voltage. Once an output voltage is fixed, it can be written as: These two voltages can be represented by: V D = OUT VIN The inductor value determines the inductor ripple current ∆VESR and affects the load transient reponse. Higher inductor value reduces the inductor’s ripple current and induces These two components constitute a large portion of the total output voltage ripple. In some applications, multiple lower output ripple voltage. The ripple current and can be approxminated by: IRIPPLE IRIPPLE 8COUTFSW = IRIPPLE × RESR ∆VCOUT = capacitors have to be paralleled to achieve the desired ESR value. If the output of the converter has to support VIN - VOUT VOUT = × VIN FSW × L another load with high pulsating current, more capacitors are needed in order to reduce the equivalent ESR Where FSW is the switching frequency of the regulator. and suppress the voltage ripple to a tolerable level. A small decoupling capacitor in parallel for bypassing Increasing the inductor value and frequency will reduce the ripple current and voltage. However, there is a the noise is also recommended, and the voltage rating of the output capacitors must also be considered. tradeoff between the inductor’s ripple current and the regulator load transient response time. To support a load transient that is faster than the switching frequency, more capacitors have to be used A smaller inductor will give the regulator a faster load transient response at the expense of higher ripple current. Increasing the switching frequency (FSW ) also to reduce the voltage excursion during load step change. Another aspect of the capacitor selection is that the reduces the ripple current and voltage, but it will increase the switching loss of the MOSFETs and the total AC current going through the capacitors has to be less than the rated RMS current specified on the ca- power dissipation of the converter. The maximum ripple current occurs at the maximum input voltage. A pacitors to prevent the capacitor from over-heating. good starting point is to choose the ripple current to be Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 16 www.anpec.com.tw APW8811 Application Information (Cont.) (CRSS) and maximum output current requirement. The Input Capacitor Selection losses in the MOSFETs have two components: conduction loss and transition loss. For the high-side and low- The input capacitor is chosen based on the voltage rating and the RMS current rating. For reliable operation, select side MOSFETs, the losses are approximately given by the following equations: the capacitor voltage rating to be at least 1.3 times higher than the maximum input voltage. The maximum RMS 2 Phigh-side = IOUT (1+ TC)(RDS(ON))D + (0.5)( IOUT)(VIN)( tSW)FSW current rating requirement is approximately IOUT/2, where IOUT is the load current. During power up, the input capaci- 2 Plow-side = IOUT (1+ TC)(RDS(ON))(1-D) tors have to handle large amount of surge current. In lowduty notebook appliactions, ceramic capacitors are Where I is the load current OUT TC is the temperature dependency of RDS(ON) remmended. The capacitors must be connected between the drain of high-side MOSFET and the source of low- FSW is the switching frequency tSW is the switching interval side MOSFET with very low-impeadance PCB layout. D is the duty cycle Note that both MOSFETs have conduction losses while MOSFET Selection The application for a notebook battery with a maximum voltage of 24V, at least a minimum 30V MOSFETs should the high-side MOSFET includes an additional transition loss. The switching internal, t SW , is the function be used. The design has to trade off the gate charge with the RDS(ON) of the MOSFET: • of the reverse transfer capacitance CRSS. The (1+TC) term is to factor in the temperature dependency of the RDS(ON) For the low-side MOSFET, before it is turned on, the and can be extracted from the “RDS(ON) vs Temperature” curve of the power MOSFET. body diode has been conducted. The low-side MOSFET driver will not charge the miller capacitor of this MOSFET. Layout Consideration In the turning off process of the low-side MOSFET, In any high switching frequency converter, a correct layout the load current will shift to the body diode first. The high dv/dt of the phase node voltage will charge the is important to ensure proper operation of the regulator. With power devices switching at higher frequency, the miller capacitor through the low-side MOSFET driver sinking current path. This results in much less resulting current transient will cause voltage spike across the interconnecting impedance and parasitic circuit switching loss of the low-side MOSFETs. The duty cycle is often very small in high battery voltage elements. As an example, consider the turn-off transition of the PWM MOSFET. Before turn-off condition, the applications, and the low-side MOSFET will conduct most of the switching cycle; therefore, the less MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is freewheeling the RDS(ON) of the low-side MOSFET, the less the power loss. The gate charge for this MOSFET is usually a by the lower MOSFET and parasitic diode. Any parasitic inductance of the circuit generates a large voltage spike secondary consideration. The high-side MOSFET does not have this zero voltage switching during the switching interval. In general, using short and wide printed circuit traces should minimize interconnect- condition, and because it conducts for less time compared to the low-side MOSFET, the switching ing impedances and the magnitude of voltage spike. And signal and power grounds are to be kept separating and loss tends to be dominant. Priority should be given to the MOSFETs with less gate charge, so that both finally combined to use the ground plane construction or single point grounding. The best tie-point between the the gate driver loss and switching loss will be minimized. signal ground and the power ground is at the negative side of the output capacitor on each channel, where there The selection of the N-channel power MOSFETs are de- is less noise. Noisy traces beneath the IC are not recommended. Below is a checklist for your layout: • termined by the RDS(ON), reversing transfer capacitance Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 17 www.anpec.com.tw APW8811 Application Information (Cont.) TQFN4x4-24A Layout Consideration (Cont.) • 4mm Keep the switching nodes (UGATEx, LGATEx, BOOTx, ThermalVia diameter 0.3mm X 4 and PHASEx) away from sensitive small signal nodes (REF, ILIMx, and FBx) since these nodes are fast moving signals. Therefore, keep traces to these nodes as short as possible and there should be no other weak 0.25mm signal traces in parallel with theses traces on any layer. The signals going through theses traces have both 2.25 mm • 0.5mm * high dv/dt and high di/dt, with high peak charging and discharging current. The traces from the gate drivers • 4mm 0.5mm to the MOSFETs (UGATEx and LGATEx) should be short and wide. 2.25 mm 0.46mm Place the source of the high-side MOSFET and the drain of the low-side MOSFET as close as possible. 0.4mm Minimizing the impedance with wide layout plane between the two pads reduces the voltage bounce of • * Just Recommend the node. Decoupling capacitor, the resistor dividers, boot capacitors, and current-limit stetting resistor should be close to their pins. (For example, place the decoupling ceramic capacitor near the drain of the high-side MOSFET as close as possible. The bulk • capacitors are also placednear the drain.) The input capacitor should be near the drain of the upper MOSFET; the high quality ceramic decoupling capacitor can be put close to the VCC and GND pins; the output capacitor should be near the loads. The input capacitor GND should be close to the output ca- • pacitor GND and the lower MOSFET GND. The drain of the MOSFETs (VIN and PHASEx nodes) should be a large plane for heat sinking. And PHASEx pin traces are also the return path for UGATEx. Con- • nect these pins to the respective converter’s upper MOSFET source. The controller used ripple mode control. Build the resistor divider close to the FB1 pin so that the high impedance trace is shorter when the output voltage is in adjustable mode. And the FB1 pin traces can’t be • close to the switching signal traces (UGATEx, LGATEx, BOOTx, and PHASEx). The PGND trace should be a separate trace, and independently go to the source of the low-side MOSFETs for current-limit accuracy. Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 18 www.anpec.com.tw APW8811 Package Information D A E b TQFN4x4-24A Pin 1 A1 A3 NX D2 aaa c L K E2 Pin 1 Corner e S Y M B O L A A1 TQFN4x4-24A MILLIMETERS MIN. INCHES MAX. MIN. MAX. 0.70 0.80 0.028 0.032 0.00 0.05 0.000 0.002 A3 0.20 REF 0.008 REF 0.007 0.012 b 0.18 0.30 D 3.90 4.10 0.154 0.161 0.098 D2 2.00 2.50 0.079 E 3.90 4.10 0.154 0.161 E2 2.00 2.50 0.079 0.098 e 0.50 BSC L 0.35 K 0.20 aaa Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 0.020 BSC 0.014 0.45 0.018 0.008 0.08 0.003 19 www.anpec.com.tw APW8811 Carrier Tape & Reel Dimensions P0 P2 P1 A B0 W F E1 OD0 K0 A0 A OD1 B B T SECTION A-A SECTION B-B H A d T1 Application TQFN4x4-24A A H T1 C d D W E1 F 330.0±2.00 50 MIN. 12.4+2.00 -0.00 13.0+0.50 -0.20 1.5 MIN. 20.2 MIN. 12.0±0.30 1.75±0.10 5.5±0.05 P0 P1 P2 D0 D1 T A0 B0 K0 2.0±0.05 1.5+0.10 -0.00 1.5 MIN. 0.6+0.00 -0.40 4.30±0.20 4.30±0.20 1.25±0.20 4.0±0.10 8.0±0.10 (mm) Devices Per Unit Package Type Unit Quantity TQFN4x4-24A Tape & Reel 3000 Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 20 www.anpec.com.tw APW8811 Taping Direction Information TQFN4x4-24A USER DIRECTION OF FEED Classification Profile Supplier Tp≧Tc User Tp≦Tc TC TC -5oC User tp Supplier tp Tp tp Temperature Max. Ramp Up Rate = 3oC/s Max. Ramp Down Rate = 6oC/s TL Tsmax TC -5oC t Preheat Area Tsmin tS 25 Time 25oC to Peak Time Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 21 www.anpec.com.tw APW8811 Classification Reflow Profiles Profile Feature Sn-Pb Eutectic Assembly Pb-Free Assembly 100 °C 150 °C 60-120 seconds 150 °C 200 °C 60-120 seconds 3 °C/second max. 3 °C/second max. 183 °C 60-150 seconds 217 °C 60-150 seconds See Classification Temp in table 1 See Classification Temp in table 2 Time (tP)** within 5°C of the specified classification temperature (Tc) 20** seconds 30** seconds Average ramp-down rate (Tp to Tsmax) 6 °C/second max. 6 °C/second max. 6 minutes max. 8 minutes max. Preheat & Soak Temperature min (Tsmin) Temperature max (Tsmax) Time (Tsmin to Tsmax) (ts) Average ramp-up rate (Tsmax to TP) Liquidous temperature (TL) Time at liquidous (tL) Peak package body Temperature (Tp)* Time 25°C to peak temperature * Tolerance for peak profile Temperature (Tp) is defined as a supplier minimum and a user maximum. ** Tolerance for time at peak profile temperature (tp) is defined as a supplier minimum and a user maximum. Classification Reflow Profiles (Cont.) Table 1. SnPb Eutectic Process – Classification Temperatures (Tc) Package Thickness <2.5 mm ≥2.5 mm Volume mm <350 235 °C 220 °C 3 Volume mm ≥350 220 °C 220 °C 3 Table 2. Pb-free Process – Classification Temperatures (Tc) Package Thickness <1.6 mm 1.6 mm – 2.5 mm ≥2.5 mm Volume mm <350 260 °C 260 °C 250 °C 3 Volume mm 350-2000 260 °C 250 °C 245 °C 3 Volume mm >2000 260 °C 245 °C 245 °C 3 Reliability Test Program Test item SOLDERABILITY HOLT PCT TCT HBM MM Latch-Up Method JESD-22, B102 JESD-22, A108 JESD-22, A102 JESD-22, A104 MIL-STD-883-3015.7 JESD-22, A115 JESD 78 Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 22 Description 5 Sec, 245°C 1000 Hrs, Bias @ Tj=125°C 168 Hrs, 100%RH, 2atm, 121°C 500 Cycles, -65°C~150°C VHBM≧2KV VMM≧200V 10ms, 1tr≧100mA www.anpec.com.tw APW8811 Customer Service Anpec Electronics Corp. Head Office : No.6, Dusing 1st Road, SBIP, Hsin-Chu, Taiwan, R.O.C. Tel : 886-3-5642000 Fax : 886-3-5642050 Taipei Branch : 2F, No. 11, Lane 218, Sec 2 Jhongsing Rd., Sindian City, Taipei County 23146, Taiwan Tel : 886-2-2910-3838 Fax : 886-2-2917-3838 Copyright ANPEC Electronics Corp. Rev. A.3 - Sep., 2012 23 www.anpec.com.tw